阅读说明：本技术 三电平两级解耦有源npc转换器 (Three-level and two-stage decoupling active NPC converter ) 是由 张迪 何江彪 萨钦·玛杜苏德哈南 于 2018-06-05 设计创作，主要内容包括：一种系统包括电压转换器和用于控制电压转换器的操作的控制器。电压转换器包括多个分支,其中每个分支包括第一组和第二组硅(Si)基功率器件。第一组硅基功率器件包括在第一互连节点处彼此连接的第一和第二硅基功率器件,并且第二组硅基功率器件包括在第二互连节点处彼此连接的第三和第四硅基功率器件。第一组和第二组硅基功率器件分别跨第一和第二DC电压源联接。第一组碳化硅(SiC)基功率器件跨第一互连节点和第二互连节点联接。该系统还包括跨第一和第二互连节点连接的缓冲电容器。(A system includes a voltage converter and a controller for controlling operation of the voltage converter. The voltage converter includes a plurality of branches, where each branch includes a first set and a second set of silicon (Si) -based power devices. The first group of silicon-based power devices includes first and second silicon-based power devices connected to each other at a first interconnect node, and the second group of silicon-based power devices includes third and fourth silicon-based power devices connected to each other at a second interconnect node. The first and second groups of silicon-based power devices are coupled across first and second DC voltage sources, respectively. A first set of silicon carbide (SiC) -based power devices is coupled across the first and second interconnect nodes. The system also includes a buffer capacitor connected across the first and second interconnect nodes.)

1. A system, comprising:

a voltage converter having a plurality of branches, each branch comprising:

a first set of silicon (Si) -based power devices coupled across a first Direct Current (DC) voltage source, wherein the first set of silicon-based power devices includes a first silicon-based power device and a second silicon-based power device connected to each other at a first interconnect node;

a second group of silicon-based power devices coupled across a second DC voltage source, wherein the second group of silicon-based power devices includes a third silicon-based power device and a fourth silicon-based power device connected to each other at a second interconnect node;

a first set of silicon carbide (SiC) -based power devices coupled across the first and second interconnect nodes, wherein the first set of silicon carbide-based power devices includes first and second silicon carbide-based power devices connected to each other at a third interconnect node;

a buffer capacitor connected across the first and second interconnect nodes;

wherein the first and second groups of silicon-based power devices are connected to each other at a fourth interconnect node; and

a controller to control operation of the voltage converter.

2. The system of claim 1, wherein the controller controls the gate pulse based on an output current polarity and an output voltage polarity.

3. The system of claim 1, wherein the controller operates the voltage converter in a plurality of operating states including a positive state, a P-type zero state, a negative state, or an N-type zero state at any given time.

4. The system of claim 3, wherein the controller is configured to, during each of the operating states, turn on one silicon-based power device from the first set of silicon-based power devices and turn on another silicon-based power device from the second set of silicon-based power devices.

5. The system of claim 4, wherein the controller is configured to turn on one of the silicon carbide-based power devices such that the one silicon carbide-based power device and the one silicon-based power device that was turned on during the corresponding operating state together carry the load current in series.

6. The system of claim 5, wherein the silicon-based power device and the silicon carbide-based power device carrying the load current in series comprise: i) the first silicon-based power device and the first silicon carbide-based power device during the positive state, ii) the third silicon-based power device and the second silicon carbide-based power device during the P-type zero state, iii) the fourth silicon-based power device and the second silicon carbide-based power device during the negative state, and iv) the second silicon-based power device and the first silicon carbide-based power device during the N-type zero state.

7. The system of claim 6, wherein when the load current is to be commutated from one silicon carbide-based power device to another, the controller is configured to cause the silicon-based power device and the silicon carbide-based power device carrying the load current in series to be turned off simultaneously.

8. The system of claim 7, wherein the controller is configured to turn on the silicon-based power device that has been simultaneously turned off with the silicon carbide-based power device after damping a resonance formed between the snubber capacitor of the voltage converter and the silicon-based power device.

9. The system of claim 7, wherein the controller is configured to cause the silicon-based power device and the silicon carbide-based power device carrying the load current in series to be simultaneously turned off if an output voltage polarity and an output current polarity of the voltage converter are opposite to each other or if the output voltage is zero and the output current polarity is positive.

10. The system of claim 5, wherein the controller is configured to cause the silicon-based power devices and the silicon carbide-based power devices that are turned on to be simultaneously turned off if the output voltage polarity and the output current polarity of the voltage converter are opposite.

11. The system of claim 3, wherein in the positive state, a voltage at the third interconnect node is positive relative to the fourth interconnect node, and in the negative state, the voltage at the third interconnect node is negative.

12. The system of claim 11, wherein in the P-type zero state and the N-type zero state, a voltage at the third interconnect node is zero relative to the fourth interconnect node.

13. The system of claim 12, wherein the controller is configured to turn on the third silicon-based power device in the P-type zero state and to turn on the second silicon-based power device in the N-type zero state.

14. A system, comprising:

a voltage converter having a plurality of branches, each branch comprising:

a first set of silicon (Si) based power devices coupled to a first DC voltage source;

a second set of silicon-based power devices coupled to a second DC voltage source, wherein the first set of silicon-based power devices and the second set of silicon-based power devices are coupled in series;

a first set of silicon carbide (SiC) -based power devices coupled to the first set of silicon-based power devices and the second set of silicon-based power devices;

a snubber capacitor connected across the first set of silicon carbide-based power devices; and

a processor configured to control the switching of each silicon carbide-based power device of the first set of silicon carbide-based power devices and each silicon-based power device of the first set of silicon-based power devices and the second set of silicon-based power devices such that, at any given time, one silicon carbide-based power device of the first set of silicon carbide-based power electronic devices conducts current in series with one silicon-based power electronic device of the first set of silicon-based power electronic devices or the second set of silicon-based power electronic devices.

15. The system of claim 14, wherein when the current is to be commutated from one silicon carbide-based power device to another silicon carbide-based power device, the controller is configured to simultaneously turn off the one silicon carbide-based power device of the first set of silicon carbide-based power electronics and the one silicon-based power electronics of the first or second set that conduct the current in series.

16. The system of claim 15, wherein the controller is configured to turn on one silicon-based power electronic device that has been turned off simultaneously with the one silicon carbide-based power electronic device after damping a resonance formed between the snubber capacitor of the voltage converter and the silicon-based power electronic device.

17. The system of claim 15, wherein the controller causes the one silicon carbide-based power electronic device and the one silicon-based power electronic device that conduct the current in series to be simultaneously turned off if an output voltage polarity and an output current polarity of the voltage converter are opposite to each other or if the output voltage is zero and the output current polarity is positive.

18. The system of claim 15, wherein the controller is configured to cause the silicon-based power electronics and the silicon carbide-based power electronics that are turned on to be simultaneously turned off if the output voltage polarity and the output current polarity of the voltage converter are opposite.

19. The system of claim 14, wherein each of the first and second groups of silicon-based power devices comprises one or more Insulated Gate Bipolar Transistors (IGBTs).

20. The voltage converter of claim 14, wherein the first set of silicon carbide-based power devices comprises one or more Metal Oxide Semiconductor Field Effect Transistors (MOSFETs).

Technical Field

Embodiments of the present technology relate to power converters, and more particularly, to a method of operating a three-level two-stage decoupled active Neutral Point Clamped (NPC) converter.

Background

Turbine electric or hybrid propulsion may improve the energy conversion efficiency of the aircraft, reduce carbon emissions, and reduce reliance on carbon-based fuels. Megawatt (MW) class lightweight, high efficiency, high reliability power converters are important components of hybrid propulsion. Furthermore, to reduce cable weight, the power converter needs to withstand moderate Direct Current (DC) voltage stress. Further, in order to reduce the weight of the motor used in hybrid propulsion, the fundamental output frequency of the power converter needs to be high, for example, higher than 1kHz, and thus the power converter is required to have a higher switching frequency.

Silicon carbide (SiC) is a semiconductor that is increasingly used in power electronic devices such as Metal Oxide Semiconductor Field Effect Transistors (MOSFETs) to meet the high switching frequency and efficiency requirements of power converters. Furthermore, a three-level NPC converter topology is the preferred choice for power converters because it can achieve higher voltage ratings without having to sequence the devices; better harmonic performance to reduce the weight of the filter; and lower switching losses to achieve high efficiency.

In order to design such a three-level converter, the most important challenge is how to minimize the commutation loop inductance. In addition to the commutation loop inductance, SiC MOSFET-based multi-level converters face other challenges compared to Si IGBT modules. SiC MOSFET modules are much more expensive than Si IGBT modules and have an inherent thermal imbalance characteristic. Methods have been proposed to effectively alleviate the thermal imbalance problem. However, such an approach transfers more switching action from the short commutation loop to the long commutation loop. Such an approach will result in higher system level switching losses.

Accordingly, a system and method that addresses the above-mentioned problems is desired.

Disclosure of Invention

In accordance with an embodiment of the present technique, a system is provided having a voltage converter and a controller that controls operation of the voltage converter. The voltage converter includes a plurality of branches, wherein each branch includes first and second sets of silicon (Si) -based power devices connected across first and second Direct Current (DC) voltage sources, respectively, and to each other at a fourth interconnection node. The first group of silicon-based power devices includes first and second silicon-based power devices connected to each other at a first interconnect node. In addition, the second set of silicon-based power devices includes third and fourth silicon-based power devices connected to each other at a second interconnect node. The system also includes a first set of silicon carbide (SiC) -based power devices coupled across the first and second interconnect nodes, wherein the first set of silicon carbide-based power devices includes first and second silicon carbide-based power devices connected to each other at a third interconnect node. The system also includes a buffer capacitor connected across the first and second interconnect nodes.

In accordance with another embodiment of the present technique, a system is provided having a voltage converter and a processor controlling operation of the voltage converter. The voltage converter includes a plurality of branches, wherein each branch includes a first set of silicon (Si) -based power devices coupled to a first DC voltage source and a second set of silicon-based power devices coupled to a second DC voltage source. The first group of silicon-based power devices and the second group of silicon-based power devices are coupled in series. The voltage converter also includes a first set of silicon carbide (SiC) -based power devices coupled with the first set of silicon-based power devices and the second set of silicon-based power devices and a snubber capacitor connected across the first set of silicon carbide-based power devices. The processor is configured to control the switching of each silicon carbide-based power device of the first group of silicon carbide-based power devices and each silicon-based power device of the first group of silicon-based power devices and the second group of silicon-based power devices such that, at any given time, one silicon carbide-based power device of the first group of silicon carbide-based power devices conducts current in series with one silicon-based power electronic device of the first group of silicon-based power electronic devices or the second group of silicon-based power electronic devices.

Drawings

These and other features and aspects of embodiments of the present invention will be better understood when the following detailed description is read with reference to the accompanying drawings in which like characters represent like parts throughout the drawings, wherein:

fig. 1 shows a schematic diagram of a three-level two-stage decoupled active neutral point clamped (3L-TDANPC) converter system according to an embodiment;

FIG. 2 is a diagram of a timing diagram of a voltage signal output by the 3L-TDANPC converter system and a gate signal provided to a switching device in the hybrid converter system of FIG. 1, according to an embodiment;

FIG. 3 illustrates a schematic diagram of four basic operating states (positive, P-type zero, negative, and N-type zero) of a 3L-TDANPC converter according to an embodiment;

FIG. 4 shows a schematic diagram depicting a commutation loop for a 3L-TDANPC converter in accordance with an embodiment;

FIG. 5 shows a schematic diagram depicting a commutation process without damping in a 3L-TDANPC converter, in accordance with an embodiment;

FIG. 6 shows a schematic diagram depicting a commutation process with single pulse damping technique in a 3L-TDANPC converter, in accordance with an embodiment;

FIG. 7 shows a schematic diagram depicting a commutation process without any damping mechanism in a 3L-TDANPC converter with output voltage positive and output current negative, in accordance with an embodiment;

FIG. 8 shows a schematic diagram depicting a commutation process with gate-off active damping technique in a 3L-TDANPC converter, in accordance with an embodiment;

FIG. 9 illustrates a graphical diagram of simulation results for a thermal modeling based 3L-TDANPC converter, according to an embodiment;

FIG. 10 shows a graphical diagram of an efficiency curve of a 3L-TDANPC converter based on simulation results according to an embodiment; and

FIG. 11 shows a schematic diagram depicting fault current in a 3L-TDANPC converter during a short circuit fault, according to an embodiment.

Detailed Description

Unless defined otherwise, technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this disclosure belongs. As used herein, the terms "first," "second," and the like, do not denote any order, quantity, or importance, but rather are used to distinguish one element from another. Also, the terms "a" and "an" do not denote a limitation of quantity, but rather denote the presence of at least one of the referenced item. The term "or" is intended to be inclusive and mean one, some, or all of the listed items. The use of "including," "comprising," or "having" and variations thereof herein is meant to encompass the items listed thereafter and equivalents thereof as well as additional items. The terms "connected" and "coupled" are not restricted to physical or mechanical connections or couplings, and may include direct or indirect electrical connections or couplings. Further, the terms "circuit" and "circuitry" and "controller" may include a single component or multiple components that are active and/or passive and are connected or otherwise coupled together to provide the described functionality.

As used herein, the term "module" refers to software, hardware or firmware, or any combination thereof, or any system, process or function that performs or facilitates the processes described herein.

A topology is proposed, which is referred to as a three-level two-stage decoupled active neutral point clamped (3L-TDANPC) converter. The 3L-TDANPC converter is internally provided with an IGBT module and a SiC MOSFET module. The commutation loops of the SiC MOSFETs are all within the module to keep switching losses low. In addition, the use of the IGBT module contributes to a significant reduction in system cost, and can limit the short-circuit current flowing through the SiC MOSFET to a very safe level.

By way of introduction, fig. 1 shows a schematic diagram of a three-level two-stage decoupled active neutral point clamped (3L-TDANPC) converter system 10 according to an embodiment of the present method, the converter system 10 incorporating Si and SiC power electronics to convert a DC voltage signal to an AC voltage signal. It should be noted that the schematic diagram of fig. 1 may represent one phase leg of a multi-phase converter system. Thus, the 3L-TDANPC converter system 10 may be employed on one or more branches (legs) of such a multiphase converter system.

In one embodiment, the Si power electronics and SiC power electronics of the 3L-TDANPC converter system 10 may be the Si IGBT 12 and SiC MOSFET14, respectively. The Si IGBT 12 may include various types of IGBTs having different ratings (e.g., 1.7kV, 3.3kV, 4.5kV, or 6.5kV IGBTs) that switch between a conducting state and a non-conducting state using Si as a semiconductor material. In the same manner, SiC MOSFETs may include various types of MOSFETs having different ratings that switch between a conductive state and a non-conductive state using SiC as a semiconductor material.

In some embodiments, multiple Si IGBTs 12 may be grouped together as part of the module 16. For example, in converter system 10, two Si IGBTs 12 may be electrically coupled in series with each other and provide three interconnection nodes (e.g., 11, 13, 15), where module 16 may be coupled to other electrical components. The interconnection nodes may be located on the collector side of one of the Si IGBTs 12, on the emitter side of one of the Si IGBTs 12, and between two Si IGBTs 12.

In one embodiment, the first Si IGBT T1 is connected to the second Si IGBT T2 at a first interconnection node 13. Similarly, the third Si IGBT T3 is connected to the fourth Si IGBT T4 at a second interconnection node 19. Further, the emitter of the first Si IGBTT1 is connected to the collector of the second Si IGBT T1 at interconnection node 13, and the emitter of the third Si IGBT T3 is connected to the collector of the fourth Si IGBT T4 at interconnection node 19. Furthermore, the two modules 16 of the Si IGBT are connected to each other at a fourth interconnection node 15.

In the same manner, a plurality of SiC MOSFETs 14 may be grouped together as part of module 18 such that two SiC MOSFETs 14(T5 and T6) may be electrically coupled to each other in series at third interconnect node 17. Further, the module 18 may be connected across the first and second interconnection nodes 13, 19. The interconnect node 17 of module 18 may be located where the drain side of SiC MOSFET T5 is connected to the source side of SiC MOSFET T6. Furthermore, a load may be connected to the interconnection node 17.

Each Si IGBT module 16 of the converter system 10 may be coupled across a DC voltage source (e.g., DC voltage source 20, DC voltage source 22). The interconnect nodes (e.g., 13, 19) or outputs of each Si IGBT module 16 can then be coupled in series with the SiC MOSFET module 18. For example, the interconnection node between the two Si IGBTs 12 of module 16 may be coupled to the source and drain sides of the SiCMOSFET of module 18. A buffer capacitor 32 is also placed between the two interconnection nodes 13 and 19. The snubber capacitor 32 divides the large commutation loop for the Si IGBT and SiC MOSFET (without the snubber capacitor 32) into two smaller loops, as will be discussed in subsequent paragraphs. The commutation loop inductance of the SiC MOSFET can be minimized by means of the snubber capacitor. Thus, even if the majority of the 3L-TDANPC converter is a Si IGBT, very low switching losses can be achieved.

The AC output voltage of converter system 10 may be provided at output terminals (e.g., 21, 23) connected to an interconnection node (e.g., 17) between the SiC MOSFETs of module 18 and to an interconnection node (e.g., 15) between voltage source 20 and voltage source 22. In some embodiments, voltage source 20 and voltage source 22 both provide the same amount of DC voltage. In this way, the Si IGBT 12 and the SiC MOSFET14 can be switched on and off in a controlled manner to convert the DC voltage signal provided via the voltage sources 20 and 22 to an AC voltage signal output by the converter system 10. The AC voltage signal output may then be provided to various types of AC powered devices, such as AC motors and the like, to perform various types of operations.

In one embodiment, the switching of Si IGBT 12 and SiC MOSFET14 may be controlled by a gate signal provided to the gates of Si IGBT 12 and SiC MOSFET 14. As such, the converter system 10 may include a converter control system (or controller) 24 that may provide a gate signal to each of the Si IGBTs 12 and SiC MOSFETs 14 in the converter system 10 to control operation of the converter system 10.

Converter control system 24 may generally include a processor 26 that determines appropriate gate signals to provide to Si IGBT 12 and SiC MOSFET14 of converter system 10 to produce a desired AC voltage output signal using DC voltage sources 20 and 22. The processor 26 may be any type of computer processor or microprocessor capable of executing computer-executable instructions (e.g., software code, programs, applications). The processor 26 may also include multiple processors that may cooperate to perform the operations described below.

In general, as described above, the processor 26 may execute a software application that includes a program that determines the gate signals to provide to the Si IGBT 12 and SiC MOSFET14 so that the resulting AC voltage output corresponds to the desired voltage signal. For example, fig. 2 shows an example timing diagram 30 of gate signals provided to the respective gates of the Si IGBT 12 and the SiC MOSFET14 of the embodiment of the converter system 10 of fig. 1.

However, when the gate signal is removed from the respective Si IGBT 12 and the Si IGBT turns off or enters a non-conductive state, the corresponding SiC MOSFET14 coupled in series with the respective Si IGBT 12 will have turned off. That is, the corresponding SiCMOSFET 14 may be in a non-conductive state earlier than its corresponding Si IGBT 12. In this way, when the gate signal of the corresponding Si IGBT 12 is removed, the current in the Si IGBT 12 is already zero since the SiC MOSFET14 has been turned off. As a result, the Si IGBT 12, which conventionally has a higher switching loss compared to the SiC MOSFET, has almost no loss during turn-off.

In certain embodiments, the processor 26 may provide gate signals to the Si IGBTs 12 and SiC MOSFETs 14 such that one Si IGBT 12 will be in series with one SiC MOSFET14 at any given time. In addition, the processor 26 may send gate signals to the Si IGBT 12 to cause the Si IGBT 12 to switch at a fundamental line frequency (e.g., 60Hz), and may send gate signals to the SiC MOSFET14 to switch at a higher frequency (e.g., >1kHz) to synthesize a desired AC voltage output waveform. Therefore, as shown in fig. 2, the frequency of change of the gate signals (e.g., G1, G2, G3, G4) supplied to the Si IGBT 12 is lower than the gate signals (e.g., G5, G6) supplied to the SiC MOSFET 14. As a result, the AC voltage output 40 (reference wave) may correspond to a desired sine wave, as shown in fig. 2. Fig. 2 also depicts a carrier wave 42. The intersection of the carrier 42 and the AC voltage output 40 typically forms a square waveform or Pulse Width Modulated (PWM) gate waveform that can be used to control the Si IGBT 12 and the SiC MOSFET 14.

It should be noted that fig. 2 is merely one example of a gate signal and that other methods of generating a gate signal are within the scope of the present technology, as described below. In one embodiment, the processor 26 may coordinate the gate signals provided to the Si IGBT 12 and the SiC MOSFET14 based on the output voltage polarity and the output current direction of the 3L-TDANPC converter 10. For example, if the output voltage polarity is positive (i.e., output terminal 21 is positive with respect to output terminal 23) and the output current is positive (i.e., the output current is flowing from the converter), the processor may provide a gate signal such that switches T1 and T5(SiC MOSFETs) are simultaneously turned off during commutation of switch T5(SiC MOSFETs). As described in the following paragraph, this results in damping of the LC resonance formed during commutation of switch T5. In one embodiment, the sensor 34 may be used to determine the direction of the output current.

FIG. 3 shows a schematic diagram of four basic operating states of a 3L-TDANPC converter according to an embodiment, namely: a positive (P) state (50); a P-type zero state (52); a negative (N) state (54); and an N-type zero state (56). As previously described, the output voltage of the 3L-TDANPC converter is an AC voltage waveform. Here, the positive state means that the voltage at the output terminal 58 is positive with respect to the DC terminal 60, and the negative state means that the voltage at the output terminal 58 is negative with respect to the DC terminal 60. Further, a P-type zero state refers to a zero voltage at output terminal 58 with respect to DC terminal 60 with switch T3 in the on position, and an N-type zero state refers to a zero voltage at output terminal 58 with respect to DC terminal 64 with switch T2 (rather than T3) in the on position.

In each operating state, the controller is configured to turn on one silicon-based power device (i.e., switch T1 or switch T2) from the first group of silicon-based power devices and turn on another silicon-based power device (i.e., switch T3 or T4) from the second group of silicon-based power devices. Further, in each operating state, the controller is configured to turn on one of the silicon carbide based power devices, i.e., switch T5 or switch T6, such that one of the silicon carbide based power devices and the turned on one of the silicon based power devices together carry the load current in series.

As can be seen from fig. 3, the silicon-based power device and the silicon carbide-based power device carrying the load current in series comprise: i) switches T1 and T5 during the positive state 50, ii) switches T3 and T6 during the P-type zero state 52, iii) switches T4 and T6 during the negative state 54, and iv) switches T2 and T5 during the N-type zero state 56.

When the 3L-TDANPC converter switches from the positive state 50 to the P-type zero state 52, current from switch T5 is commutated to switch T6 via the snubber capacitor 62. Similarly, when the converter switches from the P-type zero state 52 to the positive state 50, current from switch T6 is commutated via the snubber capacitor 62 to switch T5. Thus, as shown in fig. 4, a commutation loop 72 for the SiC MOSFET is between the switches T5, T6 and the buffer capacitor 62. It should be noted that during the positive state 50 and the P-type zero state 52, the Si switches T1 and T3 are always on.

When the polarity of the output voltage changes from positive to negative, the P-state 50 switches to the N-state 54 via the P-type zero state 52. In the negative state 54, the Si switches T2 and T4 are turned on. In other words, during the voltage change from positive to negative polarity, the current from the Si switches T1 and T3 is commutated to the switches T2 and T4. Similarly, during the voltage change from negative to positive polarity, the current from the Si switches T2 and T4 is commutated to the switches T1 and T3. Although the switches T1, T2, T3, and T4 experience hard switching (e.g., via the commutation loop 74 as shown in fig. 4), the switching losses in them are very limited because they only switch at fundamental frequencies (e.g., 60 Hz). A commutation loop 74 is formed between the snubber capacitor 62, the Si switch and a DC link capacitor 76 of the DC voltage source.

Due to parasitic inductances in the converter, resonant currents will be observed during switching of the 3L-TDANPC converter, causing thermal and electromagnetic interference (EMI) problems if not properly damped, especially under heavy load conditions. The parasitic inductance in the converter depends on the system configuration and mechanical layout. Accordingly, a mitigation method of damping such resonances is described herein.

Fig. 5 shows a schematic diagram describing the commutation process in a 3L-TDANPC converter without any damping mechanism. In the case where the output voltage and output current of the 3L-TDANPC converter are both positive, the schematic diagram 80 depicts the "forward" operation of the 3L-TDANPC converter, the schematic diagram 82 depicts the "while commutating" operation of the 3L-TDANPC converter, and the schematic diagram 84 represents the "reverse" operation of the 3L-TDANPC converter. It should be noted that each of the schematics 80, 82 and 84 show a simplified form of the 3L-TDANPC converter, and only a portion thereof. Further, the dark portion in each diagram represents the path of the load current.

The diagram 80 shows that before the commutation process starts, i.e. at the current I_LBefore commutation from switch T5 to switch T6, switches T1 and T5 carry the load current I_L. At a current I_LAfter commutation from T5 to T6, the current I_LWill continue to flow through switch T1 and charge the buffer capacitor 62 as shown in schematic 82. Thus, the snubber capacitor voltage increases and the voltage difference between the snubber capacitor and the DC bus capacitor 76 will decrease the current in switch T1 and increase the current in switch T3. After the current in the switch T1 reaches zero, the load current I_LIs fully commutated to switch T3. At this time, however, the buffer capacitor 62 is charged at a higher voltage than the DC bus voltage across capacitor 76, and this voltage difference begins to drive a negative current through switch T1 and increase the current in switch T3, thereby initiating the LC resonance, as shown in diagram 84. The resonant frequency of the LC resonance is determined by the inductance of the commutation loop 78 and the buffer capacitor 62. For example, if the inductance (L) of commutation loop 78 is 115nH,and the buffer capacitor capacitance value (C) is 2.72 μ F, the resonant frequency F of the LC resonance can be given by:

in addition, the voltage variation (Δ V) of the buffer capacitor_snub) Is determined by the value (I) of the load current. For the above case, if the load current value is 600A:

if the DC bus voltage is 1000V, the buffer capacitor voltage will be in the range of 877V to 1123V.

It should be noted that without any damping mechanism, the energy of the LC resonance will eventually be damped due to the impedance in the commutation loop. However, it is desirable to dissipate most of the energy in the Si devices, rather than other components in the commutation loop, because it is much easier to cool the Si GBT module than other components such as buffer capacitors. In addition, to minimize any EMI related issues, it is desirable to damp the resonance as quickly as possible. Thus, according to embodiments of the present technique, a single pulse damping method is used.

Fig. 6 shows a schematic diagram depicting a commutation process with single pulse damping technique in a 3L-TDANPC converter. In the case where both the output voltage and the output current of the 3L-TDANPC converter are positive, the schematic 90 depicts the "forward" operation of the 3L-TDANPC converter, while the schematic 92 depicts the "backward" operation of the 3L-TDANPC converter. It should be noted that each of the schematics 90 and 92 show a simplified form of the 3L-TDANPC converter, and only a portion thereof. Further, the dark portion in each diagram represents the path of the load current.

In the single pulse damping method, the processor 26 turns off the switch T1 together with or simultaneously with the switch T5 in order to damp the LC resonance formed during the commutation of the current from the switch T5 to T6. It should be noted that in fig. 2, applies toThe gate pulses G1 and G3 of switches T1 and T3 are shown to be the same. At this point, however, i.e., during the single pulse damping method, G1 is zero and G3 is still high, so switch T3 remains on despite switch T1 being turned off. Since switch T1 is in series with the loop inductance of commutation loop 78, turning off T1 means breaking the current in the loop inductance. The current in the open loop inductor induces an additional voltage across switch T1. This additional voltage across switch T1 forces the current through T1 to zero, thereby forcing the load current I_LCommutates to switch T3 as shown in schematic 92. Once the resonance is damped, the switch T1 can be turned on again immediately. The entire process of damping the resonance takes only a little time. In one embodiment, the resonance is damped to less than 1 μ s. Therefore, the switches T1 and T5 are turned off at the same time only for the purpose of canceling the resonance current as fast as possible.

It should be noted that fig. 6 discloses the operation of the 3L-TDANPC converter when the output voltage is positive and the output current is positive. However, in other cases, for example, when the output voltage is zero and the current is positive and when both the output voltage and the current are negative, the same method may be employed. In other words, the single pulse damping method described herein will damp the resonance as long as the output voltage polarity and the output current direction are the same or at least when the output voltage is zero.

Fig. 7 shows a schematic diagram of the commutation process without any damping mechanism in the 3L-TDANPC converter in case the output voltage is positive and the output current is negative. Diagram 100 depicts a "forward" operation of the 3L-TDANPC converter, diagram 102 depicts a "when commutating" operation of the 3L-TDANPC converter, and diagram 104 represents a "backward" operation of the 3L-TDANPC converter. It should be noted that each of the schematics 100, 102 and 104 show a simplified form of the 3L-TDANPC converter, and only a portion thereof. Further, the dark portion in each diagram represents the path of the load current.

When the output voltage is positive and the output current is negative, as shown in the diagram 100, the load current I_LThrough the anti-parallel diode 108 of switch T1. Therefore, even if the switch T1 is turned off, electricity is suppliedThe flow is not interrupted. As shown in the diagram 102, when current from switch T5 is commutated to switch T6, the load current commutated to switch T6 begins to discharge the snubber capacitor 62. The snubber capacitor 62 will be discharged and the voltage difference between the snubber capacitor 62 and the DC capacitor 76 will commutate the current in the switch T1 to the switch T3 until the current in the switch T1 decreases to zero. Thereafter, a resonant circuit will be formed between the snubber capacitor 62 and the switches T1 and T3, as shown in the schematic 104, since both switches T1 and T3 are still on. As mentioned before, without any damping mechanism, the resonant current 106 in the resonant circuit will eventually disappear due to the impedance of the resonant circuit. However, it is desirable to cancel the resonance as quickly as possible to minimize any EMI related issues.

Fig. 8 shows a schematic diagram describing the commutation process in a 3L-TDANPC converter employing Gate-Off active damping (Gate-Off active damping) technique. In the case where the output voltage is positive and the output current is negative, the diagram 110 depicts the "forward" operation of the 3L-TDANPC converter, while the diagram 112 depicts the "backward" operation of the 3L-TDANPC converter. It should be noted that each of the schematics 110 and 112 show a simplified form of the 3L-TDANPC converter, and only a portion thereof. Further, the dark portion in each diagram represents the path of the load current.

In the gate-off active damping method, the processor 26 turns off the switches T1 and T3 together throughout the entire switching cycle in which current is commutated from switch T5 to T6, or simultaneously with switch T5, to damp the LC resonance formed during commutation of current from switch T5 to T6. Since the switch T1 is in the off state, the resonant current 106 cannot reverse after commutating the current in the switch T1 to the switch T3 as shown in diagram 112. After this process, the buffer capacitor 62 is discharged via the load current, and the voltage difference between the DC capacitor 76 and the buffer capacitor 62 is blocked by the diode 108 of the switch T1. A similar procedure is followed when current is to be commutated from switch T6 to switch T5.

It should be noted that the single pulse damping method can also be used in cases where the output voltage polarity is different from the output current polarity if the current direction is unknown, e.g. when the current magnitude is not high enough to determine its direction. In this case, if resonance does occur, the effect is very limited because the current level itself is low.

FIG. 9 shows a graphical representation 120 of simulation results for a 3L-TDANPC converter based on a thermal model. The simulation was performed at a nominal DC bus voltage of 2400V. The horizontal axis 122 of the graph 120 shows the output power of the TDANPC converter in Megawatts (MW), and the vertical axis 124 of the graph 120 shows the device loss breakdown (breakdown) of the TDANPC converter in watts. As can be seen from the graph 120, the turn-on loss of the IGBT (P _ Si _ on) and the switching loss of the SiC MOSFET (P _ SiC _ switching) are two main components of the converter output power lower than or equal to 1 MW. When the output power exceeds 1MW, the conduction loss (P _ Sic _ on) of the SiCMOSFET becomes more significant.

FIG. 10 shows a graphical diagram 130 of an efficiency curve of a 3L-TDANPC converter based on simulation results. The horizontal axis 122 of the graph 130 shows the output power of the 3L-TDANPC converter in Megawatts (MW), and the vertical axis 134 of the graph 130 shows the efficiency of the 3L-TDANPC converter in percent (%). As can be seen from the graph 130, the converter efficiency of the 3L-TDANPC converter is 99.18% at 1MW output power. Under light load conditions, such as 25% load, the efficiency can be as high as 99.5%. Such high efficiency of the 3L-TDA PC converter is mainly due to the fact that only SiC MOSFETs are used for carrier frequency operation, while SiIGBTs switch only at the fundamental frequency. It should be noted that the simulation in this case is performed at a switching frequency of 16.8 kHz. Also, the efficiency simulations here only consider Si and SiC device losses and leakage losses in the commutation loop.

It should be noted that SiC MOSFETs generally have a weak short-circuit capability compared to Si IGBTs, i.e., SiC MOSFETs cannot withstand short-circuit currents (fault currents) for as long a time as Si IGBTs. For example, if a Si IGBT can withstand 10 μ s of short circuit current, a SiC MOSFET can withstand 1 or 2 μ s of short circuit current. However, in the 3L-TDANPC converter described herein, the SiC MOSFETs, i.e., switches T5 and T6, can withstand fault currents (short circuit currents) in excess of 10 μ s without the need for use in gate drivesAny other short circuit current protection scheme. As shown in fig. 11, during a short circuit condition, the fault current flowing through switch T5 or T6 is composed of two parts: (1) i energy from buffer capacitor₁And (2) i from the DC bus capacitor₂. During short-circuit faults, i due to low loop inductance₁Is very fast, but the energy stored in the buffer capacitor is limited. For example, for the schematic 140 shown in FIG. 11, the total energy is only 2J, and thus the energy dissipated in each module is negligible. Furthermore, the high loop inductance will limit the current i₂Di/dt of (1). Furthermore, the current i₂Is limited by the saturation current level of the IGBT. For example, for a 600A IGBT, when the gate voltage is 15V, i₂Will saturate at 3000A, which is further shared by the two SiC modules. Therefore, it is safe for SiC MOSFETs to carry such low short circuit currents until the fault is cleared by desat protection in the IGBT gate driver.

Advantages of this converter topology include: (1) the small commutation loop is realized due to the addition of the buffer capacitor, so that the switching loss of the IGBT and the SiC MOSFET is low; (2) low cost because SiC MOSFETs are used only on the AC side; (3) in this topology, the heat distribution between the switching devices is balanced; and (4) short-circuit fault capability of the SiC MOSFET due to the IGBT.

This written description uses examples to explain the disclosure, including the best mode, and also to enable any person skilled in the art to practice the disclosure, including making and using any devices or systems and performing any incorporated methods. The patentable scope of the disclosure is defined by the claims, and may include other examples that occur to those skilled in the art. Such other examples are intended to be within the scope of the claims if they have structural elements that do not differ from the literal language of the claims, or if they include equivalent structural elements with insubstantial differences from the literal languages of the claims.