阅读说明：本技术 一种全桥三电平llc谐振变换器的混合控制方法 (Hybrid control method of full-bridge three-level LLC resonant converter ) 是由 耿乙文 高翔 黄汉江 郑鹏飞 周奇勋 李晓辉 陈辉 于 2019-09-10 设计创作，主要内容包括：本发明涉及一种全桥三电平LLC谐振变换器的混合控制方法,属于DC/DC技术领域。本发明控制方法包括如下步骤：(1)根据实际需要的输出电压,计算出变换器的输出电压增益；(2)当变换器的输出电压增益小于1的时候,采用PWM控制方式,通过调节开关管的占空比来实现对输出电压的调节；(3)当变换器的输出电压增益大于1的时候,采用PFM控制方式,通过调节开关管的开关频率来实现对输出电压的调节。本发明解决了传统LLC谐振变换器在单一控制模式下,空载与轻载的调压问题,以及满载情况稳定性问题。(The invention relates to a hybrid control method of a full-bridge three-level LLC resonant converter, belonging to the technical field of DC/DC. The control method comprises the following steps: (1) calculating the output voltage gain of the converter according to the actually required output voltage; (2) when the output voltage gain of the converter is smaller than 1, the output voltage is adjusted by adjusting the duty ratio of a switching tube in a PWM control mode; (3) when the output voltage gain of the converter is larger than 1, the PFM control mode is adopted, and the output voltage is adjusted by adjusting the switching frequency of the switching tube. The invention solves the problem of no-load and light-load voltage regulation and the problem of full-load stability of the traditional LLC resonant converter in a single control mode.)

1. A hybrid control method of a full-bridge three-level LLC resonant converter is characterized by comprising the following steps:

(1) calculating the output voltage gain of the converter according to the actually required output voltage;

(2) when the output voltage gain of the converter is smaller than 1, the output voltage is adjusted by adjusting the duty ratio of a switching tube in a PWM control mode;

(3) when the output voltage gain of the converter is larger than 1, the PFM control mode is adopted, and the output voltage is adjusted by adjusting the switching frequency of the switching tube.

2. The hybrid control method of a full-bridge three-level LLC resonant converter as claimed in claim 1, wherein in step (2) a PWM control mode is adopted, and the switching frequency f is_sEqual to the resonant frequency f_r。

3. The hybrid control method of a full-bridge three-level LLC resonant converter according to claim 1, wherein the duty cycle of said switching tube in step (2) is switching tube M₁And a switching tube M₃The duty cycle of (c).

4. The hybrid control method of a full-bridge three-level LLC resonant converter according to claim 1, characterized in that in step (3) a PFM control mode is adopted, the switching frequency f_sLess than the resonant frequency f_r。

5. A hybrid control method of a full-bridge three-level LLC resonant converter according to claim 4, characterized in that said resonant frequency f_rIs composed of a resonant inductor L_rAnd a resonance capacitor C_rAnd (4) determining.

Technical Field

The invention relates to a hybrid control method of a full-bridge three-level LLC resonant converter, belonging to the technical field of DC/DC.

Background

With the development of power electronics technology, three-level DC/DC (direct current/direct current) converters have gained more and more attention because they can be better used in high voltage applications. Compared with a two-level converter, a high-voltage side switching tube of the three-level converter has a lower voltage withstanding value requirement, and the voltage withstanding value of the switching tube of a preceding stage inverter circuit is only half of the input voltage. Higher frequency is a means to increase the power density of the DC/DC converter, but the increase in switching loss restricts the overall efficiency improvement. Compared with other resonant converters, an LLC (inductance-capacitance) resonant converter has the capability of implementing a primary-side switching tube soft switching technique at a wider input voltage and load. Common control modes for the LLC resonant converter include a fixed frequency control mode (PWM control) and a variable frequency control mode (PFM control). Under the control of the PFM, the efficiency of the converter is high, and very high gain can be achieved, but the no-load time voltage regulation under the control of the PFM is difficult, when the gain of the converter is large, the LLC resonant converter controlled by the PFM has certain limitation, and the over-wide frequency regulation is not beneficial to the design and optimization of magnetic elements in the converter. PWM control is simpler, the fixed frequency is beneficial to the design of magnetic elements in the converter, the voltage regulation and current limiting performance of the converter is considered, the PWM control can not be suitable for occasions with wide voltage regulation range, and the gain of the converter can be influenced to a certain extent under the PWM control. Conventional diode rectification techniques also have significant drawbacks because the diode conduction voltage drop is large, which increases the cost and difficulty of thermal design of the converter in the case of large output current.

Disclosure of Invention

The invention provides a hybrid control method of a full-bridge three-level LLC resonant converter, which takes the advantages of PFM control and PWM control into account, and solves the problems of no-load and light-load voltage regulation and the stability of full-load condition of the traditional LLC resonant converter in a single control mode.

The invention adopts the following technical scheme for solving the technical problems:

a hybrid control method of a full-bridge three-level LLC resonant converter comprises the following steps:

(1) calculating the output voltage gain of the converter according to the actually required output voltage;

(2) when the output voltage gain of the converter is smaller than 1, the output voltage is adjusted by adjusting the duty ratio of a switching tube in a PWM control mode;

(3) when the output voltage gain of the converter is larger than 1, the PFM control mode is adopted, and the output voltage is adjusted by adjusting the switching frequency of the switching tube.

The invention has the following beneficial effects:

an LLC resonance tank is added to the front-stage inversion side of the traditional three-level full-bridge converter, and the rear-stage rectification side adopts a synchronous rectification mode, so that the switching loss and the diode rectification loss caused by high frequency are reduced. PFM control is employed when the converter is in boost mode. With the converter in buck mode, PWM control is employed. Therefore, the voltage regulation of the converter in a wide range can be realized, and the problem that the voltage regulation of the converter is difficult at the no-load moment is solved. Two rectifier diodes in the rear-stage full-bridge rectifier circuit are replaced by high-efficiency N-channel MOSFETs (field effect transistors) to realize synchronous rectification, and the characteristic of low on-resistance of the MOSFETs is utilized to effectively reduce loss and improve power.

Drawings

FIG. 1 is a flow chart of a hybrid control method of the present invention.

Fig. 2 is a schematic diagram of the circuit structure of the full-bridge three-level LLC resonant converter of the present invention.

Fig. 3 is a main waveform diagram in the PWM control mode of the present invention.

FIG. 4 shows the switching mode 1 t of the present invention₀To t₁Time of day]And working mode diagram of the converter.

FIG. 5 shows the switching mode 2 t of the present invention₁To t₂Time of day]And working mode diagram of the converter.

FIG. 6 shows the switching mode 3 t of the present invention₂To t₃Time of day]And working mode diagram of the converter.

FIG. 7 shows the switching mode 4 t of the present invention₃To t₄Time of day]And working mode diagram of the converter.

FIG. 8 shows the switching mode 5t of the present invention₄To t₅Time of day]And working mode diagram of the converter.

FIG. 9 shows the switching mode 6 t of the present invention₅To t₆Time of day]And working mode diagram of the converter.

Fig. 10 is an equivalent circuit diagram of the fundamental wave of the converter of the present invention.

Fig. 11 is a voltage gain graph in the PWM control mode of the present invention.

Fig. 12 is a voltage gain graph in PFM control mode of the present invention.

The main symbol names in the above figures: v_inIs used for inputting a direct current power supply; c₁、C₂Is an input voltage-dividing capacitor; m₁～M₁₀Is a switch tube; c_oss1～C_oss10Is a parasitic capacitance; d₁～D₁₀Is a body diode; d_c1～D_c4Is a clamping diode; c_s1、C_s2Is a flying capacitor; l is_rIs a resonant inductor; c_rIs a resonant capacitor; l is_mIs an excitation inductor; t is_rA high frequency isolation transformer; d_r1、D_r2Is a rectifier diode; c_oIs an output filter capacitor; r_LIs a load; v_oIs the output voltage; v_ABIs the voltage between the two points of converters a and B.

Detailed Description

The invention will be further explained with reference to the drawings

Fig. 1 is a flow chart of the hybrid control method of the present invention, illustrating the hybrid control method of the converter for different voltage gain conditions.

Fig. 2 is a circuit topology of a full bridge three level LLC resonant converter for use in the invention. As shown in fig. 2, the full-bridge three-level LLC resonant converter is composed of a three-level inverter bridge arm circuit 1, a three-level inverter bridge arm circuit 2, a clamp circuit 3, a clamp circuit 4, a resonant tank 5, a high-frequency isolation transformer 6, a diode rectifier bridge arm circuit 7, a switching tube synchronous rectifier bridge arm circuit 8, and a filter circuit 9. Wherein the input voltage-dividing capacitor C₁And an input voltage-dividing capacitor C₂The capacity is very large and equal, and the voltage is half V of the input voltage in working_C1＝V_C2＝V_in/2. Switch tube M₁～M₄And parasitic capacitance C_oss1～C_oss4And a body diode D₁～D₄A clamping diode D_c1And D_c2Flying capacitor C_s1And a three-level inverter bridge arm circuit 1 is formed. Switch tube M₅～M₈And parasitic capacitance C_oss5～C_oss8And a body diode D₅～D₈A clamping diode D_c3And D_c4Flying capacitor C_s2And a three-level inverter bridge arm circuit 2 is formed. T is_rA resonant inductor L is added between the high-frequency isolation transformer and the full-bridge three-level inverter circuit for the high-frequency isolation transformer_rAnd a resonance capacitor C_r. High-frequency isolation transformer T_rMedium excitation inductance and resonance inductance L_rResonant capacitor C_rForming an LLC resonant tank. D_r1And D_r2To output rectifier diodes, M₉And M₁₀For outputting synchronous rectification switching tubes, C_oTo output the filter capacitance. R_LIs a load.

The control method comprises the following steps:

when the output voltage is low, the converter operates in a pulse width modulation mode (PWM) with a switching frequency f_sEqual to the resonant frequency f_r. The peak value of the voltage between the points A and B of the full-bridge three-level LLC resonant converter is V_inThree-level inverter bridge arm circuit 1Upper switch tube M₁And a switching tube M₃Has a duty ratio of alpha, and a switching tube M₁And a switch tube M₄Complementary conducting, switching tube M₂And a switching tube M₃Complementary conduction three-level inverter bridge arm circuit 2 upper switch tube M₅～M₈Duty ratio is 0.5, switching tube M₅And M₆And a switching tube M₇And M₈And conducting complementarily. Switch tube M₁And a switch tube M₄And a switching tube M₂And a switching tube M₃There is a fixed phase shift angle of 0.5T, half of the switching period_s. The output voltage is regulated by regulating the duty cycle a. Rear-stage synchronous rectification switch tube M₉And M₁₀And a switching tube M₅And M₇The working mode is the same.

At higher output voltages, the converter operates in the frequency-conversion control mode (PFM), at which the switching frequency f_sLess than the resonant frequency f_r. When the PWM mode works until the gain reaches 1, the working mode of the converter is changed from a pulse width modulation mode (PWM) to a frequency conversion modulation mode (PFM), so that the voltage gain of the converter can be larger than 1. At the moment, a leading tube switch tube M on the three-level inverter bridge arm circuit 1₁And a switching tube M₄The duty cycles are both 0.5 and both are complementarily conducting. Switching tube M as a hysteresis tube₂And a switching tube M₃The duty cycles are both also 0.5 and both are complementarily conducting. Switching tube M on three-level inverter bridge arm circuit 2₅And M₆And a switching tube M₇And M₈A switching tube M which is in 180-degree complementary conduction and has a phase lagging behind that of the bridge arm 1₂And M₃。

Prior to analysis, the following assumptions were made:

all switch tubes, diodes, high-frequency transformers, capacitors and inductors in the converter are ideal components;

② switch tube M₁～M₁₀Parasitic capacitance C of_oss1～C_oss10All volume values are C_oss；

Input voltage-dividing capacitor C₁And C₂Has a sufficiently large capacitance value, the voltage on each capacitor is sufficient during operationIs a V_in/2；

Flying capacitor C₃And C₄Is large enough that each capacitor has a voltage of V_in/2；

Output capacitance C_oSufficiently large to be approximated as a voltage V_oOf the voltage source.

The main circuit structure and the switching tube modulation diagrams of fig. 3 are shown in fig. 2, and the hybrid control method of the present invention is specifically described with reference to fig. 4 to 6:

1. switch mode 0[ t ]₀Before the moment of time]

At the moment, all the switch tubes are in an off state, and the converter is used for dividing the voltage of the voltage-dividing capacitor C at the input end₁And C₂And flying capacitor C₃And C₄Charging, voltage dividing capacitor C₁And C₂Voltage on to V_in/2 flying capacitor C₃And C₄Also rises to V_inAnd/2, the stage is a pre-charging stage of the converter.

2. Switched mode 1[ t ]₀To t₁Time of day]As shown in fig. 4

Front-stage switch tube M of converter at the moment₁、M₂、M₇、M₈In a conducting state, the rear stage rectifier diode D_r1Synchronous rectification switch tube M₁₀Is in a conducting state. The voltage at two points AB of the converter is V_inCurrent i in the exciting inductance_mAnd a resonant current i_rAnd the same, are all less than 0. In mode 1, both are gradually increased.

3. Switched mode 2[ t ]₁To t₂Time of day]As shown in fig. 5

At t₁Time of day, switch tube M₁Turn-off, switch tube M₄Conducting, switching tube M₂、M₇、M₈In a conducting state, the rear stage rectifier diode D_r1Synchronous rectification switch tube M₁₀Still in the on state. At the moment, the voltage of the two points AB of the converter falls back to V_in/2, resonant current i_rGreater than 0 and at t₁The current rises to the maximum at all times, and the current will act on the switch tube M₁Parasitic electricity ofContainer C_oss1Charging until its terminal voltage rises to V_inAt the same time, the current will be applied to the switch tube M₄Parasitic capacitance C of_oss4The energy in (2) is discharged. At t₂Time of day, resonant current i_rIs equal to the excitation current i_m。

4. Switching mode 3[ t ]₂To t₃Time of day]As shown in fig. 6

At t₂Time of day, switch tube M₂、M₇、M₈And a rear stage rectifier diode D_r1And a synchronous rectification switch tube M₁₀Still in the on state. Resonant current i_rIs equal to the excitation current i_mThen the converter enters an intermittent working state, in the working mode, the secondary side of the converter is disconnected with the primary side, and the output capacitor C_oIs a load R_LProviding energy. At t₃Time of day, rectifier diode D_r1And synchronous rectifier M₁₀Turn-off, simultaneous switching of the transistor M₂、M₇、M₈Turn off, this modality ends. Due to the switch tube M₂The turn-off current is positive, so that the current is applied to the switch tube M₂Parasitic capacitance C of_oss2Charging to V_in/2. And before the next mode, the switch tube M₃、M₅、M₆The turn-on voltage of (2) is 0, so zero voltage conduction can be achieved.

Since the operation principle of the switching modes 4 to 6 is similar to that of the switching modes 1 to 3, except that the direction of the resonant current is changed, the equivalent circuit diagrams correspond to fig. 7, 8 and 9, and thus detailed description is omitted.

The fundamental equivalent circuit of the transducer can be obtained by the fundamental component method (FHA), as shown in fig. 10.

From fig. 4 to 6, the voltage V between the two points AB and AB in the PWM control mode can be obtained_ABIs expressed by the expression V_AB(t) because of V_AB(t) is a periodic function, and thus Fourier decomposition thereof gives V_AB(t) Fourier expansion of the Fourier transform.

Wherein: v_inFor the input voltage, α is the duty cycle, T_sIs a switching cycle;

wherein: f. of_sIs the switching frequency of the converter and,

wherein: v_inAlpha is the duty cycle for the voltage on the high side of the converter.

V can be obtained by the formula_AB(t) has an amplitude of

Voltage V between two points of secondary side CD_CD(t) voltage converted to primary side is V'_CD(t), its expression is as follows, V'_CD(t) has an amplitude of 4nV_oAnd/pi. Since the switching frequency is equal to the resonance frequency, so V'_CD(t) has a magnitude equal to V_AB(t) the voltage gain of the inverter under PWM control is as follows.

Wherein: v_oThe voltage of the low-voltage side of the converter, and G is the voltage gain of the converter;

from the gain expression above, a voltage gain curve of the converter under PWM control can be derived, as shown in fig. 11.

From the fundamental equivalent circuit of fig. 10, the voltage transfer function of the resonant network can be derived as:

wherein: r_eThe actual load is converted into the input end of the rear-stage rectification part, and then is reflected to the equivalent resistance L of the primary side of the transformer according to the transformer transformation ratio_mFor exciting inductance, C_rIs a resonant capacitor, L_rIs the resonance inductance, omega is the angular frequency,

when the converter is in PFM control, the circuit has two resonant frequencies f_rAnd f_r'，f_rIs composed of a resonant inductor L_rAnd a resonance capacitor C_rIs determined from_r' is composed of a resonant inductor L_rResonant capacitor C_rAnd excitation inductance L_mAnd (4) determining. Under the PFM control mode, the voltage gain of the converter is larger than 1, and the converter is in the boost mode, wherein the switching frequency f_sWill be slightly less than the resonant frequency f_r。

The voltage gain under PFM control is expressed as:

where k is the excitation inductance L_mAnd a resonant inductor L_rK is L_m/L_r；f_nTo normalize frequency, f_n＝f_s/f_r(ii) a Q is the quality factor of the resonant circuit, Q2 pi f_rL_r/R_e。

Analysis of the voltage gain expression under PFM control results in the gain plot of fig. 12. It can be seen from the figure that the voltage gain of the converter has a wide variation range under different quality factors of the resonant circuit, and the output voltage gain can be made to be larger than 1.

As can be seen from the above description, the hybrid control method proposed by the present invention has the following advantages:

because the converter adopts PWM control in a buck mode (the voltage gain is less than 1) and adopts a PFM control hybrid control mode in a boost mode (the voltage gain is more than 1), the converter has a wide voltage output range.

The problem of the traditional LLC resonant converter in single PFM control mode, because of too wide frequency adjustment, the design of converter magnetic element is difficult is solved. The problem that the output voltage gain of the converter can only be less than or equal to 1 under the single PWM control is solved.

Because the converter adopts a three-level full-bridge structure, the voltage stress of a switching tube of the converter under the working condition of high input voltage is greatly reduced, and convenience is provided for the type selection and the cost reduction of a switching device.

Because the rear stage of the converter adopts synchronous rectification, compared with full-bridge rectification, the loss caused by forward conduction voltage drop of the diode is reduced.