阅读说明：本技术 电机控制器和用于监测去磁的方法 (Motor controller and method for monitoring demagnetization ) 是由 韩磊 高桥敏男 于 2021-03-03 设计创作，主要内容包括：公开了一种电机控制器和用于在永磁同步电机(PMSM)的工作期间监测PMSM转子永磁体的去磁的方法。该电机控制器包括：电流控制器,其被配置成生成用于驱动电机的控制信号,其中,电流控制器被配置成测量电机的电压信息和电机的电流信息；磁通量估计器,其被配置成基于电压信息和电流信息计算转子磁通链；提取电路,其被配置成从所接收的转子磁通链中提取转子磁通链的磁通链幅值；以及去磁检测器,其被配置成基于磁通链幅值在电机工作期间连续地监测电机的转子永磁体的去磁,其中,去磁检测器被配置成将磁通链幅值与预定义去磁水平进行比较,并且在磁通链幅值等于或小于预定义去磁水平的条件下检测到去磁。(A motor controller and method for monitoring demagnetization of a Permanent Magnet Synchronous Motor (PMSM) rotor during operation of the PMSM is disclosed. The motor controller includes: a current controller configured to generate a control signal for driving the motor, wherein the current controller is configured to measure voltage information of the motor and current information of the motor; a magnetic flux estimator configured to calculate a rotor flux linkage based on the voltage information and the current information; an extraction circuit configured to extract a flux linkage magnitude of the rotor flux linkage from the received rotor flux linkage; and a demagnetization detector configured to continuously monitor demagnetization of a rotor permanent magnet of the electric machine during operation of the electric machine based on the flux linkage amplitude, wherein the demagnetization detector is configured to compare the flux linkage amplitude with a predefined demagnetization level and detect demagnetization if the flux linkage amplitude is equal to or less than the predefined demagnetization level.)

1. A motor controller configured to drive a permanent magnet synchronous motor PMSM, the motor controller comprising:

a current controller configured to generate a control signal for driving the PMSM, wherein the current controller is configured to measure voltage information of the PMSM and current information of the PMSM;

a magnetic flux estimator configured to receive the voltage information and the current information and to calculate a rotor flux linkage based on the voltage information and the current information;

an extraction circuit configured to receive the rotor flux linkage and to extract a flux linkage magnitude of the rotor flux linkage from the received rotor flux linkage; and

a demagnetization detector configured to continuously monitor demagnetization of a rotor permanent magnet of the PMSM during operation of the PMSM based on the flux linkage magnitude, wherein the demagnetization detector is configured to compare the flux linkage magnitude to a predefined demagnetization level, and detect the demagnetization under a first condition in which the flux linkage magnitude is equal to or less than the predefined demagnetization level.

2. The motor controller of claim 1, wherein the magnetic flux estimator is configured to calculate the rotor flux linkage by:

calculating a two-phase rotor magnetic flux based on the voltage information and the current information, wherein the two-phase rotor magnetic flux includes a first rotor magnetic flux component and a second rotor magnetic flux component, and calculating the rotor flux linkage based on the first rotor magnetic flux component and the second rotor magnetic flux component.

3. The motor controller of claim 2, wherein the magnetic flux estimator is configured to: calculating a first back electromotive force (BEMF) of the PMSM and a second BEMF of the PMSM based on the voltage information and the current information, integrating the first BEMF by a non-ideal integral to generate the first rotor magnetic flux component, and integrating the second BEMF by the non-ideal integral to generate the second rotor magnetic flux component.

4. The motor controller of claim 2, wherein the first rotor flux component is an alpha rotor flux component of an alpha-beta two-phase quantity and the second rotor flux component is a beta rotor flux component of the alpha-beta two-phase quantity.

5. The motor controller of claim 1 wherein:

the voltage information includes a first phase alternating AC voltage component of the two-phase stator voltages and a second phase AC voltage component of the two-phase stator voltages,

the current information includes a first motor phase current component in a two-phase AC current and a second motor phase current component in the two-phase AC current.

6. The motor controller of claim 5 wherein:

the first phase AC voltage component is an alpha voltage component and the second phase AC voltage component is a beta voltage component; and is

The first motor phase current component is an alpha current component and the second motor phase current component is a beta current component.

7. The motor controller of claim 1 wherein said extraction circuit is a phase locked loop.

8. The motor controller of claim 1, wherein the rotor flux linkage is a dynamically varying sinusoidal variable containing phase and angle information of the rotor flux linkage and amplitude information representing an amplitude of the flux linkage.

9. The motor controller of claim 1, wherein the flux linkage magnitude is indicative of a permanent magnet strength of the rotor permanent magnet during PMSM operation.

10. The motor controller of claim 1, wherein based on the first condition being satisfied, the demagnetization detector is configured to: monitoring a duration of time that the flux linkage amplitude remains equal to or less than the predefined demagnetization level, comparing the monitored duration to a timing threshold, and detecting the demagnetization under a second condition that the monitored duration equals or exceeds the timing threshold.

11. The motor controller of claim 10 wherein the demagnetization detector is configured to return to monitoring for the first condition based on the second condition not being met.

12. The motor controller of claim 11, wherein the demagnetization detector is configured to: determining that the second condition is not satisfied under a third condition in which the flux linkage amplitude becomes greater than the predefined demagnetization level before the monitored duration of time satisfies the timing threshold.

13. The motor controller of claim 1 wherein said demagnetization detector is configured to generate and output a demagnetization fault signal in response to detecting said demagnetization.

14. The motor controller of claim 1, wherein the demagnetization detector is configured to store a reference rotor flux linkage, and the predefined level of demagnetization is less than the reference rotor flux linkage by a predefined amount.

15. The motor controller of claim 1 wherein:

the extraction circuit is configured to calculate a motor speed of the PMSM based on the rotor flux linkage, and

the demagnetization detector is configured to: comparing the motor speed to a minimum motor speed threshold, enabling monitoring of the demagnetization if the motor speed is equal to or greater than the minimum motor speed threshold, and disabling monitoring of the demagnetization if the motor speed is less than the minimum motor speed threshold.

16. The motor controller of claim 15, wherein the minimum motor speed threshold is less than a maximum rated speed of the PMSM by a predefined amount.

17. A method for monitoring demagnetization of rotor permanent magnets of a permanent magnet synchronous motor PMSM during operation of the PMSM, the method comprising:

generating, by a current controller, a control signal for driving the PMSM;

measuring, by the current controller, voltage information of the PMSM and current information of the PMSM;

calculating, by a magnetic flux estimator, a rotor flux linkage based on the voltage information and the current information;

extracting, by an extraction circuit, a flux linkage amplitude of the rotor flux linkage from the rotor flux linkage; and

continuously monitoring, by a demagnetization detector, the demagnetization based on the flux linkage amplitude, wherein continuously monitoring comprises comparing the flux linkage amplitude to a predefined demagnetization level, and detecting the demagnetization on a first condition that the flux linkage amplitude is equal to or less than the predefined demagnetization level.

18. The method of claim 17, wherein continuously monitoring the demagnetization further comprises:

based on the first condition being met, monitoring a duration of time that the flux linkage amplitude remains equal to or less than the predefined demagnetization level, comparing the monitored duration of time to a timing threshold, and detecting the demagnetization under a second condition that the monitored duration of time equals or exceeds the timing threshold.

19. The method of claim 18, wherein continuously monitoring the demagnetization further comprises:

returning to monitoring for the first condition based on the second condition not being satisfied.

20. The method of claim 17, further comprising:

generating, by the demagnetization detector, a demagnetization fault signal in response to detecting the demagnetization; and

outputting, by the demagnetization detector, the demagnetization fault signal.

21. The method of claim 17, further comprising:

calculating, by the extraction circuit, a motor speed of the PMSM based on the rotor flux linkage;

comparing, by the demagnetization detector, the motor speed to a minimum motor speed threshold;

enabling, by the demagnetization detector, monitoring of the demagnetization if the motor speed is equal to or greater than the minimum motor speed threshold; and

disabling, by the demagnetization detector, monitoring of the demagnetization if the motor speed is less than the minimum motor speed threshold.

Technical Field

The present disclosure relates generally to an apparatus and method for monitoring a permanent magnet flux linkage of a Permanent Magnet Synchronous Machine (PMSM).

Background

Permanent Magnet Synchronous Machines (PMSM) are increasingly used in domestic appliance applications due to their high energy efficiency and reliable operation. The motor controller controls the PMSM using a motor control algorithm. However, one of the challenges facing PMSM-type machines is the risk of demagnetization of the permanent magnet material used in the rotor. Causes of demagnetization of the permanent magnet include excessive Alternating Current (AC) applied in the stator, excessive field weakening control, temperature rise, and vibration. Demagnetization of the permanent magnets may result in reduced torque capacity per ampere, reduced motor operating efficiency, overheating, and even failure of the entire motor system.

Accordingly, it may be desirable to develop a method for continuously monitoring the permanent magnet flux linkage and detecting the occurrence of permanent magnet demagnetization throughout the life cycle of a PMSM drive system.

Disclosure of Invention

One or more embodiments provide a motor controller configured to drive a Permanent Magnet Synchronous Motor (PMSM). The motor controller includes: a current controller configured to generate a control signal for driving the PMSM, wherein the current controller is configured to measure voltage information of the PMSM and current information of the PMSM; a flux estimator configured to receive voltage information and current information and to calculate a rotor flux linkage based on the voltage information and the current information; an extraction circuit configured to receive the rotor flux linkage and to extract a flux linkage amplitude of the rotor flux linkage from the received rotor flux linkage; and a demagnetization detector configured to continuously monitor demagnetization of a rotor permanent magnet of the PMSM during operation of the PMSM based on the flux linkage magnitude, wherein the demagnetization detector is configured to compare the flux linkage magnitude with a predefined demagnetization level and detect demagnetization under a first condition in which the flux linkage magnitude is equal to or less than the predefined demagnetization level.

One or more embodiments provide a method for monitoring demagnetization of rotor permanent magnets of a Permanent Magnet Synchronous Machine (PMSM) during operation of the PMSM. The method comprises the following steps: generating, by a current controller, a control signal for driving a PMSM; measuring voltage information of the PMSM and current information of the PMSM by a current controller; calculating, by a flux estimator, a rotor flux linkage based on the voltage information and the current information; extracting, by an extraction circuit, a flux linkage amplitude of the rotor flux linkage from the rotor flux linkage; and continuously monitoring demagnetization based on the flux linkage amplitude by a demagnetization detector, wherein continuously monitoring comprises comparing the flux linkage amplitude to a predefined demagnetization level and detecting demagnetization according to a first condition that the flux linkage amplitude is equal to or less than the predefined demagnetization level.

Drawings

Embodiments are described herein with reference to the accompanying drawings.

Fig. 1A is a schematic block diagram illustrating a motor controlled actuator of a power semiconductor device according to one or more embodiments;

fig. 1B is a schematic diagram illustrating a power inverter with single-tap current sensing in accordance with one or more embodiments;

FIGS. 2A and 2B show schematic block diagrams of a motor control algorithm implemented by a motor controller in accordance with one or more embodiments;

fig. 3 is a schematic diagram of a magnetic flux estimator and a magnetic flux Phase Locked Loop (PLL) of a motor control algorithm implemented by a motor controller in accordance with one or more embodiments.

Detailed Description

In the following, details are set forth to provide a more thorough description of the exemplary embodiments. It will be apparent, however, to one skilled in the art that the embodiments may be practiced without these specific details. In other instances, well-known structures and devices are shown in block diagram form or in schematic form, rather than in detail, in order to avoid obscuring the embodiments. In addition, the features of the different embodiments described below may be combined with each other, unless specifically noted otherwise.

Further, in the following description, the same or similar elements or elements having the same or similar functions are denoted by the same or similar reference numerals. Since the same or functionally equivalent elements are given the same reference numerals in the drawings, a repetitive description of the elements provided with the same reference numerals may be omitted. Thus, the descriptions provided for elements having the same or similar reference numbers are interchangeable.

In this regard, directional terminology, such as "top," "bottom," "below," "above," "front," "back," "front," "rear," etc., may be used with reference to the orientation of the figure(s) being described. Because components of embodiments can be positioned in a number of different orientations, the directional terminology is used for purposes of illustration and is in no way limiting. It is to be understood that other embodiments may be utilized and structural or logical changes may be made without departing from the scope defined by the claims. The following detailed description is, therefore, not to be taken in a limiting sense.

It will be understood that when an element is referred to as being "connected" or "coupled" to another element, it can be directly connected or coupled to the other element or intervening elements may be present. In contrast, when an element is referred to as being "directly connected" or "directly coupled" to another element, there are no intervening elements present. Other words used to describe the relationship between elements should be interpreted in a similar manner (e.g., "between … …" and "directly between … …", "adjacent" and "directly adjacent", etc.).

In the embodiments described herein or shown in the drawings, any direct electrical connection or coupling, i.e. any connection or coupling without further intermediate elements, may also be achieved by an indirect connection or coupling, i.e. a connection or coupling with one or more further intermediate elements, and vice versa, as long as the general purpose of the connection or coupling is substantially maintained, e.g. to send a certain signal or to send a certain information. Features from different embodiments may be combined to form further embodiments. For example, unless stated to the contrary, variations or modifications described with respect to one of the embodiments may also be applicable to the other embodiments.

The term "substantially" may be used herein to explain small manufacturing tolerances (e.g., within 5%) that are considered acceptable in the industry without departing from aspects of the embodiments described herein.

A sensor may refer to a component that converts a physical quantity to be measured into an electrical signal, such as a current signal or a voltage signal. For example, the physical quantity may be a current or a voltage at the shunt resistor in the single shunt resistor system.

The signal processing circuitry and/or signal conditioning circuitry may receive one or more signals from one or more components and perform signal conditioning or signal processing on the one or more signals. As used herein, signal conditioning refers to manipulating a signal in the following manner: so that the signal meets the requirements of the next stage for further processing. Signal conditioning may include conversion from analog to digital (e.g., via an analog-to-digital converter), amplification, filtering, conversion, biasing, range matching, isolation, and any other processing required to adapt the signal for processing after conditioning.

Thus, the signal processing circuit may comprise an analog-to-digital converter (ADC) to convert analog signals from the one or more sensor elements into digital signals. The signal processing circuit may also include a Digital Signal Processor (DSP) that performs some processing on the digital signal.

Many functions of modern devices in automotive, consumer and industrial applications, such as converting electrical energy and driving electric motors or motors, rely on power semiconductor devices. For example, Insulated Gate Bipolar Transistors (IGBTs), Metal Oxide Semiconductor Field Effect Transistors (MOSFETs), and diodes have been used in a variety of applications including, but not limited to, power supplies and switches in power converters.

Power semiconductor devices typically include a semiconductor structure configured to conduct a load current along a load current path between two load terminal structures of the device. In addition, the load current path may be controlled by means of a control electrode, sometimes referred to as a gate electrode. For example, the control electrode may set the power semiconductor device to one of a conducting state and a blocking state upon receiving a corresponding control signal from, for example, a driver unit. The control signal may be realized by a voltage signal or a current signal having a controlled value.

Power transistors are power semiconductor devices that may be used to drive a load current. For example, an IGBT is turned "on" or "off" by activating and deactivating its gate terminal. Applying a positive input voltage signal across the gate and emitter will cause it to be in an "on" state, while making the input gate signal zero or slightly negative will cause it to become "off. There are on-processes and off-processes for turning the power transistor on and off. During the turn-on process, a gate driver Integrated Circuit (IC) may be used to provide (pull) a gate current (i.e., a turn-on current) to the gate of the power transistor to charge the gate to a sufficient voltage to turn on the device. Conversely, during turn-off, the gate driver IC is used to draw (sink) gate current (i.e., turn-off current) from the gate of the power transistor to discharge the gate sufficiently to turn off the device. The current pulses may be output from the gate driver IC as the control signal according to a Pulse Width Modulation (PWM) scheme. Thus, during a PWM period for controlling the power transistor, the control signal may be switched between an on current level and an off current level. This in turn charges and discharges the gate voltage to turn the power transistor on and off, respectively.

In particular, the gate of the power transistor is a capacitive load and, upon initiating a switching event, designates an on current (i.e., a gate source current) and an off current (i.e., a gate sink current) as initial currents. During the turn-off event, after a small amount of time (small compared to the PWM period), the gate current decreases and reaches a zero value when the gate reaches 0V. After a short time (small compared to the PWM period) during the on-conduction event, the gate current decreases and reaches a zero value when the gate reaches 15V.

The transistors may include Insulated Gate Bipolar Transistors (IGBTs) and Metal Oxide Semiconductor Field Effect Transistors (MOSFETs) (e.g., silicon MOSFETs or silicon carbide MOSFETs). Although an IGBT may be used as an example in the following embodiments, it should be understood that a MOSFET may replace an IGBT, and conversely an IGBT may replace a MOSFET. In this case, in any of the examples described herein, when the IGBT is replaced with the MOSFET, the collector of the IGBT may be replaced with the drain of the MOSFET, the emitter of the IGBT may be replaced with the source of the MOSFET, and the drain-source voltage V of the MOSFET_DSCollector-emitter voltage V that can replace IGBT_CE. Thus, any IGBT module can be replaced by a MOSFET module, whereas any MOSFET module can be replaced by an IGBT module.

The specific embodiments described in this specification pertain to, but are not limited to, power semiconductor devices that may be used within a power converter or power supply. Thus, in an embodiment, the power semiconductor device may be configured to carry a load current to be provided to the load and/or by the power supply, respectively. For example, the semiconductor device may include one or more power semiconductor cells such as monolithically integrated diode cells and/or monolithically integrated transistor cells. Such a diode cell and/or such a transistor cell may be integrated in the power semiconductor module.

Power semiconductor devices comprising transistors suitably connected to form a half bridge are commonly used in the field of power electronics. For example, a half-bridge may be used to drive a motor or a switched mode power supply.

For example, a multi-phase inverter is configured to provide multi-phase power by providing a multi-phase load (e.g., a three-phase motor). For example, three-phase power includes three symmetrical sine waves that are 120 electrical degrees out of phase with each other. In a symmetrical three-phase power supply system, the three conductors each carry an Alternating Current (AC) having the same frequency and voltage amplitude but a phase difference of one-third of the period with respect to a common reference. Due to the phase difference, the voltage on any conductor reaches its peak at one-third of the period after one of the other conductors and one-third of the period before the remaining conductors. The phase delay provides constant power transfer for a balanced linear load. This also makes it possible to generate a rotating magnetic field in the motor.

In a three-phase system feeding balanced and linear loads, the sum of the instantaneous currents of the three conductors is zero. In other words, the current in each conductor is equal in magnitude but opposite in sign to the sum of the currents in the other two conductors. The return path for the current in any phase conductor is the other two phase conductors. The instantaneous current produces a current space vector.

The three-phase inverter includes three inverter legs, one for each of the three phases, and each of the inverter legs is connected in parallel with each other to a Direct Current (DC) voltage source. Each inverter leg includes a pair of transistors arranged, for example, in a half-bridge configuration for converting DC to AC. In other words, each inverter leg includes two complementary transistors (i.e., a high-side transistor and a low-side transistor) connected in series, and they are turned on and off complementarily to each other for driving the phase load. However, a multi-phase inverter is not limited to three phases and may include two or more than three phases, with each phase having inverter legs.

Fig. 1A is a schematic block diagram illustrating a motor controlled actuator 100 of a power semiconductor device according to one or more embodiments. In particular, the motor-controlled actuator 100 includes a power inverter 1 and an inverter control unit 2. The inverter control unit 2 functions as a motor control unit, and thus may also be referred to as a motor controller or a motor control IC. The motor control unit may be a single IC or may be split into a microcontroller and gate drivers on two or more ICs.

The motor controlled actuator 100 is also coupled to a three-phase motor M that includes three phases U, V and W. The power inverter 1 is a three-phase voltage generator configured to supply three-phase power by supplying three-phase voltage to drive the motor M. It should also be understood that power inverter 1 and inverter control unit 2 may be placed on the same circuit board, or on separate circuit boards.

Deviations in both amplitude and phase may result in power and torque losses in the motor M. Thus, the motor controlled actuator 100 may be configured to monitor and control the magnitude and phase of the voltage provided to the motor M in real time to ensure that proper current balance is maintained based on a feedback control loop. Open loop motor control units also exist and can be implemented.

The power inverter 1 for a three-phase motor M comprises a switch array of six transistor modules 3u +, 3u-, 3v +, 3v-, 3w +, and 3w- (collectively referred to as transistor modules 3) arranged in complementary pairs. Each complementary pair constitutes an inverter branch supplying phase currents to the three-phase motor M. Each inverter leg therefore comprises an upper (high-side) transistor module 3 and a lower (low-side) transistor module 3. Each transistor module may include a power transistor and may also include a diode (not shown). Thus, each inverter leg includes an upper transistor and a lower transistor. Load current paths U, V and W extend from the output of each inverter leg (i.e., the output of each half bridge) between the complementary transistors and are configured to be coupled to a load, such as motor M. The power inverter 1 is coupled to a DC power source 4 (e.g., a battery or a diode bridge rectifier) and an inverter control unit 2.

In this example, the inverter control unit 2 includes a motor control circuit and a gate driver circuit for controlling the switch array. In some examples, the inverter control unit 2 may be monolithic, with the motor control circuitry and the gate driver circuitry integrated onto a single die. In other examples, the motor control circuit and the gate driver circuit may be divided into separate ICs. A "monolithic" gate driver is a gate driver on a single silicon chip and can further be made by a specific High Voltage (HV) technology. In addition, the gate driver IC may be integrated on the power inverter 1.

The motor controller performs a motor control function of the motor controlled actuator 100 in real time and transmits a PWM control signal to the gate driver. The motor control function may include controlling a permanent magnet motor or an induction motor, and may be configured as sensor-based control or sensorless control that does not require rotor position sensing, as is the case with sensor-based control having hall sensors and/or encoder devices. Alternatively, the motor control function may include a combination of both sensor-based control (e.g., used at lower rotor speeds) and sensorless control (e.g., used at higher rotor speeds).

For example, the inverter control unit 2 includes a controller and driver unit 5, and the controller and driver unit 5 includes a microcontroller unit (MCU)6 as a motor controller and a gate driver 7 for generating a driver signal for controlling the transistor of each transistor module 3. Thus, the load current paths U, V and W may be controlled by the controller and driver unit 5 by controlling the control electrode (i.e., gate electrode) of the transistor 3. For example, upon receiving the PWM control signal from the microcontroller, the gate driver IC may set the respective transistor to one of a conducting state (i.e., an on state) and a blocking state (i.e., an off state).

The gate driver IC may be configured to receive instructions from the microcontroller including power transistor control signals and to switch the respective transistor 3 on or off in accordance with the received instructions and control signals. For example, during the turn-on process of the respective transistor 3, the gate driver IC may be used to provide (pull) a gate current to the gate of the respective transistor 3 in order to charge the gate. Conversely, during the turn-off process, the gate driver IC may be used to draw (sink) the gate current from the gate of the transistor 3 in order to discharge the gate.

The inverter control unit 2 or the controller and drive unit 5 itself may include a PWM controller, ADC, DSP and/or clock source (i.e., timer or counter) for implementing a PWM scheme for controlling the state of each transistor and ultimately each phase current provided on the respective load current paths U, V and W.

In particular, the microcontroller of the controller and driver unit 5 may use a motor control algorithm, such as a Field Oriented Control (FOC) algorithm, for providing current control in real time for each phase current output to a multi-phase load, such as a multi-phase motor. Thus, the control loop of the magnetic field orientation may be referred to as a current control loop. The motor speed can also be controlled by adding an acceleration constant control loop over the FOC control providing speed constant control. Thus, the FOC (i.e., the current control loop) may be considered an inner control loop, and the constant speed control loop may be considered an outer control loop. In addition, the motor power and hence the motor speed may also be controlled by a power constant control loop above the speed constant control loop. Thus, the power constant control loop may be considered as the outermost control loop, at least with respect to the current control loop and the speed constant control loop. In other words, the current control loop may be considered an inner control loop, the speed constant control loop may be considered an intermediate control loop, and the power constant control loop may be considered an outer control loop.

The current control loop and the constant speed control loop are always kept active or enabled during motor control (i.e., during operation of the motor). Likewise, the constant power control loop may remain active or enabled during motor control. However, the power-constant control loop may also be switchably activated/deactivated (enabled/disabled) during motor control. In case the power constant control loop is activated, the controller and driver unit 5 is considered to be in the power constant control mode even if the speed constant control loop is also activated. In case the power constant control loop is deactivated, the controller and driver unit 5 is considered to be in the speed constant control mode.

In some cases, a fourth control loop (e.g., a position control loop), also outside of the constant speed control loop, may be used to control motor position.

For example, during an FOC, motor phase currents should be measured so that accurate rotor position can be determined in real time. To enable determination of motor phase currents, MCU 6 may employ an algorithm (e.g., Space Vector Modulation (SVM), also known as Space Vector Pulse Width Modulation (SVPWM)) using single-tap current sensing.

Furthermore, the switches 3 (i.e. transistors) of the power inverter 1 are controlled such that at no time are both switches in the same inverter leg rendered conductive, otherwise the DC power supply will be short-circuited. This requirement can be met by complementary operation of the switches 3 within the inverter legs, according to the motor control algorithm.

Fig. 1B is a schematic diagram illustrating a power inverter 1 utilizing single-tap current sensing in accordance with one or more embodiments. Specifically, the power inverter 1 includes a shunt resistor Rs placed on the negative DC link of the power inverter 1. Transistors 3u +, 3u-, 3v +, 3v-, 3w +, and 3 w-are shown as switches, and motor M is shown with a winding for each of its phases. Here, UO denotes the line to neutral voltage from the bridge center point U to the motor neutral point O; UN represents the U-bridge voltage from the bridge midpoint U to the negative bus supply rail N; UV represents the line-to-line voltage from U-phase to V-phase; VW represents the line-to-line voltage from V phase to W phase; and WV denotes a line-to-line voltage from the W phase to the V phase.

The MCU 6 in fig. 1A may receive samples of the current drawn from the shunt resistor Rs and then use an algorithm (i.e., software) to reconstruct the three-phase current in real-time. For example, SVPWM is a vector control based algorithm that requires sensing of three motor phase currents. The DC link current pulses are sampled at precise time intervals by using a single shunt resistor Rs. The voltage drop across the shunt resistor Rs may be amplified by an operational amplifier within the inverter control unit 2 and shifted up by 1.65V, for example. The resulting voltage may be converted by an ADC within the inverter control unit 2. Based on the actual combination of the switches, the SVPWM algorithm is used to reconstruct the three-phase current of the motor M. The ADC may measure the DC link current during the active vector of the PWM period. In each sector, two-phase current measurements are available. Since the sum of the three winding currents is zero, a third phase current value can be calculated.

SVPWM itself is an algorithm for controlling PWM in real time. The algorithm is used to generate an AC waveform and may be used to drive a three-phase AC powered motor at variable speeds from a DC source using a plurality of switching transistors. Although the examples herein are described in the context of a three-phase motor, the examples are not limited thereto and may be applied to any load scheme.

Further, it should be understood that other implementations besides a single shunt resistor may be used for current sensing, and other motor control algorithms may be used to control the load, and the embodiments described herein are not limited thereto.

Fig. 2A and 2B show schematic block diagrams of a motor control algorithm 200 according to one or more embodiments. In particular, fig. 2 consists of two parts, fig. 2A and 2B, which are joined at respective boundaries (a) to form a complete motor control algorithm 200. The motor control algorithm 200 may be implemented as firmware programmed into the motor controller 6 or by a combination of firmware and circuit components. The motor controller 6 itself may include one or more controllers, one or more processing circuits, and/or one or more signal processors configured to implement the motor control algorithm 200.

In particular, the motor control algorithm 200 includes a constant speed control loop 12, which may be sensor-based or sensorless-based, and a current control loop 13 implemented by the motor controller 6 shown in fig. 1A. Thus, the motor controller 6 includes a speed controller that implements outer loop control and a current controller that implements inner loop control. The constant speed control loop 12 may be used interchangeably with the speed controller 12. Similarly, the current control loop 13 may be used interchangeably with the current controller 13.

As used herein, Vq and Vd denote the stator Q-axis voltage and D-axis voltage, respectively, of the motor in the DQ coordinate system. That is, Vq is a motor voltage component on the Q-axis of the DQ coordinate system, and Vd is a motor voltage component on the D-axis of the DQ coordinate system. Likewise, Iq and Id represent stator Q-axis current and D-axis current of the motor in the DQ coordinate system, respectively. That is, Iq is a motor current component on the Q-axis of the DQ coordinate system, and Id is a motor current component on the D-axis of the DQ coordinate system. In addition, each proportional-integral (PI) controller receives a proportional gain KP and an integral gain KI and generates an output according to equations 1 and 2:

PI output KP Δ + KI ═ Δ dt equation 1

Where Δ is the error or deviation of the actual measured value (PV) from the Setpoint (SP).

Δ ═ SP-PV equation 2

The FOC software supports driving two types of Permanent Magnet Synchronous Machines (PMSM), namely, a constant air gap surface mount magneto and an interior mount magneto with variable reluctance. The sensorless FOC algorithm structure is shown in fig. 2A and 2B, and follows a cascaded control structure with an outer speed constant control loop and an inner current control loop, each respectively functioning in varying the motor winding voltage to drive the motor at a target power or target speed. As long as the magnetic flux estimator and PLL unit 43 is active, a sensor based FOC algorithm structure may also be used. The speed constant control loop receives the target speed TargetSpeed from, for example, an external signal generator 15 (e.g., universal asynchronous receiver/transmitter (UART), Variable Speed Pump (VSP), frequency and duty cycle). For example, the external signal generator 15 may be configured to generate an external digital or analog signal 15s based on one or more input parameters intended to set a target speed of the motor, and provide the target speed of the motor, TargetSpeed, to the speed slope spdrampwate block 21 of the speed controller 12. It should be appreciated that the target speed, TargetSpeed, may be provided by another source, such as another controller loop that functions as an additional outer control loop, making the speed control loop an intermediate control loop.

The speed controller 12 calculates a motor torque required to follow a target speed (TargetSpeed). TargetSpeed is a variable that sets the target speed of the motor. The target speed is a constant value at the time of setting; the speed slope SpdRampRate block 21 changes it to a ramp-up value SpdRef according to the speed slope. The error generator 22 receives the SpdRef signal and the actual (measured) motor speed value MotorSpeed (i.e., estimated speed) from the magnetic flux estimator and Phase Locked Loop (PLL) unit 43 and generates a speed error ErrSpeed, which is the deviation between the SpdRef signal and the actual (estimated) motor speed.

The PI compensator 23 acts on the error ErrSpeed. The integral term forces the velocity steady state error to zero, while the proportional term improves the high frequency response. The PI compensator gains KP and KI are adjusted depending on the motor characteristics and the load characteristics to meet the target dynamic performance. The output of the PI compensator 23 is a torque current TrqRef capable of maintaining the motor speed SpRef. The limit function block 24 applies one or more limit functions to the output of the PI compensator 23. For example, the limit function block 24 performs a motor limit function MotorLim on the output of the PI compensator 23 to prevent integral saturation (integral windup) and to keep the motor current within the motor maximum current. The limit function block 24 performs a low speed limit function LowSpeedLim on the output of the PI compensator 23 to limit the motor current at low speed. The limit function block 24 executes a regenerative current limit function RegenLim on the output of the PI compensator 23 to limit the regenerative current of the motor.

The current loop of the current controller 13 drives the motor current required to generate this torque current TrqRef. An Interior Permanent Magnet (IPM) controller 31 is configured to split the torque current TrqRef into IdRef and IqRef for an interior mounted magneto having variable reluctance based on the difference between Ld and Lq. For a constant air gap Surface Mount Magnetic (SMM) motor, IqRef equals TrqRef, and IdRef equals 0. IqRef is the current command on the Q-axis (i.e., the reference current value). In other words, IqRef is a target current value of the Iq current component. Similarly, IdRef is the value of the target current (i.e., the reference current value) of the Id current component. The IPM controller 31 also receives a field weakening current (IdFwk) which is limited by the limiting function block 45 based on FwkCurRatio. The field weakening module 44 calculates a weak magnetic current (flux weakening current) IdFwk based on Vdq (i.e., the square root of Vd and Vq) and FwkVoltLvl, which sets the field weakening level. The field weakening current IdFwk is added to IdRef in IPM controller 31 for all Interior Permanent Magnet Synchronous Machines (IPMSM) and Surface Permanent Magnet Synchronous Machines (SPMSM).

The current Iq loop PI compensator 34, also referred to as Iq controller 34, acts on the error ErrIq between IqRef and Iq. The integral term forces the steady state error to zero, while the proportional term improves the high frequency response. The PI compensator gains KP and KI are adjusted depending on the motor characteristics and the load characteristics to meet the target dynamic performance. The limit function block 36 applies one or more limit functions to the output of the PI compensator 34 to prevent integral saturation and maintain the inverter output voltage based on the VdqLim.

Similarly, the current Id loop PI compensator 35, also referred to as Id controller 35, acts on the error ErrId between IdRef and Id. The PI compensator gains KP and KI are also adjusted depending on motor characteristics and load characteristics to meet the target dynamic performance, but they are typically the same as the current Iq loop PI compensator 34. The limit function block 37 applies one or more limit functions to the output of the PI compensator 35 to prevent integral saturation and maintain the inverter output voltage based on VdqLim.

The forward vector rotation unit 38 applies forward vector rotation to the current loop output voltages Vd and Vq, and converts the current loop output voltages Vd and Vq into two-phase AC voltage components V α and V β based on the rotor angle θ calculated by the magnetic flux estimator and PLL unit 43. The space vector pulse width modulator 39 receives the two-phase alternating-current voltage components V α and V β, and generates inverter switching signals (i.e., six-way PWM control signals output from the motor controller 6) based on the V α and V β voltage inputs and SVPWM. Then, the gate driver 7 turns on/off the corresponding power transistor 3 based on the PWM control signal. Pwmfeq provides the frequency of the PWM control signal to space vector pulse width modulator 39. In the event of a FAULT being detected, FAULT provides a FAULT signal to space vector pulse width modulator 39.

The current loop of the current controller 13 calculates the inverter voltage to drive the motor current required to generate the desired torque. The phase current reconstruction circuit 40 reconstructs each of the phase currents Iu, Iv and Iw of the respective phases U, V and W using single-tap reconstruction. In particular, the phase current reconstruction circuit 40 measures the DC link current in the shunt resistor during the active vector of the PWM period. In each PWM period, there are two different active vectors, and the DC link current in each active vector represents the current on one motor phase. Since the sum of all three winding currents is zero in equilibrium conditions, it is possible to calculate the third phase current value.

Magnetic Field Orientation Control (FOC) uses Clarke transformation at Clarke transformation unit 41 to apply an alpha-beta (α - β) transformation to the three-phase currents to derive an α current I α and a β current I β. The FOC also uses vector rotation (i.e., cordic rotation) at the vector rotation unit 42 to transform the motor winding current with the alpha and beta currents I α and I β into two quasi-DC current components, i.e., an Id current component that boosts or attenuates the rotor field and an Iq current component that produces the motor torque.

Two error generators (e.g., subtractors) 32 and 33 generate error values ErrIq and ErrId, respectively. Specifically, the error generator 32 receives the reference current value IqRef as a Set Point (SP) value from the IPM control block 31 and the Iq current value as an actual measurement value (PV) from the vector rotation unit 42, and generates an error value ErrIq. Similarly, the error generator 33 receives the reference current value IdRef (i.e., the reference current value on the D-axis) as the Set Point (SP) value from the IPM control block 31 and the Id current value as the actual measurement value (PV) from the vector rotation unit 42, and generates the error value ErrId.

Generally, the IPM control block 31 separates the torque reference current TrqRef from the speed controller into Iqref and Idref according to the difference between the motor inductances Ld, Lq. Iqref and Idref represent target currents. Typically, IdRef is zero for SMM motors or scaled to the negative value of the torque current TrqRef for IPM motors. However, above a certain speed (referred to as the base speed), the inverter output voltage becomes limited by the DC bus voltage. In this case, field-weakening controller 44 generates a negative Id and decouples that Id from the torque reference current to oppose the rotor field that reduces the back-wound electromotive force (EMF). This enables operation at higher speeds but with lower torque output. The field-weakening controller 44 is used to adjust the Id current to keep the motor voltage magnitude within the bus voltage limit.

The rotor magnet position estimator includes a magnetic flux estimator and a PLL 43. The magnetic flux estimator and the magnetic flux PLL operate to detect the rotor position and to measure the motor speed of the operating motor. As provided in equations 3 and 4 below, the magnetic flux is calculated based on the feedback currents (i.e., using the alpha and beta currents I α and I β), the estimated voltages V α and V β (based on the DC bus feedback voltage and modulation index), and the motor parameters (inductance and resistance). The output of the flux estimator 51 represents the rotor flux in the alpha-beta (fixed quadrature frame, u phase aligned with alpha) two-phase quantities.

An angle and frequency Phase Locked Loop (PLL)55 of the flux estimator and PLL 43 estimates the flux angle (i.e., the estimated rotor angle θ) and the motor speed from the rotor flux vector in the α - β component. The vector rotation 53 of the PLL calculates the error between the rotor flux angle and the estimated angle. The PI compensator of the PLL 55 in the closed loop path forces the angle estimate and the frequency estimate to track the angle and frequency of the rotor flux. The motor speed is derived from the rotor frequency based on the number of rotor poles.

When driving an Interior Permanent Magnet (IPM) machine, rotor salient poles (saliency) can generate a reluctance torque component to increase the torque generated by the rotor magnets. When driving a surface magneto (SMM), the saliency is zero (Ld ═ Lq) and Id is set to zero to achieve maximum efficiency. In the case of an IPM machine with salient poles (Ld < Lq), negative Id will produce positive reluctance torque. The most efficient operating point is to maximize the total torque at a given current magnitude. The most efficient operating points for both surface magneto (SMM) and Interior Permanent Magnet (IPM) machines are calculated by IPM control block 31.

It should be understood that the illustrated speed controller 12 and current controller 13 illustrate only one example configuration and are not limited thereto. For example, typically, the speed controller 12 is configured with a speed control loop that outputs a torque current TrqRef based on the target speed TargetSpeed. In addition, the current controller 13 is configured to calculate voltage and current information for driving the motor based on the torque current TrqRef output from the speed controller 12. Specifically, the current controller 13 determines a stator Q-axis voltage Vq and a D-axis voltage Vd, and a stator Q-axis current Iq and a D-axis current Id.

In the catch spin method, the controller may track the back emf to determine if the motor is rotating and if so, in that direction. The conventional trapping spin sequence starts after the charging phase of the bootstrap capacitor is completed. During spin capture, both IqRef and IdRef are set to 0 (speed regulator disabled) while the flux PLL attempts to lock onto actual motor speed (MotorSpeed) and rotor angle (rotoangle). The capture spin time is defined by the TCatchSpin parameter. Once the capture spin time has elapsed, the calculated motor speed is checked using the "DirectStartThr" parameter value. If the motor speed is greater than or equal to the "DirectStartThr" parameter value, normal speed control is initiated and the current motor speed will become the initial speed reference and also be set as the speed ramp starting point. Depending on the set target speed, the electric machine will either slow down (via regenerative braking) or accelerate to reach the desired speed. If the motor speed is less than the DirectStartThr parameter value, the motor state changes to an Angle sensing (ANGLESENSING) state.

During the forward capture spin sequence, where the motor spins in the same desired direction, no motor current is injected. After the capture spin time TCatchSpin has elapsed, assuming that the flux PLL of block 43 locks to the actual motor speed, the motor speed in this case will become the starting point for the initial speed reference and the speed ramp reference SpdRef used by the speed slope SpdRampRate block 21. The motor continues to accelerate or decelerate following the speed ramp reference SpdRef to reach the set target speed TargetSpeed.

During the reverse capture spin sequence, in which the motor spins in the desired opposite direction, no motor current is injected. After the TCatchSpin time has elapsed, the motor is still spinning in the opposite direction at a speed above the regeneration speed threshold (RegensdThr). Thus, the injected torque, limited by the value defined in the RegenLim parameter, forces the electric machine to decelerate via regenerative braking. Once the speed of the reverse spinning motor falls below the regeneration speed threshold (regenstdthr), the injection torque is limited by MotorLim (RegenLim ≦ MotorLim). The injected torque forces the motor to stop and begin accelerating in the desired spin direction to follow the speed ramp reference SpdRef to reach the set target speed TargetSpeed.

The described embodiment uses a magnetic flux estimator 51 and a magnetic flux magnitude and angle extraction block 55, which may be implemented, for example, by a PLL. The flux estimator and PLL unit 43 continuously monitors the permanent magnet flux linkage during the entire life cycle of the PMSM drive system in order to detect the occurrence of demagnetization of the permanent magnets of the motor rotor.

Fig. 3 is a schematic diagram of a magnetic flux estimator, a magnetic flux PLL and a demagnetization detector in a magnetic flux estimator and PLL unit 43 according to one or more embodiments. Rotor flux linkageThe magnetic flux amplitude and angle extraction is performed subsequently by the PLL 35 via the magnetic flux estimator 51 estimation.

The magnetic flux estimator 51 is configured to receive the two-phase AC voltage components V α and V β from the forward vector rotation unit 38 and the α current I α and β current I β from the Clarke transformation unit 41 to perform non-ideal integration of the motor back EMF. In particular, the magnetic flux estimator 51 obtains the α - β 2 phasor (fixed quadrature frame) by non-ideal integration of the motor back EMF) The non-ideal integral of the motor back EMF is determined by using the corresponding motor phase current I α or I β, the corresponding stator voltage V α or V β, the motor winding resistance R as shown in equations 3 and 4 below_sAnd motor winding inductance L_sTo calculate:

wherein the content of the first and second substances,represents the alpha rotor flux, andrepresenting the beta rotor flux.

Rotor fluxAndare provided to a vector rotator 53, the vector rotator 53 being configured to generate D and Q components in a rotating frame of a DQ coordinate system. The D rotor flux component represents the amplitude of the rotor flux. The Q rotor flux component represents the error between the rotor flux angle alpha and the estimated angle theta.

Rotor flux linkageDerived by vector rotator 53 according to equation 5 below:

specifically, the vector rotator performed by vector rotator 53 calculates as follows:

as a result of this, it is possible to,and is

Therefore, the following is true;

when α ═ θ, the following is true:

as mentioned above, the D rotor flux component (i.e., D in the vector rotator calculation) represents the amplitude of the rotor fluxThe Q rotor flux component (i.e., Q in the vector rotator calculation) represents the error between the rotor flux angle α and the estimated rotor angle θ。

Alternatively, the rotor flux linkageMay be calculated by the magnetic flux estimator 51 by the application 5, or the vector rotator 53 may be incorporated by the magnetic flux estimator 51.

Rotor flux linkageIs a dynamically changing sinusoidal variable containing phase and angle information ω t of the rotor flux linkage and amplitude information of the rotor flux linkageE.g. from rotor flux linkageThe angle and amplitude of the rotor flux linkage can be extracted by the PLL 55. The extracted (i.e. estimated) flux linkage amplitude represents the permanent magnet strength at a given time of motor operation, since its self-inductance (Ls) -related flux component is excluded. Thus, as the motor flux strength changes with age, its magnitude also changes (i.e., decreases) proportionally without being affected by self-inductance, which changes reflect the degradation. Thus, the system can identify and detect the amount of demagnetization of the permanent magnet machine by continuously monitoring the estimated flux linkage amplitude.

In particular, flux linkage amplitude is provided from PLL 55 to demagnetization detector 57To make an estimate. The demagnetization detector 57 is further configured to receive and store a reference magnetic flux magnitudeAnd amplitude of flux linkageBased on the reference magnetic flux amplitudeThe predefined demagnetization level DemagThresh. In particular, the predefined demagnetization level DemagThresh is the reference magnetic flux amplitude according to equation 6 belowPredefined percentage (Demag Level%) or fraction:

the predefined demagnetization level DemagThresh is determined to be smaller than the reference flux amplitude(e.g., at least 10% less) and representing a threshold value, the magnitude of the flux linkage to be measuredCompared to the threshold for demagnetization monitoring and detection. Although Demag Level% is not limited to 90% or less, it should be used from a reference magnetic flux magnitudeTo allow for small variations in the measured flux linkage amplitude and to safely prevent detection of false demagnetization.

The demagnetization detector 57 may be, for example, at flux linkage amplitudeAbove a predefined demagnetization level DemagThresh, it is determined that no demagnetization fault exists. In this case, the demagnetization detector 57 does not generate a failure signal or generates an OK signal.

On the other hand, demagnetization detector 57 can be at flux linkage magnitudeIt is determined that a demagnetization fault exists on condition that it is equal to or less than a predefined demagnetization level DemagThresh. In this case, the demagnetization detector 57 may generate a demagnetization fault signal indicating that demagnetization of the rotor permanent magnet has occurred.

In addition, timing thresholds may also be considered in demagnetization monitoring and detection. For example, the demagnetization detector 57 may not only couple flux linkage magnitudeCompared to a predefined demagnetization level DemagThresh, and the demagnetization detector 57 can also monitor flux linkage amplitude via a counter or other meansAn amount of time equal to or less than a predefined demagnetization level DemagThresh, and the measured time may be compared to a predefined timing threshold. Demagnetization of the rotor permanent magnets may be detected on condition that the measured time equals or exceeds a predefined timing threshold. The timing condition may also help to tolerate variations in the measured flux linkage amplitude for a small duration without triggering a demagnetization fault signal. Therefore, erroneous demagnetization detection can be prevented.

As a result, demagnetization detector 57 can apply a dual condition analysis, where it monitors the first condition to be met (i.e., flux linkage amplitude)Equal to or less than a predefined demagnetization level DemagThresh) and, once the first condition is satisfied, the demagnetization detector 57 monitors the second condition to be satisfied (i.e. the flux linkage amplitude value)A duration equal to or less than a predefined demagnetization level DemagThresh meets or exceeds a predefined timing threshold). If the second condition is met, the demagnetization detector 57 generates a demagnetization fault signal. On the other hand, if the monitoring of the second condition is before the predetermined timing threshold is metAmplitude of flux linkage during measurementBeyond the predefined demagnetization level DemagThresh, the demagnetization detector 57 does not generate a demagnetization fault signal and is reset back to monitoring for the first condition.

As a result, the rotor flux linkage is continuously estimated via the flux estimator 51 and the vector rotator 53 during operation of the electric machine M (i.e., during rotation) and throughout the life cycle of the PMSM drive systemFurthermore, the demagnetization detector 57 is configured to continuously monitor the occurrence of demagnetization of the rotor permanent magnets during operation of the motor M and throughout the life cycle of the PMSM drive system.

In addition, the demagnetization detector 57 includes a storage device 58, such as a flight recorder, the storage device 58 being configured to periodically store the estimated flux linkage amplitude when the magnetic flux estimator 51 and the magnetic flux PLL 55 are locked and in a steady state

Initially, the flux linkage amplitude may be estimated at zero hours of service timeDuring which the motor M operates under a predefined load and the flux linkage amplitudeIs measured. In other words, a zero hour service time is a test period during which the motor is tested and its characteristics may be measured to develop a system model, for example, for one or more aspects of a motor control algorithm. Initial flux linkage amplitude determined during the test periodCan be stored inIn the storage device 58 and is set to the reference flux linkage amplitudeAnd thus used in demagnetization detection during the remaining life of the motor control system in the manner described above.

If the actual flux linkage amplitude measured during the operating phase of the machine M is presentDetected as being below the predefined demagnetization level for a configurable amount of time, a demagnetization fault is triggered by the demagnetization detector 57. One advantage of this concept is that the rotor flux linkage estimation does not require any additional dedicated calculations. Instead, the rotor flux linkage estimate is obtained from the same flux estimator and PLL unit 43 used to estimate the rotor speed and rotor angle. In addition, the rotor flux linkage estimation method is not affected by motor load conditions and motor speed.

One limitation of the rotor flux linkage estimation method is that there is a minimum motor speed at which the flux estimator 51 can effectively operate such that if the motor speed is less than the minimum motor speed, the measured flux linkage amplitude is less than the minimum motor speedAnd is unreliable. This minimum speed limit is necessary to avoid saturation of the integrator used in the flux estimator 51. The turn-off frequency ω of the non-ideal integrator of the magnetic flux estimator 51_bProvided by equation 7 below:

ωb＝1/τ_Flxequation 7

Wherein, tau_FlxIs the magnetic flux estimator time constant of the magnetic flux estimator 51. When the motor speed is less than the turn-off frequency omega of the non-ideal integrator_bThe gain of the non-ideal integrator flattens out and no longer continues to increase. Therefore, if the rotor fundamental frequency ω is expressed by equation 8_rLower than the turn-off frequency omega of a non-ideal integrator_bEstimated rotor flux linkageIs no longer accurate. Equation 8 is provided as follows:

typically, the minimum motor speed limit is about 5% of the maximum rated motor speed. Accordingly, the memory device 58 may store a minimum motor speed limit, and the minimum motor speed limit may be based on a predetermined percentage or fraction of the maximum rated motor speed of the motor M. In this case, the demagnetization detector 57 is configured to receive the measured (estimated) motor speed (MotorSpeed) from the magnetic flux PLL 55 and compare the estimated measured motor speed with the minimum motor speed limit stored in the memory device 58. The demagnetization detection monitoring of the demagnetization detector 57 is enabled on condition that the estimated measured motor speed is equal to or greater than the minimum motor speed limit. For example, the demagnetization detection monitoring itself may be initialized and activated such that the rotor flux linkage estimate is considered in the determination of the demagnetization faultOn the other hand, under the condition that the estimated measured motor speed is less than the minimum motor speed limit, the demagnetization detection monitoring by the demagnetization detector 57 is disabled. For example, when the estimated measured motor speed is less than the minimum motor speed limit, the demagnetization detector 57 may ignore or disregard any rotor flux linkage estimates received from the PLL 55Thus, the demagnetization detector 57 does not generate or output a demagnetization fault when the estimated measured motor speed is less than the minimum motor speed limit.

In view of the above, a digital motor control system for household appliance applications including, but not limited to, air conditioning units, fan controllers, refrigerators, washing machines, air cleaners, vacuum cleaners, blowers, pumps, etc., is configured to accurately estimate a rotor flux linkage using flux equations 3, 4, and 5, and to detect demagnetization of a permanent magnet of a rotor based on the estimated rotor flux linkage.

While various embodiments have been described, it will be apparent to those of ordinary skill in the art that many more embodiments and implementations are possible within the scope of the disclosure. For example, while certain embodiments may be directed to sensorless FOC, these embodiments may also use sensor-based FOC as long as the magnetic flux estimator and PLL unit 43 are active. Accordingly, the invention is not to be restricted except in light of the attached claims and their equivalents. In regard to the various functions performed by the above described components or structures (assemblies, devices, circuits, systems, etc.), the terms (including a reference to a "means") used to describe such components are intended to correspond, unless otherwise indicated, to any component or structure which performs the specified function of the described component (i.e., that is functionally equivalent), even though not structurally equivalent to the disclosed structure which performs the function in the herein illustrated exemplary implementations of the invention.

Furthermore, the following claims are hereby incorporated into the detailed description, with each claim standing on its own as a separate example embodiment. Although each claim may stand on its own as a separate example embodiment, it should be noted that, although a dependent claim may refer in the claims to a particular combination with one or more other claims, other example embodiments may also include a combination of a dependent claim with the subject matter of each other independent claim or dependent claims. Such combinations are presented herein unless the statement is not intended to be specific. Furthermore, even if the claim does not directly refer to an independent claim, it is intended to include features of the claim in any other independent claim.

It should also be noted that the methods disclosed in the specification or claims may be implemented by an apparatus having means for performing each respective action of the methods.

Further, it should be understood that the disclosure of multiple acts or functions disclosed in the specification or claims may not be construed as limited to a particular sequence. Thus, unless such acts or functions are not interchangeable for technical reasons, the disclosure of multiple acts or functions does not limit the acts or functions to a particular order. Further, in some implementations, a single action may include multiple sub-actions or may be divided into multiple sub-actions. Such sub-actions may be included and such single actions may be part of the disclosure unless expressly excluded.

Embodiments provided herein may be implemented in hardware or software, according to certain implementation requirements. The implementation can be performed using a digital storage medium, e.g. a floppy disk, a DVD, a blu-ray, a CD, a RAM, a ROM, a PROM, an EPROM, an EEPROM or a flash memory, in which electronically readable control signals are stored, which cooperate (or are capable of cooperating) with a programmable computer system such that the respective method is performed. Accordingly, the digital storage medium may be computer-readable.

The instructions may be executed by one or more processors, such as one or more Central Processing Units (CPUs), Digital Signal Processors (DSPs), general purpose microprocessors, Application Specific Integrated Circuits (ASICs), field programmable logic arrays (FPGAs), or other equivalent integrated or discrete logic circuitry. Thus, the term "processor," as used herein, refers to any of the foregoing structure or any other structure suitable for implementing the techniques described herein. In addition, in some aspects, the functions described herein may be provided within dedicated hardware and/or software modules. Also, the techniques may be fully implemented in one or more circuits or logic elements.

Thus, the techniques described in this disclosure may be implemented, at least in part, in hardware, software, firmware, or any combination thereof. For example, various aspects of the described techniques may be implemented within one or more processors, including one or more microprocessors, DSPs, ASICs, or any other equivalent integrated or discrete logic circuitry, as well as any combinations of such components.

The control unit, including hardware, may also perform one or more of the techniques described in this disclosure. Such hardware, software, and firmware may be implemented within the same device or within separate devices to support the various techniques described in this disclosure. The software may be stored on a non-transitory computer readable medium such that the non-transitory computer readable medium comprises program code or program algorithms stored thereon which, when executed, cause a computer program to perform the steps of the method.

Although various exemplary embodiments have been disclosed, it will be apparent to those skilled in the art that various changes and modifications can be made which will achieve some of the advantages of the concepts disclosed herein without departing from the spirit and scope of the invention. It will be obvious to those skilled in the art that other components performing the same function may be substituted as appropriate. It is to be understood that other embodiments may be utilized and structural or logical changes may be made without departing from the scope of the present invention. It should be mentioned that features explained with reference to a particular figure may be combined with features of other figures, even in those cases not explicitly mentioned. Such modifications to the general inventive concept are intended to be covered by the appended claims and their legal equivalents.