阅读说明：本技术 双三相永磁同步电机pmsm锯齿载波双随机svpwm控制方法 (Double-phase permanent magnet synchronous motor PMSM sawtooth carrier double-random SVPWM control method ) 是由 赵文祥 李亮 吉敬华 和阳 陶涛 田伟 于 2021-05-19 设计创作，主要内容包括：本发明公开了一种双三相永磁同步电机PMSM锯齿载波双随机SVPWM控制方法。双三相永磁同步电机采用SVPWM时会导致相电流在开关频率及其整数倍处存在集中的高频谐波,带来振动噪声问题。此外,对双三相电机开关序列的中心化处理,也会增加谐波含量。为解决上述问题,本发明结合随机零矢量SVPWM与变延时SVPWM,提出基于锯齿载波的双随机SVPWM控制方法。在不影响矢量控制运行性能的情况下,将开关频率及其整数倍处的谐波分散到指定频域内,显著降低相电流的高频谐波幅值。与此同时,采用锯齿载波代替传统的三角载波,避免了开关序列的中心化处理,降低了谐波含量。(The invention discloses a double-random SVPWM control method for a double-three-phase permanent magnet synchronous motor PMSM sawtooth carrier. When the double three-phase permanent magnet synchronous motor adopts SVPWM, concentrated high-frequency harmonic waves exist in the phase current at the switching frequency and integral multiple of the switching frequency, and the problem of vibration noise is caused. In addition, the harmonic content is also increased by the centralization treatment of the double three-phase motor switching sequence. In order to solve the problems, the invention provides a double-random SVPWM control method based on sawtooth carrier waves by combining random zero vector SVPWM and variable delay SVPWM. Under the condition of not influencing the operation performance of vector control, the harmonic waves at the switching frequency and integral multiples thereof are dispersed into a specified frequency domain, and the high-frequency harmonic amplitude of the phase current is obviously reduced. Meanwhile, the sawtooth carrier wave is adopted to replace the traditional triangular carrier wave, so that the centralization processing of a switching sequence is avoided, and the harmonic content is reduced.)

1. The double-random SVPWM control method of the double-three-phase permanent magnet synchronous motor PMSM sawtooth carrier is characterized by comprising the following steps:

step 1: firstly, obtaining the projection u of a reference voltage vector on an alpha-beta axis of a fundamental wave sub-plane and a z1-z2 axis of a third harmonic sub-plane through a PI controller and coordinate transformation_α、u_β、u_z1、u_z2；

Step 2: secondly, analyzing the maximum four-vector SVPWM principle and deducing the action time T of four large vectors in one switching period₁、T₂、T₃、T₄And the action time T of the zero vector₀₀、T₇₇Further calculating the duty ratio of each phase of the inverter in one period;

and step 3: the action time T of the zero vector is redistributed by adopting the random zero vector SVPWM₀₀And T₇₇And combining with variable delay SVPWM to make switching period T_sRandomly changing, dispersing harmonic waves at the switching frequency and integral multiples thereof into a specified frequency domain, and greatly reducing the amplitude of the harmonic waves;

and 4, step 4: finally, the duty ratio and the switching period T of each phase of the inverter are calculated_sThrough sawtooth carrier modulation, obtain the asymmetric switch sequence in center, centering of switch sequence when avoiding using the triangle carrier is handled, reduces harmonic content.

2. The dual-random SVPWM control method of dual-three-phase permanent magnet synchronous motor PMSM sawtooth carrier according to claim 1, characterized in that the specific process of step 2 is as follows:

the six-phase inverter obtains 64 space voltage vectors according to the combination of different switch states of each phase, equally divides two orthogonal subspaces into 12 sectors, adopts four-vector SVPWM for enabling a harmonic plane to be controllable, adopts the maximum four-vector SVPWM for having higher bus voltage utilization rate, namely selects four large vectors nearest to a target vector as reference voltage vectors, and the calculation process of the maximum four-vector SVPWM can be expressed as follows:

in the formula，T_y(y 1,2,3,4) represents the time during which the y-th voltage vector acts in one switching cycle; u shape_xy(x ═ α, β, z1, z2) represents the projection of the y-th voltage vector on the x-axis; u. of_α、u_β、u_z1、u_z2Representing a projection of the reference voltage vector on the respective axis; t is_sRepresents a switching cycle;

the action time of the zero vector does not affect the amplitude of the output voltage, and the action time of two zero vectors is generally equally distributed in the conventional modulation algorithm, namely:

3. the dual-random SVPWM control method of dual-three-phase permanent magnet synchronous motor PMSM sawtooth carrier according to claim 1, characterized in that the specific process of step 3 is as follows:

combining random zero vector SVPWM and variable delay SVPWM to obtain double random SVPWM, dispersing harmonic waves at the switching frequency and integral multiples thereof into a specified frequency domain, and greatly reducing the amplitude of the harmonic waves; the action time of the zero vector will not affect the amplitude of the output voltage if two zero vectors u₀₀And u₇₇The action time of the zero vector can be represented as:

wherein R is [0,1 ]]Uniformly distributed random numbers; t is₀The action time of the zero vector; t is₀₀And T₇₇Are respectively two zero vectors u₀₀And u₇₇The action time of (c);

the sampling frequency is kept unchanged while the switching frequency is randomly changed by the variable-delay SVPWM, and the average value of the switching frequency is ensured to be equal to the sampling frequency, so that the advantage of obvious harmonic suppression effect of the random switching frequency SVPWM is reserved, the control performance is not influenced, the delay time delta t of the next switching period is calculated in each sampling period, and then the expression of the nth switching period is as follows:

T_s(n)＝T_samp+Δt_n-Δt_n-1

Δt＝R_vtdT_samp

in the formula, T_sampIs a sampling period; r_vtdIs [0,1 ]]Uniformly distributed random numbers.

4. The dual-random SVPWM control method of dual-three-phase permanent magnet synchronous motor PMSM sawtooth carrier according to claim 1, characterized in that: the specific process of step 4 is as follows:

when a sawtooth carrier is adopted, each phase has two comparison values in each period, and when the carrier value is equal to the comparison value 1, the switching signal S is changed from 0 to 1; when the carrier value is equal to the comparison value 2, the switching signal S is changed from 1 to 0; changing the values of the comparison value 1 and the comparison value 2 can make each phase pulse be positioned at any position in a switching period, and directly generate a non-centrosymmetric switching sequence; the sawtooth carrier modulation is introduced into the double three-phase motor, an asymmetric switching sequence can be directly generated, and the error of a target vector caused by centralization processing is eliminated.

Technical Field

The invention relates to the field of harmonic suppression and low vibration noise of a double three-phase permanent magnet synchronous motor, in particular to a double three-phase PMSM (permanent magnet synchronous motor) sawtooth carrier double random SVPWM (space vector pulse width modulation) strategy, which is beneficial to reducing harmonic content, reducing switching frequency and harmonic amplitude at integral multiple of the switching frequency, and further reducing electromagnetic interference and vibration noise.

Background

Compared with the traditional three-phase motor, the multi-phase motor driving system has the advantages of higher power output, smaller torque ripple, higher bus voltage utilization rate, more control freedom degree, better fault-tolerant performance and the like, thereby having good application prospect in the fields of electric airplanes, ship propulsion, submarines and the like. The torque ripple of the double three-phase permanent magnet synchronous motor with the phase shift of 30 degrees is smaller than that of a common five-phase or six-phase motor, and the double three-phase permanent magnet synchronous motor has greater advantages.

The double three-phase permanent magnet synchronous motor PMSM is driven by a six-phase voltage source inverter, and concentrated high-frequency harmonic waves are generated at the switching frequency and integral multiples of the switching frequency when the power tube is switched on and switched off, so that high-frequency vibration noise and electromagnetic interference are caused. In order to solve this problem, a modulation strategy is generally used to suppress concentrated high-frequency harmonics. The main method comprises the following steps: 1) harmonic waves at odd multiples of the switching frequency are eliminated by using the improved SVPWM, but the method is only suitable for three-phase motors. 2) The harmonic waves at the even-numbered times of the switching frequency are eliminated by utilizing the staggered parallel inverters, but the method has higher requirements on the structure of the motor and is complex to realize. 3) The spread spectrum modulation technology is used, high-frequency harmonic waves are dispersed in a wider frequency range, the harmonic amplitude is reduced, the topological structure of the system does not need to be changed, the method has great research value and application prospect, and the method is a modulation strategy for inhibiting the high-frequency harmonic waves which is researched more at the present stage.

At present, the commonly used random spread spectrum modulation mainly includes random zero vector SVPWM and random switching frequency SVPWM. In contrast, the random switching frequency SVPWM has a better harmonic dispersion effect, but it causes the sampling frequency to vary randomly with the switching frequency, resulting in the regulator parameters no longer being suitable. The variable-delay SVPWM randomly changes the switching frequency and keeps the sampling frequency unchanged, overcomes the defect of the traditional random switching frequency modulation, and improves the control performance of the random modulation. The random zero vector and the random switching frequency modulation are combined, so that the switching frequency and the high-frequency harmonic waves at integral multiples of the switching frequency can be effectively dispersed into a wider frequency domain, the amplitude of the harmonic waves is greatly reduced, and better high-frequency performance is obtained.

On the other hand, in order to obtain a higher bus voltage utilization rate and more effectively control a harmonic space, the dual three-phase motor vector control system generally adopts a maximum four-vector SVPWM method, but a PWM switching sequence generated by the dual three-phase motor vector control system is not centrosymmetric, so that hardware implementation is difficult in the conventional triangular carrier modulation. In order to solve the problem, an asymmetric switching sequence is generally subjected to centering treatment, but the method can change a reference voltage vector, increase harmonic content and influence the control performance of a system.

Disclosure of Invention

Aiming at the problem of vibration noise caused by double three-phase PMSM current harmonics, the invention adopts random zero vector-variable delay double random SVPWM, disperses concentrated high-frequency harmonics into a specified frequency domain under the condition of not influencing the performance of a control system, and obviously reduces the harmonic amplitude. In addition, a non-centrosymmetric switching sequence required by the maximum four-vector SVPWM is directly obtained by using sawtooth carrier modulation, so that the centralization processing of the switching sequence when a triangular carrier is used is avoided, the control performance is more stable, the harmonic content is reduced, and the vibration noise is reduced.

In order to achieve the purpose, the invention adopts the following technical scheme:

the double-random SVPWM control method of the double-three-phase permanent magnet synchronous motor PMSM sawtooth carrier comprises the following steps:

step 1: firstly, obtaining the projection u of a reference voltage vector on an alpha-beta axis of a fundamental wave sub-plane and a z1-z2 axis of a third harmonic sub-plane through a PI controller and coordinate transformation_α、u_β、u_z1、u_z2；

Step 2: it is composed ofThe maximum four-vector SVPWM principle is analyzed, and the action time T of four large vectors in one switching period is deduced₁、T₂、T₃、T₄And the action time T of the zero vector₀₀、T₇₇Further calculating the duty ratio of each phase of the inverter in one period;

and step 3: the action time T of the zero vector is redistributed by adopting the random zero vector SVPWM₀₀And T₇₇And combining with variable delay SVPWM to make switching period T_sRandomly changing, dispersing harmonic waves at the switching frequency and integral multiples thereof into a specified frequency domain, and greatly reducing the amplitude of the harmonic waves;

and 4, step 4: finally, the duty ratio and the switching period T of each phase of the inverter are calculated_sThrough sawtooth carrier modulation, obtain the asymmetric switch sequence in center, centering of switch sequence when avoiding using the triangle carrier is handled, reduces harmonic content.

Further, the specific process of step 2 is as follows:

the six-phase inverter obtains 64 space voltage vectors according to the combination of different switch states of each phase, equally divides two orthogonal subspaces into 12 sectors, adopts four-vector SVPWM for enabling a harmonic plane to be controllable, adopts the maximum four-vector SVPWM for having higher bus voltage utilization rate, namely selects four large vectors nearest to a target vector as reference voltage vectors, and the calculation process of the maximum four-vector SVPWM can be expressed as follows:

in the formula, T_y(y 1,2,3,4) represents the time during which the y-th voltage vector acts in one switching cycle; u shape_xy(x ═ α, β, z1, z2) represents the projection of the y-th voltage vector on the x-axis; u. of_α、u_β、u_z1、u_z2Representing a projection of the reference voltage vector on the respective axis; t is_sRepresents a switching cycle;

the action time of the zero vector does not affect the amplitude of the output voltage, and the action time of two zero vectors is generally equally distributed in the conventional modulation algorithm, namely:

further, the specific process of step 3 is as follows:

combining random zero vector SVPWM and variable delay SVPWM to obtain double random SVPWM, dispersing harmonic waves at the switching frequency and integral multiples thereof into a specified frequency domain, and greatly reducing the amplitude of the harmonic waves; the action time of the zero vector will not affect the amplitude of the output voltage if two zero vectors u₀₀And u₇₇The action time of the zero vector can be represented as:

wherein R is [0,1 ]]Uniformly distributed random numbers; t is₀The action time of the zero vector; t is₀₀And T₇₇Are respectively two zero vectors u₀₀And u₇₇The action time of (c);

the sampling frequency is kept unchanged while the switching frequency is randomly changed by the variable-delay SVPWM, and the average value of the switching frequency is ensured to be equal to the sampling frequency, so that the advantage of obvious harmonic suppression effect of the random switching frequency SVPWM is reserved, the control performance is not influenced, the delay time delta t of the next switching period is calculated in each sampling period, and then the expression of the nth switching period is as follows:

T_s(n)＝T_samp+Δt_n-Δt_n-1

Δt＝R_vtdT_samp

in the formula, T_sampIs a sampling period; r_vtdIs [0,1 ]]Uniformly distributed random numbers.

Further, the specific process of step 4 is as follows:

when a sawtooth carrier is adopted, each phase has two comparison values in each period, and when the carrier value is equal to the comparison value 1, the switching signal S is changed from 0 to 1; when the carrier value is equal to the comparison value 2, the switching signal S is changed from 1 to 0; changing the values of the comparison value 1 and the comparison value 2 can make each phase pulse be positioned at any position in a switching period, and directly generate a non-centrosymmetric switching sequence; the sawtooth carrier modulation is introduced into the double three-phase motor, an asymmetric switching sequence can be directly generated, and the error of a target vector caused by centralization processing is eliminated.

The invention combines random zero vector SVPWM and variable delay SVPWM to obtain double random SVPWM, disperses harmonic waves at the switching frequency and integral multiple thereof into a specified frequency domain, greatly reduces the amplitude of the harmonic waves, and reduces electromagnetic interference and vibration noise.

The double random SVPWM combining the variable-delay SVPWM and the random zero vector SVPWM can disperse high-frequency harmonics at the switching frequency and integral multiples thereof into a specified frequency domain, obviously reduce the harmonic amplitude, does not influence the control performance of a system, and has a harmonic suppression effect superior to that of any single random method.

The invention uses sawtooth carrier wave to replace traditional triangle carrier wave, avoids the centralization processing of switch sequence, and reduces harmonic content. The sawtooth carrier modulation is introduced into the double three-phase motor, an asymmetric switching sequence can be directly generated, and the error of a target vector caused by centralization processing is eliminated, so that the harmonic content is reduced, the control performance of the motor is improved, and the defects of the double three-phase motor when a triangular carrier is used are overcome.

The invention has the beneficial effects that:

1. the invention uses the variable delay SVPWM with invariable sampling frequency, overcomes the defect of the non-adaptation of the parameters of the regulator caused by the random change of the sampling frequency of the traditional random switching frequency SVPWM, and improves the stability of the system.

2. The invention combines the random zero vector SVPWM and the variable delay SVPWM to obtain the double random SVPWM, combines the advantages of two random methods, has more excellent harmonic dispersion performance than any single random method, can effectively disperse the harmonic at the switching frequency and integral multiple thereof into a specified frequency domain, obviously reduces the harmonic amplitude, and further reduces the electromagnetic interference and the vibration noise.

3. In the double three-phase motor, the sawtooth carrier modulation is adopted to replace the traditional triangular carrier modulation, so that the problem of increasing the harmonic content caused by the centralized processing of a switching sequence can be avoided, the harmonic content is favorably reduced, and the electromagnetic interference and the vibration noise are further reduced.

Drawings

FIG. 1 is a control block diagram of a dual-phase PMSM sawtooth carrier dual-random SVPWM method;

FIG. 2 is a six-phase voltage source inverter topology with neutral isolation;

FIG. 3 is a six-phase inverter space voltage vector distribution diagram; wherein, (a) is a voltage vector distribution diagram of a fundamental sub-plane, and (b) is a voltage vector distribution diagram of a third harmonic sub-plane;

FIG. 4 is a schematic diagram of a sawtooth carrier dual random SVPWM;

FIG. 5 is a schematic diagram of the principle of sawtooth carrier modulation and switching sequence; wherein, (a) is a schematic diagram of sawtooth carrier modulation, and (b) is a switching sequence of sawtooth carrier modulation;

FIG. 6 is a graph of phase current versus harmonic content effects; wherein, (a) is a phase current and harmonic content effect diagram during sawtooth carrier modulation, and (b) is a phase current and harmonic content effect diagram during triangular carrier modulation;

FIG. 7 is a graph of a power spectrum analysis of phase currents; wherein, (a) is a power spectrum analysis diagram of the time phase current by adopting the traditional SVPWM, and (b) is a power spectrum analysis diagram of the time phase current by adopting the double random SVPWM.

Detailed description of the invention

The technical solution in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention.

Reference will now be made in detail to embodiments of the present invention, examples of which are illustrated in the accompanying drawings, wherein like or similar reference numerals refer to the same or similar elements or elements having the same or similar function throughout. The embodiments described below with reference to the accompanying drawings are illustrative only for the purpose of explaining the present invention, and are not to be construed as limiting the present invention.

As shown in the structural block diagram of fig. 1, the invention relates to a dual-random SVPWM strategy of a dual-three-phase PMSM sawtooth carrier, which mainly comprises dual-random SVPWM and sawtooth carrier modulation.

The invention takes a double three-phase PMSM as a control object, and carries out harmonic suppression on phase current, and the specific measures are as follows:

step 1: as shown in the structural block diagram of FIG. 1, firstly, the projections u of the reference voltage vector on the alpha-beta axis of the fundamental wave sub-plane and the z1-z2 axis of the third harmonic sub-plane are obtained through PI controller and coordinate transformation_α、u_β、u_z1、u_z2And the maximum four-vector SVPWM is used as the input of the dual three-phase PMSM.

Step 2: analyzing the maximum four-vector SVPWM principle and deducing the action time T of four large vectors in one switching period₁、T₂、T₃、T₄And the action time T of the zero vector₀₀、T₇₇. And further calculating the duty ratio of each phase of the inverter in one period.

The topology of a neutral point isolated six-phase voltage source inverter is shown in fig. 2. In the figure, the DC bus voltage is U_dcEach phase is provided with an upper bridge arm and a lower bridge arm, the on and off of a power switch tube of each phase are controlled by a PWM signal, and only one switch tube of each phase is switched on at the same time. Defining a switching function S ═ S_a S_b S_c S_d S_e S_f]Taking phase a as an example, S is generated when the upper bridge arm switching tube is conducted_aWhen the lower bridge arm switching tube is conducted, S is equal to 1_a0, but also for other phases, the voltage vectors in the α - β subspace and the z1-z2 subspace can be expressed as:

wherein a ═ e^jπ/6. The subscript of each voltage vector is written as a two-bit octal number, converted by a binary switching function, for example: u. of₄₅Indicates that the switching function S is [ 100101 ]]。

According to the combination of different switch states of each phase, 64 space voltage vectors can be obtained, including 60 non-zero vectors and 4 zero vectors (u)₀₀，u₀₇，u₇₀，u₇₇). Fig. 3 shows the voltage vector distribution for the α - β subspace and the z1-z2 subspace, with 64 voltage vectors equally dividing the two orthogonal subspaces into 12 sectors. To make the harmonic plane controllable, six-phase voltage source inverters typically employ four-vector SVPWM. In order to have higher bus voltage utilization rate, the maximum four-vector SVPWM is adopted, namely four large vectors which are closest to a target vector are selected as reference voltage vectors. Take sector I as an example, u is selected₄₄、u₄₅、u₅₅、u₆₄Is a reference voltage vector, as shown by the voltage vector within the circle in fig. 3. The calculation process of the maximum four-vector SVPWM can be expressed as:

in the formula, T_yRepresents the time during which the y-th voltage vector acts during a switching cycle; u shape_xyRepresents the projection of the y-th voltage vector on the x-axis; u. of_α、u_β、u_z1And u_z2Representing a projection of the reference voltage vector on the respective axis; t is_sIndicating the switching period.

The action time of the zero vector does not influence the amplitude of the output voltage, and the action time of two zero vectors is generally evenly distributed in the traditional modulation algorithm, namely

After the action time of the four large vectors and the two zero vectors is obtained, the switching sequence shown in fig. 5 can be deduced according to the principle that the switching state of each phase changes only once in each period, and further the duty ratio of each phase of the inverter is obtained.

And step 3: the action time T of the zero vector is redistributed by adopting the random zero vector SVPWM₀₀And T₇₇. And combining with variable delay SVPWM to make switching period T_sAnd randomly changing, dispersing the harmonic waves at the switching frequency and integral multiples thereof into a specified frequency domain, and greatly reducing the amplitude of the harmonic waves.

The action time of the zero vector does not affect the amplitude of the output voltage. If two zero vectors u are to be combined₀₀And u₇₇While keeping the effective vector function time constant, as shown in fig. 4, the output fundamental voltage synthesized by the inverter does not change, but only the switching frequency and the harmonic amplitudes at integral multiples thereof are reduced. The action time of the zero vector can be expressed as:

wherein R is [0,1 ]]Uniformly distributed random numbers. T is₀The action time of the zero vector; t is₀₀And T₇₇Are respectively two zero vectors u₀₀And u₇₇The action time of (1). It is noted that as the motor speed increases, the modulation ratio increases and the action time of the zero vector decreases gradually. Therefore, the high frequency harmonic suppression effect of the random zero vector SVPWM is reduced with the increase of the motor rotation speed.

The switching frequency is randomly changed by the variable-delay SVPWM, meanwhile, the sampling frequency is kept unchanged, and the average value of the switching frequency is ensured to be equal to the sampling frequency. Therefore, the advantage of obvious harmonic suppression effect of the random switching frequency SVPWM is reserved, and the control performance is not influenced. As shown in fig. 4, the delay time Δ t of the next switching period is calculated in each sampling period, and the expression of the nth switching period is:

T_s(n)＝T_samp+Δt_n-Δt_n-1

Δt＝R_vtdT_samp

in the formula, T_sampIs a sampling period; r_vtdIs [0,1 ]]Uniformly distributed random numbers. As can be seen from fig. 4, the maximum value of the delay time Δ T of the switching period cannot exceed T_sampTherefore, the next switching period can be ensured not to influence the output of the current switching period, so that the maximum value of the switching period is 2T_samp. In order to prevent too small a switching period from occurring, the minimum value of the switching period must be limited to T_smin. In order to obtain a better high-frequency harmonic dispersion effect, the switching period needs to be randomly changed within a maximum range. The switching frequency varies randomly in the range of

T_smin≤T_s≤2T_samp

The double random SVPWM combining the variable-delay SVPWM and the random zero vector SVPWM can disperse high-frequency harmonics of the switching frequency and integral multiples thereof into a specified frequency domain, obviously reduce harmonic amplitude, reduce electromagnetic interference and vibration noise, does not influence the control performance of a system, and has a harmonic suppression effect superior to that of any single random method.

And 4, step 4: finally, the duty ratio and the switching period T of each phase of the inverter are calculated_sThrough sawtooth carrier modulation, obtain the asymmetric switch sequence in center, centering of switch sequence when avoiding using the triangle carrier is handled, reduces harmonic content.

Fig. 5(a) shows the principle of sawtooth carrier modulation, with a carrier amplitude of 1 and two comparison values per phase per cycle. When the carrier value is equal to the comparison value 1, the switching signal S is changed from 0 to 1; when the carrier value is equal to the comparison value 2, the switching signal S changes from 1 to 0. Fig. 5(b) shows the switching sequence of the maximum four-vector SVPWM during the sawtooth carrier modulation, and it can be seen from the figure that when the sawtooth carrier is used, the values of the comparison value 1 and the comparison value 2 are changed, so that each phase pulse can be positioned at any position in one switching period, and a non-centrosymmetric switching sequence is directly generated.

The sawtooth carrier modulation is introduced into the double three-phase motor, an asymmetric switching sequence can be directly generated, and the error of a target vector caused by centralization processing is eliminated, so that the harmonic content is reduced, the control performance of the motor is improved, and the defects of the double three-phase motor when a triangular carrier is used are overcome.

FIG. 6(a) is a graph showing the effect of phase current and harmonic content under sawtooth carrier modulation, where the THD of the phase current is 6.45%; fig. 6(b) is a graph showing the effect of phase current and harmonic content under triangular carrier modulation, and the THD of the phase current is 9.55%. It can be seen that the sawtooth carrier modulation proposed by the present invention can effectively reduce the harmonic content.

FIG. 7 is a power spectrum analysis diagram of phase current under sawtooth carrier modulation, from which it can be seen that the current spectrum of the common SVPWM algorithm has an obvious peak at the switching frequency and its integral multiple, and the maximum harmonic amplitude is-23.7 dB; when the double random SVPWM is adopted, the power spectrum has no obvious peak and is close to a white noise spectrum, the maximum harmonic amplitude is-49.2 dB, and the maximum harmonic amplitude is reduced by 25.5 dB. According to the analysis, the double-random SVPWM provided by the invention can effectively disperse the harmonic waves at the switching frequency and integral multiples thereof into a specified frequency domain, obviously reduce the amplitude of the harmonic waves and further reduce the vibration noise.

In the description herein, references to the description of the terms "one embodiment," "some embodiments," "an illustrative embodiment," "an example," "a specific example," or "some examples" or the like mean that a particular feature, structure, material, or characteristic described in connection with the embodiment or example is included in at least one embodiment or example of the invention. In this specification, the schematic representations of the terms used above do not necessarily refer to the same embodiment or example. Furthermore, the particular features, structures, materials, or characteristics described may be combined in any suitable manner in any one or more embodiments or examples.

While embodiments of the invention have been shown and described, it will be understood by those of ordinary skill in the art that: various changes, modifications, substitutions and alterations can be made to the embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the claims and their equivalents.