阅读说明：本技术 一种电磁流量转换器 (Electromagnetic flow converter ) 是由 黄建彪 孙宏泉 邓君 强欢 张昕 余武 姜云贺 刘伟海 于 2021-08-09 设计创作，主要内容包括：本发明涉及一种电磁流量转换器,包括电源电路、可控恒流励磁电路、可控恒流源、励磁检测电路、可控放大电路、AD转换电路、空管检测电路、键盘与显示电路、通信接口电路。将可控电流源和可控增益放大器结合使用,分别对运放电路放大倍率和传感器励磁电流进行自动调控；通过当前计算的流量值大小,自动选择合适的放大倍率和合适的励磁电流,从而使流量计的测量范围变得更宽,量程比更大,适用范围更广。不同的放大倍率与不同的励磁电流构成了多个传统流量计的组合形式,任一组合下,相当于独立的传统电磁流量计在工作,并可以进行无缝切换,仅使用同一套硬件电路和同一套软件算法,极大地节约了成本,提高了资源的使用效率。(The invention relates to an electromagnetic flow converter, which comprises a power supply circuit, a controllable constant-current excitation circuit, a controllable constant-current source, an excitation detection circuit, a controllable amplification circuit, an AD conversion circuit, an air traffic control detection circuit, a keyboard and display circuit and a communication interface circuit. The controllable current source and the controllable gain amplifier are combined for use, and the amplification factor of the operational amplifier circuit and the excitation current of the sensor are respectively and automatically regulated and controlled; through the current calculated flow value, the proper amplification factor and the proper exciting current are automatically selected, so that the measuring range of the flowmeter is wider, the measuring range ratio is larger, and the application range is wider. Different amplification factors and different exciting currents form a combination form of a plurality of traditional flowmeters, and under any combination, the electromagnetic flowmeter works equivalently to an independent traditional electromagnetic flowmeter, seamless switching can be performed, only the same hardware circuit and the same software algorithm are used, the cost is greatly saved, and the use efficiency of resources is improved.)

1. An electromagnetic flow converter, comprising: the device comprises a power supply circuit, a controllable constant-current excitation circuit, a controllable constant-current source, an excitation detection circuit, a controllable amplification circuit, an AD conversion circuit, an air traffic control detection circuit, a keyboard and display circuit and a communication interface circuit; the power circuit uses an isolated switch power supply to provide a plurality of groups of different voltages and provide a stable isolated power supply for the electromagnetic flow converter; the controllable constant-current excitation circuit is used for providing direction controllable current for the excitation coil of the sensor; the controllable constant current source is used for controlling the current of the controllable constant current excitation circuit flowing through the excitation coil; the excitation detection circuit is used for judging whether the controllable constant-current excitation circuit works normally or not; the controllable amplifying circuit is used for carrying out differential acquisition, band-pass filtering and level shifting processing on the analog voltage signal so as to facilitate AD conversion on the output voltage signal; the AD conversion circuit is used for processing the voltage signal output by the controllable amplification circuit and converting the analog quantity into the digital quantity, so that the MCU can conveniently process the voltage signal; the empty pipe detection circuit is used for judging whether the liquid is full; the keyboard and the display circuit are used for realizing local human-computer interaction; and the communication interface circuit is used for realizing remote data interaction.

2. The electromagnetic flow converter of claim 1, wherein: the controllable amplifying circuit includes: CD4053BPWR chip, operational amplifier U10A, operational amplifier U10B, resistors R30, R31, R32, R33, R34, R35, R36, R37, R38, R39, R40, R41, and capacitors C12, C13, C14, C15, C16, C17, C18;

pin 16 of the CD4053BPWR chip is connected with a +5V power supply; pins 6 and 8 of the CD4053BPWR chip are grounded; pins 9 and 10 of the CD4053BPWR chip are connected to pins of the single chip microcomputer; pin 2 of the CD4053BPWR chip is connected to pin 4; pin 7 of the CD4053BPWR chip is connected with a-5V power supply and is connected with the ground through a capacitor C18; the resistors R38, R39, R40 and R41 are connected in series, and one end of the resistor R41, which is not connected with the resistor R40, is grounded; pin 1 of the CD4053BPWR chip is connected to a node between the resistors R38, R39, pin 3 of the chip is connected to a node between the resistors R39, R40, and pin 5 of the chip is connected to a node between the resistors R40, R41; the end of the resistor R38 which is not connected with the resistor R39 is respectively connected with one end of the resistor R35 and the output end of the operational amplifier U10A; the other end of the resistor R35 is connected with one end of a resistor R36 and the inverting input end of the operational amplifier U10B respectively, and the output end of the operational amplifier U10B is connected with the other end of a resistor R36 and one end of a resistor R34 respectively; the other end of the resistor R34 is connected with one end of the capacitor C15 and the input end of the ADC respectively, and the other end of the capacitor C15 is grounded; the non-inverting input end of the operational amplifier U10B is connected with one end of a capacitor C13, one end of a resistor R30 and one end of a resistor R31 respectively, and the other end of a resistor R31 is connected with the other end of a capacitor C13 and one end of a capacitor C12 and is grounded; the other end of the capacitor C12 is connected with the other end of the resistor R30 and connected with a +5V power supply in parallel; the inverting input terminal of the operational amplifier U10A is connected to pin 15 of the CD4053BPWR chip; the non-inverting input end of the operational amplifier U10A is connected with one end of a resistor R33, and the other end of the resistor R33 is connected with an analog voltage signal; the positive power supply end of the operational amplifier U10A is respectively connected with one end of a resistor R32 and one end of a capacitor C14, the other end of the resistor R32 is connected with a +5V power supply, and the other end of the capacitor C14 is grounded; the negative power end of the operational amplifier U10A is respectively connected with one end of a resistor R37 and one end of a capacitor C16, the other end of the resistor R37 is connected with a-5V power supply, and the other end of the capacitor C16 is grounded.

3. The electromagnetic flow converter of claim 2, wherein: the operational amplifier U10A and operational amplifier U10B are model numbers TL072IDR or TLC2272 AI.

4. The electromagnetic flow converter of claim 1, wherein: the controllable constant current excitation circuit comprises: a photoelectric coupler, an inverter U5, an operational amplifier U6B, an operational amplifier U6A, a metal-oxide semiconductor field effect transistor, an H bridge circuit, a resistor R12, R13, R14, R15, R16, R17, R18, R19, R20, R21, R22, R23, R24, a capacitor C4, C5, C6, C7, C8, C9, C10;

pin 2 of the photoelectric coupler is connected to a power supply of 3.3V through a resistor R13; a pin 3 of the photoelectric coupler is connected with a PWM signal; pin 5 of the photoelectric coupler is grounded; a pin 6 of the photoelectric coupler is respectively connected with one end of a resistor R14 and one end of a resistor R15, and a pin 8 of the photoelectric coupler is connected with one end of a resistor R11; the other end of the resistor R14, the other end of the resistor R11 and one end of the capacitor C5 are connected and connected with V1; the other end of the capacitor C5 is grounded; the other end of the resistor R15 is connected with the input end of the inverter U5, the output end of the inverter U5 is connected with one end of a resistor R16, the other end of the resistor R16 is respectively connected with one end of a resistor R17 and one end of a capacitor C8, and the other end of the capacitor C8 is grounded; the other end of the resistor R17 is connected with one end of a resistor R18 and one end of a capacitor C7 respectively, the other end of the resistor R18 is connected with the non-inverting input end of the operational amplifier U6B and one end of a capacitor C9 respectively, and the other end of the capacitor C9 is grounded; the inverting input end of the operational amplifier U6B is connected with one end of a resistor R21, the other end of the resistor R21, the other end of a capacitor C7 and the output end of the operational amplifier U6B are connected with the non-inverting input end of the operational amplifier U6A, the negative power end of the operational amplifier U6A is grounded, and the positive power end of the operational amplifier U6A uses a low-power filter formed by the resistor R12 and the capacitor C6 to suppress power supply noise; the inverting input end of the operational amplifier U6A is respectively connected with one end of the resistor R20 and one end of the capacitor C10; the other end of the resistor R20 is connected with one end of each of the resistors R22, R23 and R24, and the other ends of the resistors R22, R23 and R24 are connected and grounded; the other end of the capacitor C10 is connected with the output end of the operational amplifier U6A; the output end of the operational amplifier U6A is connected with one end of a resistor R19, the other end of the resistor R19 is connected with the grid of a metal-oxide semiconductor field effect transistor, the drain of the metal-oxide semiconductor field effect transistor is connected with an H-bridge circuit, and the source of the H-bridge circuit is respectively connected with one end of a resistor R22, a resistor R23 and a resistor R24.

5. The electromagnetic flow converter of claim 4, wherein: the model of the inverter U5 is SN74AHC1G 04; the photoelectric coupler adopts a high-speed photoelectric coupler, and the model is 6N 137; the models of the operational amplifier U6B and the operational amplifier U6A are TLC27L2 IDR; the metal-oxide semiconductor field effect transistor is a MOS transistor with model number ZVN 2120.

6. The electromagnetic flow converter of claim 4, wherein: the supply voltage V2 of the inverter U5 is provided by a high precision voltage reference chip.

7. The electromagnetic flow converter of claim 4, wherein: the power supply voltage V2 of the inverter U5 is provided by a reference voltage generating circuit, and the reference voltage generating circuit adopts a mode of dividing reference voltage and adding an operational amplifier; a voltage reference chip of 1.25V or 2.5V is used for obtaining a reference voltage, a reference voltage V2ref is obtained through resistance voltage division, and the power supply voltage V2 is obtained after the V2ref is subjected to operational amplification.

8. The electromagnetic flow converter of claim 1, wherein: the communication interface circuit comprises an RS485 module, a 4-20mA module, a hart module, a Bluetooth module, a wifi module and a sub-g module.

9. The electromagnetic flow converter of claim 1, wherein: the electromagnetic flow converter has the functions of storing and calling multi-gear correction parameters, and automatically adjusts the controllable amplifying circuit and the controllable exciting current according to the current calculated flow; different amplification factors and different exciting currents form a plurality of measurement combination forms, each combination sets a measurement range, so that an overlapped measurement interval exists between the combinations, and when the calculated flow exceeds the measurement range of the gear, gear switching is executed, and frequent gear switching is avoided; in order to improve the measurement accuracy, a calibrated instrument coefficient is arranged under each combination; in order to reduce the calibration complexity, influence factors of independently changing the amplification factor and the exciting current are calibrated respectively by carrying out influence factor decomposition on the amplification factor and the exciting current, and further, all combined instrument coefficients are obtained.

Technical Field

The invention relates to an electromagnetic flowmeter, in particular to an electromagnetic flow converter.

Background

The traditional electromagnetic flow converter is a core component of an electromagnetic flowmeter, and the setting of a meter coefficient is realized by setting a converter coefficient and a sensor coefficient. The converter coefficient is mainly used for calibrating the performance of the converter, so that the same tool works on the converter to achieve the same flow output, and further obtain the corresponding converter coefficient, and the converter coefficient is mainly used for enabling the converter coefficients to achieve the consistent level among the converters, and finally achieving the purpose of direct replacement. The sensor coefficient is usually obtained by calibration through a water flow calibration device, and the sensor and the converter are usually used together in order to improve the field use precision. Although the principle and the target are replaceable, in practical application, the measurement device is used together with the measurement device in order to ensure the measurement accuracy.

The traditional electromagnetic flow converter has only one set of calibration parameters, and once the traditional electromagnetic flow converter is subjected to factory calibration, the exciting current and the instrument coefficient are fixed. The precision is only effective in a calibration range, and only one precision index is generally given. The traditional electromagnetic flow converter is pale and powerless for the application environment with wide range ratio and places with certain requirements on precision.

In order to realize normal use at low flow rate on site, an operational amplifier circuit is usually designed as a multi-stage adjustable circuit, and voltage amplification is realized by artificially setting an amplification stage, so that measurement of low flow rate signals is realized. However, this solution can only be used in the case where the flow rate is always relatively low, and when the flow rate is increased, the signal output by the sensor is also increased, and after the amplification by the amplifying circuit, the operational amplifier circuit may reach a saturation state, and the linear amplification function is lost, which may eventually cause a measurement error. The scheme needs manual intervention multiplying power setting, cannot measure large and small flows at the same time, and only one set of calibration parameters exist in the design of the traditional electromagnetic flowmeter, so that the precision may have great difference between small flows and large flows.

In order to avoid the trouble of large and small flow precision difference, the precision is usually improved by using a multi-stage correction mode, and the method is a method which is purely temporary, permanent and non-permanent, when the flow speed is too small, the effective signal output by the sensor is too small, so that the sensor is easily interfered, and meanwhile, because the voltage after the operational amplifier is amplified is too low, the noise of the operational amplifier circuit contributes to the flow, and finally the deviation of the measurement result is caused.

The matching of the sensor is realized by artificially setting the exciting current, and the exciting current is usually changed by disassembling and assembling the sampling resistor. In this way, the matching of the sensors is completely achieved in a manual way, and the automatic assembly can be achieved by using a robot assembly. However, when it comes to the field, if irregular installation occurs, for example, for convenience, the sensor is taken out and installed on the pipeline, and then the converter is installed. Finally, there is a problem that the shipment is completed, but the shipment is not completed when it is installed, resulting in mixed loading and use. For sensors and converters with the same excitation current, it is possible to use the same by changing the aperture, but if the excitation current is different, this directly leads to measurement bias problems.

Disclosure of Invention

Traditional electromagnetic flowmeter has and only has one set of instrument coefficient, greatly restricts intelligent flowmeter's development, has leaded to it to realize great range ratio, has the problem that can't realize simultaneously taking into account big or small flow.

Therefore, the range ratio of the flowmeter is improved by a method of automatically adjusting the amplification factor of the operational amplifier and automatically adjusting the exciting current. The measurement precision is ensured by the method of multi-gear parameter storage and automatic calling.

The problem of can't realize great range ratio to traditional electromagnetic flowmeter in the use, being simultaneously considered to big flow promptly, through the magnification of automatic adjustment operational amplifier, avoid artifical manual setting, realize automatic many grades of signal amplification to realize wide range ratio. Because ideal electronic components cannot exist, the amplification factor is corrected by using the calibration parameters switched between gears, and when the gears are switched, the corresponding calibration parameters are automatically multiplied to realize flow calculation so as to achieve the target of wide range ratio. In order to realize measurement with higher precision, the flow calibration can be carried out under different gears, the calibration parameters of the corresponding gears are used for flow calculation when the gears are switched, and the precision of flow conversion is improved while the wide-range ratio is realized.

The acquisition circuit uses an amplifier circuit with variable amplification factor, a plurality of amplification gears are designed, each gear corresponds to a measurement range, and when the calculated flow rate exceeds the range, the gears can be automatically switched. There is the overlapping measurement interval between the gear, when being higher than this shelves and measuring the upper limit or be less than this shelves and measure the time limit, switch over automatically, because there is the overlapping measurement interval between the gear, can effectively avoid because factors such as interference cause erroneous judgement and frequent switching. The processing has the advantages that the amplitude of the output of the analog amplifying circuit is improved, and the subsequent data acquisition is facilitated.

In order to further improve the range ratio, the excitation circuit is adjusted by using a controllable constant current source, and in order to avoid mutual interference, a constant current control circuit which can implement a photoelectric isolation technology is preferred. The constant current control circuit based on PWM control realizes current regulation by changing duty ratio, corrects the exciting current in order to improve the output precision of the exciting current, and stores the corrected result. The isolated constant current control circuit can be realized in various modes, wherein the PWM mode has fewer pins and can accurately control the current, so the method is preferred in the invention. Of course, the multi-step current regulation can also be realized by using a voltage reference and a controllable gain amplifier.

Under different excitation currents, the signal amplitudes output by the sensor are different, so that the same instrument coefficient cannot be used, and in order to ensure the normal use of the flowmeter, the multi-gear excitation current is calibrated to obtain the relation between the excitation current and the output of the sensor, so that the instrument coefficient is corrected.

The technical scheme of the invention is as follows:

an electromagnetic flow converter comprising: the device comprises a power supply circuit, a controllable constant-current excitation circuit, a controllable constant-current source, an excitation detection circuit, a controllable amplification circuit, an AD conversion circuit (namely an ADC chip for converting analog quantity into digital quantity), an air traffic control detection circuit, a keyboard and display circuit and a communication interface circuit. The power circuit uses an isolated switch power supply to provide a plurality of groups of different voltages and provide a stable isolated power supply for the electromagnetic flow converter; the controllable constant-current excitation circuit is used for providing direction controllable current for the excitation coil of the sensor; the controllable constant current source is used for controlling the current of the controllable constant current excitation circuit flowing through the excitation coil; the excitation detection circuit is used for judging whether the controllable constant-current excitation circuit works normally or not; the controllable amplifying circuit is used for carrying out differential acquisition, band-pass filtering, level shifting and other processing on the analog voltage signal so as to enable the output voltage signal to be convenient for AD conversion; the AD conversion circuit is used for processing the voltage signal output by the controllable amplification circuit and converting the analog quantity into the digital quantity, so that the MCU can conveniently process the voltage signal; the empty pipe detection circuit is used for judging whether the liquid is full; the keyboard and the display circuit are used for realizing local human-computer interaction; the communication interface circuit is used for realizing remote data interaction.

In circuit design, RC filters are used for power supplies of the operational amplifiers to suppress power supply ripples, and the core of the controllable amplification circuit is to dynamically adjust a feedback network.

The controllable amplifying circuit includes: CD4053BPWR chip, operational amplifier U10A, operational amplifier U10B, resistors R30, R31, R32, R33, R34, R35, R36, R37, R38, R39, R40, R41, and capacitors C12, C13, C14, C15, C16, C17, C18;

pin 16 of the CD4053BPWR chip is connected with a +5V power supply; pins 6 and 8 of the CD4053BPWR chip are grounded; pins 9 and 10 of the CD4053BPWR chip are connected to pins of the single chip microcomputer (the single chip microcomputer controls the on-off state of the CD4053BPWR chip through the high and low levels of the pins); pin 2 of the CD4053BPWR chip is connected to pin 4; pin 7 of the CD4053BPWR chip is connected with a-5V power supply and is connected with the ground through a capacitor C18; the resistors R38, R39, R40 and R41 are connected in series, and one end of the resistor R41, which is not connected with the resistor R40, is grounded; pin 1 of the CD4053BPWR chip is connected to a node between the resistors R38, R39, pin 3 of the chip is connected to a node between the resistors R39, R40, and pin 5 of the chip is connected to a node between the resistors R40, R41; the end of the resistor R38 which is not connected with the resistor R39 is respectively connected with one end of the resistor R35 and the output end of the operational amplifier U10A; the other end of the resistor R35 is connected with one end of a resistor R36 and the inverting input end of the operational amplifier U10B respectively, and the output end of the operational amplifier U10B is connected with the other end of a resistor R36 and one end of a resistor R34 respectively; the other end of the resistor R34 is connected with one end of the capacitor C15 and the input end of the ADC respectively, and the other end of the capacitor C15 is grounded; the non-inverting input end of the operational amplifier U10B is connected with one end of a capacitor C13, one end of a resistor R30 and one end of a resistor R31 respectively, and the other end of a resistor R31 is connected with the other end of a capacitor C13 and one end of a capacitor C12 and is grounded; the other end of the capacitor C12 is connected with the other end of the resistor R30 and connected with a +5V power supply in parallel; the inverting input terminal of the operational amplifier U10A is connected to pin 15 of the CD4053BPWR chip; the non-inverting input end of the operational amplifier U10A is connected with one end of a resistor R33, and the other end of the resistor R33 is connected with an analog voltage signal; the positive power supply end of the operational amplifier U10A is respectively connected with one end of a resistor R32 and one end of a capacitor C14, the other end of the resistor R32 is connected with a +5V power supply, and the other end of the capacitor C14 is grounded; the negative power end of the operational amplifier U10A is respectively connected with one end of a resistor R37 and one end of a capacitor C16, the other end of the resistor R37 is connected with a-5V power supply, and the other end of the capacitor C16 is grounded.

The models of the operational amplifier U10A and the operational amplifier U10B are TL072IDR or TLC2272 AI.

The controllable constant current excitation circuit comprises: a photoelectric coupler, an inverter U5, an operational amplifier U6B, an operational amplifier U6A, a metal-oxide semiconductor field effect transistor, an H bridge circuit, a resistor R12, R13, R14, R15, R16, R17, R18, R19, R20, R21, R22, R23, R24, a capacitor C4, C5, C6, C7, C8, C9, C10;

the model of the inverter U5 is SN74AHC1G 04; the optical coupler uses a high-speed optical coupler, such as 6N 137; the models of the operational amplifier U6B and the operational amplifier U6A are TLC27L2 IDR; the metal-oxide semiconductor field effect transistor is a MOS transistor with model number ZVN 2120.

Pin 2 of the photoelectric coupler is connected to a power supply of 3.3V through a resistor R13; a pin 3 of the photoelectric coupler is connected with a PWM signal; pin 5 of the photoelectric coupler is grounded; a pin 6 of the photoelectric coupler is respectively connected with one end of a resistor R14 and one end of a resistor R15, and a pin 8 of the photoelectric coupler is connected with one end of a resistor R11; the other end of the resistor R14, the other end of the resistor R11 and one end of the capacitor C5 are connected and connected with the V1(V1 is a power supply of the optical coupler, the rear inverter U5 shapes the output waveform of the optical coupler, and the waveform voltage is determined by the voltage V2 of the inverter U5, so that the voltage range of the V1 is wide and the values are relatively random); the other end of the capacitor C5 is grounded; the other end of the resistor R15 is connected with the input end of the inverter U5, the output end of the inverter U5 is connected with one end of a resistor R16, the other end of the resistor R16 is respectively connected with one end of a resistor R17 and one end of a capacitor C8, and the other end of the capacitor C8 is grounded; the other end of the resistor R17 is connected with one end of a resistor R18 and one end of a capacitor C7 respectively, the other end of the resistor R18 is connected with the non-inverting input end of an operational amplifier U6B (a first-stage operational amplifier uses a 3-order low-pass filter to effectively avoid high-frequency leakage) and one end of a capacitor C9 respectively, and the other end of the capacitor C9 is grounded; the inverting input end of the operational amplifier U6B is connected with one end of a resistor R21, the other end of the resistor R21, the other end of a capacitor C7 and the output end of the operational amplifier U6B are connected with the non-inverting input end of the operational amplifier U6A, the negative power end of the operational amplifier U6A is grounded, and the positive power end of the operational amplifier U6A uses a low-power filter formed by the resistor R12 and the capacitor C6 to suppress power supply noise; the inverting input end of the operational amplifier U6A is respectively connected with one end of the resistor R20 and one end of the capacitor C10; the other end of the resistor R20 is connected with one end of each of the resistors R22, R23 and R24, and the other ends of the resistors R22, R23 and R24 are connected and grounded; the other end of the capacitor C10 is connected with the output end of the operational amplifier U6A; the output end of the operational amplifier U6A is connected with one end of a resistor R19, the other end of the resistor R19 is connected with the grid of a metal-oxide semiconductor field effect transistor, the drain of the metal-oxide semiconductor field effect transistor is connected with an H-bridge circuit, and the source of the H-bridge circuit is respectively connected with one end of a resistor R22, a resistor R23 and a resistor R24.

The supply voltage V2 of inverter U5 is provided by a high precision voltage reference chip,

alternatively, the power supply voltage V2 of the inverter U5 is supplied by a reference voltage generation circuit that uses a reference voltage division and operation following manner (a reference voltage is obtained using a 1.25V or 2.5V voltage reference chip, a reference voltage V2ref is obtained through resistance division, and then the power supply voltage V2 is obtained through operation following V2 ref).

The communication interface circuit comprises an RS485 module, a 4-20mA module, a hart module, a Bluetooth module, a wifi module and a sub-g module.

The electromagnetic flow converter has the functions of storing and calling multi-gear correction parameters, and automatically adjusts the controllable amplifying circuit and the controllable exciting current according to the current calculated flow; different amplification factors and different exciting currents form a plurality of measurement combination forms, each combination sets a measurement range, so that an overlapped measurement interval exists between the combinations, and when the calculated flow exceeds the measurement range of the gear, gear switching is executed, and frequent gear switching is avoided; in order to improve the measurement accuracy, a calibrated instrument coefficient is arranged under each combination; in order to reduce the calibration complexity, influence factors of independently changing the amplification factor and the exciting current are calibrated respectively by carrying out influence factor decomposition on the amplification factor and the exciting current, and further, all combined instrument coefficients are obtained.

The method can also reduce the calibration times, and the saved calibration times are m × n- (m + n-1), wherein m and n are the gear numbers of the exciting current and the amplification factor respectively.

The invention has the beneficial effects that: the controllable current source and the controllable gain amplifier are combined for use, and the amplification factor of the operational amplifier circuit and the excitation current of the sensor are respectively and automatically regulated and controlled; the controllable current source controls exciting current, the controllable gain amplifier controls amplification factor, and proper amplification factor and proper exciting current are automatically selected according to the current calculated flow value, so that the measuring range of the flowmeter is wider, the measuring range ratio is larger, and the application range is wider. When the flow rate becomes low or high, the final output voltage of the operational amplifier circuit is ensured not to be distorted and to be as large as possible by adjusting the amplification factor, and the output voltage meets the voltage requirement acquired by the ADC; when the flow rate is very low or very high, the output voltage of the sensor is changed by adjusting the exciting current, so that the amplified voltage of the sensor after passing through the amplifying circuit meets the voltage requirement acquired by the ADC. By using the two methods together, the range ratio can be greatly widened.

By using the controllable gain amplifier, the flowmeter can be calibrated by using a smaller flow device, and the problem that the flow range of the device cannot meet the measurement range in the traditional calibration method is solved.

The combination of the controllable current source and the controllable gain amplifier is used, so that the measuring range of the flowmeter becomes wider, the measuring range ratio is larger, and the application range is wider.

And correcting, storing and automatically calling a plurality of gear parameters to ensure the measurement precision.

The measurement ranges between the gears are overlapped, and frequent gear switching is avoided through a model of the hysteresis comparator.

The operation mode of the controllable circuit can be set to automatic and manual, and parameter correction is convenient in the manual mode.

Different amplification factors and different exciting currents form a combination form of a plurality of traditional flowmeters, and under any combination, the electromagnetic flowmeter works equivalently to an independent traditional electromagnetic flowmeter, seamless switching can be performed, only the same hardware circuit and the same software algorithm are used, the cost is greatly saved, and the use efficiency of resources is improved.

In order to improve the measurement accuracy, each combination condition is calibrated in a traditional manner to obtain a plurality of instrument coefficients. When the electromagnetic flowmeter works under a certain combination, the corresponding meter coefficient is automatically called to carry out flow calculation, so that the output precision of the flowmeter is ensured, and the measuring range of the electromagnetic flowmeter is expanded.

In order to reduce the calibration complexity, two dimensions of the influence of amplification factor (gain) and the influence of excitation current are respectively decomposed to form a matrix flowmeter model. At the moment, the influence factors of the exciting current and the gain are respectively calculated through the fixed amplification factor (gain) or the fixed exciting current, and then the coefficients of all the combinations are deduced, so that the calibration requirement is reduced, and the manpower and the material resources are saved.

Generally, the invention uses a controllable current source and a controllable gain amplifier, so that signals are easier to collect, and then the complexity of calibration is reduced by using a method of influencing factor decomposition, so that the electromagnetic flow converter with the structure has more practical value.

Drawings

FIG. 1 is a block diagram of an electromagnetic flowmeter system;

FIG. 2 is a schematic diagram of a controllable gain amplifier circuit;

FIG. 3 is a magnification circuit diagram with controllable magnification according to an embodiment of the present invention;

FIG. 4 is a conventional constant current excitation circuit diagram (constant current section);

FIG. 5 is a circuit diagram of a controllable constant current excitation circuit according to an embodiment of the present invention;

FIG. 6 is an enlarged partial view of FIG. 5;

FIG. 7 a reference voltage generation circuit;

fig. 8 shows a basic processing mode of a conventional electromagnetic flowmeter acquisition system.

Detailed Description

The present invention is described in further detail below with reference to figures 1-8.

The basic processing mode of the traditional electromagnetic flowmeter acquisition system is shown in fig. 8, the voltage output by the sensor is very weak, so that the pre-amplification circuit usually uses an instrumentation amplifier to process signals, then uses a low-pass filter to filter the signals, removes direct current components through blocking capacitor coupling to obtain low-frequency alternating current signals, amplifies the low-frequency alternating current signals through a amplifier circuit, obtains positive-value signals convenient for ADC processing after level shifting, and then enters the ADC.

The electromagnetic flowmeter aims at the problem that a large range ratio cannot be realized due to the fact that the amplification factor and the exciting current of the traditional electromagnetic flowmeter are fixed values in the using process, namely the problem that large and small flow rates are considered simultaneously is solved. Because the exciting current is fixed, when the flow velocity is lower, the signal output by the sensor is very low and is influenced by noise interference according to the electromagnetic induction law, and the signal output by the amplifying circuit has a large noise contribution value, so that the accuracy of the calculated flow value is poor, even the accuracy of the calculated flow value is beyond the standard. When the flow rate is high, the output signal of the sensor is large, and when the amplification factor is set to be large, the operational amplifier output may be saturated, resulting in signal distortion.

In order to solve the problems, the amplification factor of the operational amplifier circuit and the exciting current of the sensor are respectively and automatically regulated and controlled by the combined use of the controllable gain amplifier and the controllable current source, so that the final output voltage of the operational amplifier circuit is not distorted and is as large as possible, the acquisition condition of an ADC (analog-to-digital converter) is met, and the flowmeter with the wide range ratio is realized.

After the hardware circuit is fixed, the interrelation of different amplification factor (gain) gears is determined, that is, each amplification circuit can correspond to a coefficient by which the output voltage value of the amplification circuit can be converted into the equivalent output voltage value of the fixed amplification factor.

The exciting current influences the output voltage of the sensor, and the exciting current is in direct proportion to the magnetic field intensity through the relation between the exciting current and the magnetic field intensity, and the magnetic field intensity is in direct proportion to the induced voltage according to the electromagnetic induction law, and the induced voltage is in direct proportion to the flow velocity of the fluid, so the exciting current is in direct proportion to the flow velocity of the fluid. That is, different exciting currents have a proportional relationship, and equivalent conversion can be performed by a coefficient.

The invention avoids manual setting by automatically adjusting the amplification factor of the operational amplifier, realizes automatic multi-gear signal amplification and further realizes wide range ratio. The basic principle of the controllable gain amplifier is shown in fig. 2, and different amplification factors can be obtained by controlling the switches from S1 to S3. In order to realize intelligent control, a multiplexing analog switch is usually used to replace switches shown in S1-S3, and the multiplexing analog switch can be directly controlled by a single chip microcomputer, namely, the amplification factor is automatically set by the single chip microcomputer, so that which amplification gear the current operational amplifier circuit is in can be determined, and the calculation of flow is facilitated. The analog switch is various, and has single-path multi-path and multi-path, such as a common 74HC series analog switch and a CD405x series switch.

The acquisition circuit is provided with a plurality of amplification gears, each gear corresponds to a measurement range, the range is limited by the acquisition voltage of the ADC, and when the calculated flow rate (ADC input voltage in one-to-one correspondence) exceeds the range, the gears are automatically switched. The overlapped measuring interval is set between the gears, when the measuring interval is higher than the upper limit of the gear measurement or lower than the lower limit of the gear measurement, the switching is automatically carried out, and due to the fact that the overlapped measuring interval exists between the gears, the working model is similar to a hysteresis comparator, misjudgment and frequent switching caused by factors such as interference can be effectively avoided. The processing has the advantages that the amplitude of the output of the analog amplifying circuit is improved, and the subsequent data acquisition is facilitated.

In order to reduce the influence of power supply ripple, an RC filter is applied to the power supply terminal of the operational amplifier, the basic principle of the rate-controllable amplifier is to change the feedback network, as shown in fig. 3 (the amplification circuit diagram with controllable rate according to an embodiment of the present invention), the feedback network is composed of R38, R39, R40, and R41, pins 9 and 10 of the CD4053BPWR chip are connected to pins of the single chip, and the single chip can obtain different amplification rates by controlling the control levels of the UFB and the UFC.

Fig. 4 shows a constant current circuit generally used in a conventional constant current excitation circuit. In order to change the exciting current, it is usually realized by changing a resistance, but this method is implemented once, the exciting current becomes a fixed value during use, that is, the output range of the sensor is limited. The two ways are realized: 1. and changing a voltage division resistor, 2, and changing a sampling resistor. The essence of changing the voltage dividing resistors (R3, R6 and R7) is to change the reference voltage of the non-inverting input end of the operational amplifier, the essence of changing the sampling resistors (R8, R9 and R10) is to change the feedback voltage (voltage of the inverting input end) of the circuit, and when the voltages of the non-inverting and inverting input ends of the operational amplifier are equal, the voltages reach an equilibrium state. The excitation current is calculated by the following formula:

in order to further improve the range ratio, the exciting circuit is adjusted by using a controllable constant current source, in order to avoid mutual interference, a constant current control circuit of a photoelectric isolation technology is implemented, namely, the constant current control circuit based on PWM control is used, the current is adjusted by changing the duty ratio of PWM, in order to improve the output precision of the exciting current, the exciting current is corrected, and the corrected result is stored. Different from the conventional excitation constant current circuit (fig. 4), the generation mode of Vref is different, the conventional mode is directly obtained by resistance voltage division, and the reference voltage Vref obtained by performing photocoupler isolation, shaping and filtering on pulses in a PWM mode is used in the invention, as shown in fig. 5 (a controllable constant current excitation circuit diagram according to an embodiment of the invention).

The PWM waveform sent by the single chip microcomputer is shaped by the inverter U5 after passing through the high-speed photoelectric coupler U4 to obtain the PWM waveform with controllable amplitude, the amplitude of the waveform is determined by V2, and the resistor R15 is used for carrying out current-limiting protection on the input port of the inverter. The PWM waveform output from the inverter U5 passes through a third-order low-pass filter (where R16 and C8 have the meaning of preventing leakage of ac signals) composed of resistors R16, R17, R18, R21, capacitors C7, C8, C9, and an operational amplifier U6B, to obtain a reference voltage Vref, which is proportional to the PWM duty cycle.

The output current of the H-bridge can be calculated by equation (1). In order to reduce the interference of power supply noise to the operational amplifier U6A, an RC filter (R12 and C6) is added to the positive power port of the operational amplifier U6A, so that the ripple of the positive power end of the operational amplifier U6A is improved, the stability of the output of the operational amplifier U6A is improved, and the precision of the output current is ensured. According to the following characteristic of the operational amplifier U6A, the voltage of the inverting input end is consistent with that of the non-inverting input end, namely the voltages at two ends of the sampling resistors R22, R23 and R24 are consistent with the voltage (Vref) obtained by the pre-filter, according to ohm's law, the current on the sampling resistor is in proportion to the PWM duty ratio, and the output current of the circuit is controlled by adjusting the PWM duty ratio.

For the generation of the V2, the voltage reference chip with high precision can be used for providing, and the voltage reference chip can also be used for providing in a mode of dividing the reference voltage and adding the operational amplifier, in order to obtain the corresponding reference voltage more conveniently, the latter is adopted, namely, the reference chip is used for generating a reference voltage, then the resistance voltage divider is used for obtaining the target voltage, the final reference voltage is obtained after the operational amplifier follower is used, and the use of the follower can improve the driving capability. Fig. 7 (reference voltage generating circuit) supplies a power supply voltage V2 to the inverter U5 in fig. 5. A voltage reference chip U9(TL431 or TL432) of 1.25V or 2.5V is used for obtaining a reference voltage, a reference voltage V2ref is obtained through resistance voltage division, the V2ref is subjected to operational amplifier follow-up, an output voltage V2 is obtained, and a voltage follower is used for reducing output impedance, improving the driving capability of V2, avoiding the influence of voltage fluctuation caused by load change of V2 on a high-level voltage value of a PWM waveform, and also being an important measure for ensuring current output accuracy.

According to the electromagnetic induction law, under different excitation currents, the magnetic field intensity generated on the excitation coil is different, and the signal amplitude output by the sensor is also different. And the magnetic induction intensity is in direct proportion to the excitation current, and a proportionality coefficient can be used for representing the influence of the excitation current on the output signal of the sensor. In order to ensure the accuracy of flow calculation, the multi-gear exciting current is calibrated to obtain the relation between the exciting current and the output of the sensor, thereby realizing the correction of the instrument coefficient.

When the flow rate of the fluid is low, the output voltage of the sensor is extremely small, at the moment, the signal-to-noise ratio is low under the influence of interference noise, and the accuracy of flow calculation is inevitably reduced only by improving the amplification factor of the operational amplifier. Therefore, under the condition that the flow velocity of the fluid is relatively low, the signal voltage output by the sensor is improved in a mode of increasing the exciting current, and the purpose of improving the signal to noise ratio is achieved, so that the measurement precision is ensured. When the flow rate of the fluid is large, the output voltage of the sensor may reach the limit voltage of the ADC after passing through the amplifying circuit (under the condition of minimum amplification factor), and therefore, the output voltage of the sensor is reduced by reducing the exciting current, so as to obtain a larger range ratio.

Specifically, the method comprises the following steps:

in order to accurately control the output of the current, correction parameters of a PWM current control circuit are stored in the flowmeter, and the duty ratio of the target current can be calculated through the correction parameters so as to control the exciting current. In order to ensure the consistency of output flow when the magnification of the amplifier is switched, calibrating coefficients under each magnification to obtain a plurality of coefficients under the magnification, so that when the magnification is switched, calling corresponding coefficients to calculate the flow, hereinafter referred to as magnification coefficients for short, and obtaining the magnification coefficients by changing the magnification of the amplifier under the fixed flow; in order to ensure the flow calculation accuracy under different exciting currents, calibration is performed under each exciting current to obtain proportionality coefficients under a plurality of currents, which are referred to as current coefficients for short.

The correction method of the exciting current comprises the following steps: the exciting coil is replaced by a standard resistor, excitation is controlled to enable the H bridge not to be conducted alternately, the single chip microcomputer gives a PWM wave with a given duty ratio, voltage values at two ends of the standard resistor (divided by the resistor to obtain current) are recorded, and therefore the relation between the exciting current and the PWM duty ratio is obtained. Obtaining linear relation by using a two-point method, namely determining two duty ratios, measuring corresponding voltage values to obtain current values, and assuming that the two duty ratios are a and b respectively, and the measured voltages are V respectively_a、V_bAnd the resistance value of the standard resistor is R.

Duty cycle	Voltage of	Calculating the current
			a	Va	Va/R
b	Vb	Vb/R

Then the current at any duty cycle x is:

the duty ratio x corresponding to any current I is as follows:

storing (a, V) in the correction parameters_a)(b，V_b) And calculating the duty ratio under the specified current according to the formula.

The multiplying power coefficient calibration method comprises the following steps: calibration is carried out under the condition of fixed flow (namely, the output signal of the sensor is fixed, and simulation can be carried out by using a signal generator), and a corresponding ADC value (the ADC value is the digital quantity output by the ADC chip) is obtained and recorded by adjusting the amplification factor. For convenience of explanation, it is assumed that the amplification factors of the respective gears M0, M1, and M2 are respectively, and when the sensor inputs are the same, the corresponding ADC values are ADC0, ADC1, and ADC2, and since the sensor inputs are the same and the ADC values and the amplification circuit output values have a one-to-one correspondence relationship, the ADC values and the operational amplification factors have the following relationships:

let K0 equal to 1Then

Since the ADC value is in a proportional relationship with the flow rate when the magnification is fixed, the ADC values used for calculating the flow rate are matched after the normalization conversion of the respective ADC values by the formula (5) at the time of the shift change, that is, after the conversion by the formula.

The calibration method of the current coefficient comprises the following steps: when switching between multiple levels of current, the output voltage of the same sensor at the same flow rate can be different, and theoretically, the exciting current is in direct proportion to the output signal of the sensor. In order to improve the precision, a proportional relation is obtained by using a calibration method, and the actual flow can be better represented. Under the condition of the same flow speed and the same magnification, changing the exciting current, recording ADC values, and setting the exciting current as Ia, Ib and Ic respectively, and the ADC values as ADC _ a, ADC _ b and ADC _ c respectively, wherein a group of coefficients K inevitably exist due to the fixed flow speed, so that the following formula is established:

ADC_a*Ka＝ADC_b*Kb＝ADC_c*Kc

the current can be switched among a plurality of gears, and the smaller the exciting current is, the smaller the output signal of the sensor is; the magnification can be switched between a plurality of steps, the higher the magnification, the larger the value of the ADC. Assuming that a flowmeter has 3 (adjustable) excitation current steps and 3 (adjustable) magnification steps, and assuming that calibration is performed at the magnification M0 and the excitation current step Ia, the coefficients are (1, K1, K2) (1, Kb, Kc), respectively, other coefficients can be expanded, as shown in the table.

(Ia- > Ib- > Ic M0- > M1- > M2 are enlarged in sequence)

When switching between gears, the values of the ADC can be normalized according to the coefficients in the table, and then the flow calculation can be performed according to the flow relation of the calibration point M0.

When the flowmeter is powered on, the flowmeter is started with default preset exciting current and amplification factor, whether the flowmeter is in an automatic amplification factor mode is detected, if yes, the amplification factor can be automatically switched, whether the flowmeter is in an automatic current control mode is detected, and if yes, the exciting current can be automatically switched.

When the exciting current is maximum and the amplification factor is maximum, namely (Ic M2), the lower limit of the detectable flow is minimum; when the excitation current is minimum and the amplification factor is minimum, that is, (Ia M0), the upper limit of the detectable flow rate is maximum.

When the detection ADC value (corresponding to the flow) is smaller, the adjustment is carried out by increasing the magnification and increasing the exciting current, namely the adjustment is carried out towards the lower right of the chart; when the detected ADC value (which corresponds to the ADC flow rate) is large, the adjustment is made by the equation of reducing the amplification factor and reducing the excitation current, that is, toward the upper left of the graph.

The measurable range of each gear is limited by the fact that the output of the operational amplifier is not distorted and the input voltage of the ADC meets the input requirement (generally smaller than the power supply voltage), and the range with guaranteed precision is limited by the signal-to-noise ratio of the output signal of the sensor, and the smaller the output signal of the sensor, the higher the noise contribution degree is, and the poorer the precision is.

Due to the existence of multiple gears, reasonable distribution can be flexibly carried out on the upper limit and the lower limit of measurement in the gears, when a measured value exceeds the upper limit or the lower limit, switching is carried out, a superposition area exists in the measurement range, the current gear is kept unchanged in the superposition area, and the gear is changed only when the measurement range exceeds the range.

The above embodiments are merely illustrative, and not restrictive, and those skilled in the relevant art can make various changes and modifications without departing from the spirit and scope of the invention, and therefore all equivalent technical solutions also belong to the scope of the invention.

Those not described in detail in this specification are within the skill of the art.