阅读说明：本技术 永磁同步电动机的速度恒定控制和功率恒定控制 (Constant speed control and constant power control of permanent magnet synchronous motor ) 是由 蹇龙飞 高桥敏男 于 2020-11-09 设计创作，主要内容包括：电动机控制器包括：电流控制器,被配置为生成用于驱动永磁同步电动机(PMSM)的控制信号,其中电流控制器被配置为测量PMSM的电压信息和电流信息；功率恒定控制器,被配置为接收电压信息和电流信息,并且基于PMSM的目标功率、以及基于电压信息和电流信息来生成第一目标速度；被配置为生成第二目标速度的第一信号生成器；耦合在功率恒定控制器和电流控制器之间的速度恒定控制器,其中速度恒定控制器被配置为可切换地接收第一目标速度和第二目标速度,并且基于接收到的第一目标速度或接收到的第二目标速度来调节PMSM的电动机速度。(The motor controller includes: a current controller configured to generate a control signal for driving a Permanent Magnet Synchronous Motor (PMSM), wherein the current controller is configured to measure voltage information and current information of the PMSM; a power constant controller configured to receive the voltage information and the current information, and generate a first target speed based on a target power of the PMSM and based on the voltage information and the current information; a first signal generator configured to generate a second target speed; a speed constant controller coupled between the power constant controller and the current controller, wherein the speed constant controller is configured to switchably receive a first target speed and a second target speed, and to adjust a motor speed of the PMSM based on the received first target speed or the received second target speed.)

1. A motor controller configured to drive a Permanent Magnet Synchronous Motor (PMSM) with sensorless Field Oriented Control (FOC), the motor controller comprising:

a current controller configured to generate a control signal for driving the PMSM, wherein the current controller is configured to measure voltage information of the PMSM and current information of the PMSM;

a power constant controller configured to receive the voltage information and the current information and generate a first target speed based on a target power of the PMSM and based on the voltage information and the current information;

a first signal generator configured to generate a second target speed;

a constant speed controller coupled between the constant power controller and the current controller, wherein the constant speed controller is configured to switchably receive the first target speed and the second target speed and adjust a motor speed of the PMSM based on the received first target speed or the received second target speed;

a first switch configured to switchably couple the speed constant controller to the power constant controller in a first switch state to receive the first target speed or to the first signal generator in a second switch state to receive the second target speed; and

a first switch controller configured to control a switching state of the first switch.

2. The motor controller of claim 1, wherein:

the speed constancy controller is configured to generate a first torque current based on the first target speed and a second torque current based on the second target speed, wherein the first torque current corresponds to a first motor torque applied to the PMSM to achieve the first target speed and the second torque current corresponds to a second motor torque applied to the PMSM to achieve the second target speed, and

the current controller is configured to generate the control signal based on the first torque current or the second torque current.

3. The motor controller of claim 1, further comprising:

a second signal generator configured to generate a third target speed;

a second switch configured to switchably couple the speed constant controller to the first switch in a first switch state to receive the first target speed or the second target speed or to the second signal generator in a second switch state to receive the third target speed; and

a second switch controller configured to control a switching state of the second switch,

wherein the speed constant controller is coupled to the second switch and configured to receive the first target speed, the second target speed, or the third target speed based on the switch state of the first switch and the switch state of the second switch.

4. The motor controller of claim 1, wherein:

the voltage information includes a first motor voltage component corresponding to a q-axis of a DQ coordinate system and a second motor voltage component corresponding to a d-axis of the DQ coordinate system, an

The current information includes a first current component corresponding to the q-axis of the DQ coordinate system and a second motor current component corresponding to the d-axis of the DQ coordinate system.

5. The motor controller according to claim 1, wherein the power constant controller comprises:

a power feedback estimator configured to receive the voltage information and the current information from the current controller and to calculate a feedback power of the PMSM based on the voltage information and the current information,

wherein the power constant controller is configured to generate the first target speed based on the target power and the calculated feedback power.

6. The motor controller of claim 5, wherein the power constant controller comprises:

a first scaler configured to receive the feedback power from the power feedback estimator and convert the feedback power into a scaled feedback power,

wherein the power constant controller is configured to generate the first target speed based on the target power and the scaled feedback power.

7. The motor controller of claim 6, wherein the first sealer converts the feedback power to the scaled feedback power by normalizing the feedback power.

8. The motor controller of claim 6, wherein the power constant controller comprises:

a power regulator configured to receive the scaled feedback power and convert the scaled feedback power to an estimated input power,

wherein the power constancy controller is configured to generate the first target speed based on the target power and the estimated input power.

9. The motor controller of claim 6 wherein the estimated input power represents an input power of a power inverter coupled to an output of the current controller.

10. The motor controller of claim 8, wherein the power constant controller comprises:

an error generator configured to receive the target power and the estimated input power and compare the target power and the estimated input power to generate an error signal,

wherein the power constant controller is configured to generate the first target speed based on the error signal.

11. The motor controller of claim 10, wherein the power constant controller comprises:

a proportional-integral (PI) controller configured to receive the error signal and generate the first target speed based on the error signal.

12. The motor controller of claim 1, wherein:

the current controller is configured to measure the motor speed of the PMSM, an

The first switch controller is configured to receive the measured motor speed and control the switch state of the first switch based on the measured motor speed.

13. The motor controller of claim 12, wherein:

the first switch controller is configured to compare the measured motor speed to a motor speed threshold, control the first switch to be in the second switch state on a condition that the measured motor speed is less than the motor speed threshold, and control the first switch to be in the first switch state on a condition that the measured motor speed is equal to or greater than the motor speed threshold.

14. The motor controller of claim 13 wherein the second target speed is greater than a first motor speed threshold and the first target speed is greater than the second target speed.

15. A method of driving a Permanent Magnet Synchronous Motor (PMSM) with sensorless Field Oriented Control (FOC), the method comprising:

generating, by a current controller, a control signal for driving the PMSM;

measuring, by the current controller, voltage information of the PMSM and current information of the PMSM;

generating, by a power constancy controller, a first target speed based on a target power of the PMSM and based on the voltage information and the current information;

generating, by a first signal generator, a second target speed;

switchably receiving, by a speed constant controller, the first target speed and the second target speed, the speed constant controller being coupled between the power constant controller and the current controller;

adjusting, by the constant speed controller, a motor speed of the PMSM based on the received first target speed or the received second target speed;

switchably coupling, by a first switch, the speed constant controller to the power constant controller in a first switch state to receive the first target speed or to the first signal generator in a second switch state to receive the second target speed; and

the switching state of the first switch is controlled by a first switch controller.

16. The method of claim 15, wherein:

the voltage information includes a first motor voltage component corresponding to a q-axis of a DQ coordinate system and a second motor voltage component corresponding to a d-axis of the DQ coordinate system, an

The current information includes a first current component corresponding to the q-axis of the DQ coordinate system and a second motor current component corresponding to the d-axis of the DQ coordinate system.

17. The method of claim 15, further comprising:

calculating, by the power constancy controller, a feedback power of the PMSM based on the voltage information and the current information;

converting, by the power constant controller, the feedback power to a scaled feedback power;

converting, by the power constant controller, the scaled feedback power to an estimated input power;

generating, by the power constant controller, an error signal by comparing the target power and the estimated input power; and

generating, by the power constant controller, the first target speed based on the error signal.

18. The method of claim 15, further comprising:

measuring, by the current controller, the motor speed of the PMSM; and

controlling, by the first switch controller, the switch state of the first switch based on the measured motor speed.

19. The method of claim 18, further comprising:

comparing, by the first switch controller, the measured motor speed to a motor speed threshold;

controlling, by the first switch controller, the first switch in the second switch state on a condition that the measured motor speed is less than the motor speed threshold; and

controlling, by the first switch controller, the first switch to be in the first switch state on a condition that the measured motor speed is equal to or greater than the motor speed threshold.

20. The method of claim 19, wherein the second target speed is greater than a first motor speed threshold and the first target speed is greater than the second target speed.

Technical Field

The present disclosure relates generally to an apparatus and method for implementing constant speed control and constant power control of a Permanent Magnet Synchronous Motor (PMSM), and more particularly to a PMSM utilizing sensorless Field Oriented Control (FOC).

Background

The motor controller uses a motor control algorithm that implements a constant speed control or a constant power control for controlling a Permanent Magnet Synchronous Motor (PMSM). The PMSM may be used in a fan motor such as those used in vacuum cleaners, exhaust fans, fume extraction fans, or other similar applications. However, there are at least four problems.

First, when the air intake has changed, the speed constant control cannot maintain a constant air flow of the fan. For example, the air inlet of a vacuum cleaner may become partially blocked or completely blocked.

Second, because the speed remains constant, the motor power will increase rapidly if the load increases, so the speed constant control cannot protect the battery power of the motor. For example, the load may increase in response to the intake opening being more open than during normal operation.

Third, the frequency width of the general power constant control is generally narrow. As a result, the dynamic response is slow. If a fast start of the motor is required, as in the case of a vacuum cleaner, the constant power control may not be able to meet this requirement because the start speed is too slow.

Fourth, the general power constant control controls only the motor power, which is also the output power of the inverter. However, it does not reflect the input power to protect the battery power.

Accordingly, an improved device having power constant control implemented in a PMSM utilizing sensorless Field Oriented Control (FOC) that can address these issues may be desired.

In addition, problems can arise in the application of vacuum cleaners and other suction devices when the inlet of the vacuum cleaner or suction device is blocked by one or more objects. Blocked inlets will cause many other problems including burning out the motor and circuit board as they will reach high temperatures without an effective cooling air flow. In addition, if there is no airflow, dust, dirt, etc. cannot be sucked.

Accordingly, an improved device with constant power control may be desired with the additional ability to automatically (i.e., without user intervention) remove inlet blockage during operation of the motor.

Disclosure of Invention

One or more embodiments provide a motor controller configured to drive a Permanent Magnet Synchronous Motor (PMSM) with sensorless Field Oriented Control (FOC). The motor controller includes: a current controller configured to generate a control signal for driving the PMSM, wherein the current controller is configured to measure voltage information of the PMSM and current information of the PMSM; a power constant controller configured to receive the voltage information and the current information, and generate a first target speed based on a target power of the PMSM and based on the voltage information and the current information; a first signal generator configured to generate a second target speed; a constant speed controller coupled between the constant power controller and the current controller, wherein the constant speed controller is configured to switchably receive a first target speed and a second target speed, and to adjust a motor speed of the PMSM based on the received first target speed or the received second target speed; a first switch configured to switchably couple the speed constant controller to the power constant controller in a first switch state to receive a first target speed or to the first signal generator in a second switch state to receive a second target speed; and a first switch controller configured to control a switching state of the first switch.

One or more embodiments provide a method of driving a PMSM using a sensorless FOC. The method comprises the following steps: generating a control signal for driving the PMSM by a current controller; measuring voltage information of the PMSM and current information of the PMSM by a current controller; generating, by a power constancy controller, a first target speed based on a target power of the PMSM and based on the voltage information and the current information; generating, by a first signal generator, a second target speed; switchably receiving, by a speed constant controller coupled between the power constant controller and the current controller, a first target speed and a second target speed; adjusting, by the constant speed controller, a motor speed of the PMSM based on the received first target speed or the received second target speed; switchably coupling, by the first switch, the speed constant controller to the power constant controller in a first switch state to receive the first target speed or to the first signal generator in a second switch state to receive the second target speed; the switching state of the first switch is controlled by a first switch controller.

Drawings

Embodiments are described herein with reference to the accompanying drawings.

FIG. 1A is a schematic block diagram illustrating a motor controlled actuator of a power semiconductor device in accordance with one or more embodiments;

fig. 1B is a schematic diagram illustrating a power inverter with single-tap current sensing in accordance with one or more embodiments;

FIG. 2A is a graph illustrating a constant speed control feature in accordance with one or more embodiments;

FIG. 2B is a diagram illustrating power constant control features in accordance with one or more embodiments;

3A-1 and 3A-2 are schematic block diagrams of a motor control algorithm implemented by a motor controller in accordance with one or more embodiments;

FIG. 3B is a schematic block diagram of a fast start logic module included in a motor control algorithm and used to implement a speed constant control mode and a power constant control mode in accordance with one or more embodiments;

FIG. 3C is a schematic block diagram of a fail-safe logic module included in a motor control algorithm and used to implement a fail-safe state during a power-constant control mode in accordance with one or more embodiments;

FIG. 4 illustrates a fitted curve (i.e., a normal speed fitted curve) in which a normal motor speed is fitted to a target power in accordance with one or more embodiments;

FIG. 5 is a diagram illustrating a motor rapid start-up flow in accordance with one or more embodiments as compared to a conventional power constant start-up flow;

fig. 6 is a graph illustrating motor speed and input power of the inverter 1 according to a fail-safe state of the motor controller 6 in accordance with one or more embodiments;

fig. 7 is a flow diagram of a fail-safe logic flow implemented by fail-safe logic module 60 in accordance with one or more embodiments.

Detailed Description

In the following, details are set forth to provide a more thorough explanation of the exemplary embodiments. It will be apparent, however, to one skilled in the art that the embodiments may be practiced without these specific details. In other instances, well-known structures and devices are shown in block diagram form or schematic form, not in detail, in order to avoid obscuring the embodiments. In addition, the features of the different embodiments described below may be combined with each other, unless specifically noted otherwise.

Further, in the following description, equivalent or similar elements having equivalent or similar functionality are denoted by equivalent or similar reference numerals. Since the same or functionally equivalent elements are given the same reference numerals in the drawings, a repetitive description of the elements having the same reference numerals may be omitted. Thus, the descriptions provided for elements having the same or similar reference numbers are interchangeable.

In this regard, directional terminology, such as "top," "bottom," "below," "over," "front," "back," "forward," "backward," etc., may be used with reference to the orientation of the figure(s) being described. Because portions of the embodiments can be positioned in a number of different orientations, the directional terminology is used for purposes of illustration and is in no way limiting. It is to be understood that other embodiments may be utilized and structural or logical changes may be made without departing from the scope defined by the claims. The following detailed description is, therefore, not to be taken in a limiting sense.

It will be understood that when an element is referred to as being "connected" or "coupled" to another element, it can be directly connected or coupled to the other element or intervening elements may be present. In contrast, when an element is referred to as being "directly connected" or "directly coupled" to another element, there are no intervening elements present. Other words used to describe the relationship between elements (e.g., "between," directly between, "" adjacent …, "directly adjacent …," etc.) should be interpreted in a similar manner.

In the embodiments shown in the drawings or described herein, any direct electrical connection or coupling, i.e., any connection or coupling without additional intermediate elements, can also be achieved through indirect connection or coupling, i.e., connection or coupling with one or more additional intermediate elements, and vice versa, provided that the general purpose of the connection or coupling is substantially maintained (e.g., to transmit some signal or to transmit some information). Features from different embodiments may be combined to form further embodiments. For example, unless indicated to the contrary, variations or modifications described with respect to one of the embodiments may also be applicable to the other embodiments.

The term "substantially" may be used herein to describe manufacturing tolerances that are considered industrially acceptable to be small (e.g., within 5%) without departing from aspects of the embodiments described herein.

A sensor may refer to a component that converts a physical quantity to be measured into an electrical signal (e.g., a current signal or a voltage signal). The physical quantity may be, for example, a current or a voltage over a shunt resistor in a single shunt resistor system.

Signal processing circuitry and/or signal conditioning circuitry may receive one or more signals from one or more components and perform signal conditioning or processing thereon. As used herein, signal conditioning refers to manipulating a signal in such a way that the signal meets the requirements for the next stage of further processing. Signal conditioning may include conversion from analog to digital (e.g., via an analog-to-digital converter), amplification, filtering, conversion, biasing, range matching, isolation, and any other process needed to make the signal suitable for processing after conditioning.

Thus, the signal processing circuit may comprise an analog-to-digital converter (ADC) which converts analog signals from one or more sensor elements into digital signals. The signal processing circuit may also include a Digital Signal Processor (DSP) that performs some processing on the digital signal.

Many functions of modern equipment in automotive consumer and industrial applications, such as converting electrical energy and driving electric motors or motors, rely on power semiconductor devices. For example, Insulated Gate Bipolar Transistors (IGBTs), Metal Oxide Semiconductor Field Effect Transistors (MOSFETs), and diodes have been used in a variety of applications, including but not limited to power supplies and switches in power converters, to name a few.

A power semiconductor device typically includes a semiconductor structure configured to conduct a load current along a load current path between two load terminal structures of a device. Furthermore, the load current path may be controlled by means of a control electrode (sometimes referred to as gate electrode). For example, the control electrode may set the power semiconductor device to one of an on-state and an off-state upon receiving a corresponding control signal from, for example, a driver unit. The control signal may be a voltage signal or a current signal having a controlled value.

Power transistors are power semiconductor devices that may be used to drive a load current. For example, an IGBT is turned "on" or "off" by activating and deactivating its gate terminal. Applying a positive input voltage signal across the gate and emitter will keep the device in its "on" state, while making the input gate signal zero or slightly negative will make it "off. There are turn-on and turn-off processes for turning on and off the power transistor. During the turn-on process, a gate driver Integrated Circuit (IC) may be used to provide a (source) gate current (i.e., a turn-on current) to the gate of the power transistor in order to charge the gate to a sufficient voltage to turn on the device. Conversely, during the turn-off process, the gate driver IC is used to draw (sink) the gate current (i.e., the turn-off current) from the gate of the power transistor in order to sufficiently discharge the gate to turn off the device. The current pulses may be output from the gate driver IC as a control signal according to a Pulse Width Modulation (PWM) scheme. Thus, the control signal may be switched between an on current level and an off current level during a PWM period for controlling the power transistor. This in turn charges and discharges the gate voltage to turn the power transistor on and off, respectively.

In particular, the gate of the power transistor is a capacitive load and designates an on current (i.e., gate source current) and an off current (i.e., gate sink current) as initial currents when a switching event is initiated. During the turn-off event, after a small amount of time (less compared to the PWM period) has elapsed, the gate current decreases and reaches a zero value when the gate reaches 0V. During the turn-on event, after a small amount of time (less compared to the PWM period) has elapsed, the gate current decreases and reaches a zero value when the gate reaches 15V.

The transistors may include Insulated Gate Bipolar Transistors (IGBTs) and Metal Oxide Semiconductor Field Effect Transistors (MOSFETs) (e.g., silicon MOSFETs or silicon carbide MOSFETs). Although IGBTs may be used as examples in the following embodiments, it should be understood that MOSFETs may be used instead of IGBTs and vice versa. In this context, when replacing an IGBT with a MOSFET, the collector of the IGBT in any of the examples described herein may be replaced with the drain of the MOSFET, the emitter of the IGBT may be replaced with the source of the MOSFET, and the collector-emitter voltage V of the IGBT_CEThe drain-source voltage V of the MOSFET can be used_DSInstead of this. Thus, any IGBT module can be replaced with a MOSFET module and vice versa.

Certain embodiments described herein relate to, but are not limited to, power semiconductor devices that may be used within a power converter or power supply. Thus, in an embodiment, the power semiconductor device may be configured to carry a load current to be supplied to the load and/or, correspondingly, a load current provided by the power supply. For example, the semiconductor device may include one or more power semiconductor cells, such as monolithically integrated diode cells and/or monolithically integrated transistor cells. Such a diode cell and/or such a transistor cell may be integrated in a power semiconductor module.

In the field of power electronics, power semiconductor devices are generally used which comprise transistors which are suitably connected to form a half bridge. For example, a half bridge may be used to drive an electric motor or a switched mode power supply.

For example, a multi-phase inverter is configured to provide multi-phase power by supplying a multi-phase load (e.g., a three-phase motor). For example, three-phase power involves three symmetrical sine waves that are 120 electrical degrees out of phase with each other. In a symmetrical three-phase power supply system, the three conductors each carry an Alternating Current (AC) of the same frequency and voltage amplitude relative to a common reference, but with a phase difference of one third of the period. Due to the phase difference, the voltage on any one conductor reaches its peak one-third of a period after one of the other conductors and one-third of a period before the remaining conductor. This phase delay delivers constant power to a balanced linear load. A rotating magnetic field can also be generated in the motor.

In a three-phase system feeding a balanced and linear load, the sum of the instantaneous currents of the three conductors is zero. In other words, the current in each conductor is equal in magnitude to the sum of the currents in the other two conductors, but of opposite sign. The return path for current in any phase conductor is the other two phase conductor. The instantaneous current results in a current space vector.

The three-phase inverter includes three inverter legs (leg), one for each of the three phases, and each connected in parallel with each other to a Direct Current (DC) voltage source. Each inverter leg includes a pair of transistors, for example arranged in a half-bridge configuration, for converting DC to AC. In other words, each inverter branch includes two complementary transistors (i.e., a high-side transistor and a low-side transistor) connected in series, and they are turned on and off complementarily to each other to drive the phase load. However, the multi-phase inverter is not limited to three phases and may include two or more than three phases, each phase having inverter branches.

Fig. 1A is a schematic block diagram illustrating a motor controlled actuator 100 of a power semiconductor device in accordance with one or more embodiments. Specifically, the motor-controlled actuator 100 includes a power inverter 1 and an inverter control unit 2. The inverter control unit 2 functions as a motor control unit, and thus may also be referred to as a motor controller or a motor control IC. The motor control unit may be a single IC or may be divided into a microcontroller and gate drivers on two or more ICs.

The motor controlled actuator 100 is also coupled to a three-phase motor M that includes three phases U, V and W. The power inverter 1 is a three-phase voltage generator configured to supply three-phase power by supplying three-phase voltages to drive the motor M. It will further be appreciated that the power inverter 1 and the inverter control unit 2 may be placed on the same circuit board, or on separate circuit boards.

Deviations in both amplitude and phase may result in power and torque losses in the motor M. Thus, the motor control actuator 100 may be configured to monitor and control the amplitude and phase of the voltage supplied to the motor M in real time to ensure that proper current balance is maintained based on a feedback control loop. An open loop motor control unit also exists and may be implemented.

The power inverter 1 for a three-phase motor M includes a switch array of six transistor modules 3u +, 3u-, 3v +, 3v-, 3w +, and 3w- (collectively referred to as transistor modules 3) arranged in complementary pairs. Each complementary pair constitutes one inverter branch supplying phase voltages to the three-phase motor M. Each inverter branch therefore comprises an upper (high-side) transistor module 3 and a lower (low-side) transistor module 3. Each transistor module may include one power transistor and may also include a diode (not shown). Thus, each inverter branch comprises an upper transistor and a lower transistor. Load current paths U, V and W extend from the output of each inverter leg (i.e., the output of each half-bridge) between the complementary transistors and are configured to be coupled to a load, such as motor M. The power inverter 1 is coupled to a DC power source 4 (e.g., a battery or a diode bridge rectifier) and to an inverter control unit 2.

In this example, the inverter control unit 2 includes a motor control circuit and a gate driver circuit for controlling the switch array. In some examples, the inverter control unit 2 may be monolithic, with the motor control circuitry and the gate driver circuitry integrated onto a single die. In other examples, the motor control circuit and the gate driver circuit may be divided into separate ICs. A "monolithic" gate driver is a gate driver on a single silicon chip and can further be made with a specific High Voltage (HV) technology. In addition, the gate driver IC may be integrated on the power inverter 1.

The motor controller performs a motor control function of the motor controlled actuator 100 in real time and transmits a PWM control signal to the gate driver. The motor control function may include controlling a permanent magnet motor or controlling an induction motor, and may be configured as a sensorless control that does not require rotor position sensing, such as is the case with sensor-based control of hall sensors and/or encoder devices. Alternatively, the motor control functions may include a combination of sensor-based control (e.g., used at lower rotor speeds) and sensorless control (e.g., used at higher rotor speeds).

For example, the inverter control unit 2 comprises a controller and driver unit 5, the controller and driver unit 5 comprising a microcontroller unit (MCU)6 as a motor controller, and a gate driver 7, the gate driver 7 for generating driver signals for controlling the transistors of each transistor module 3. Thus, the load current paths U, V and W may be controlled by the controller and driver unit 5 by means of controlling the control electrodes (i.e. gate electrodes) of the transistors 3. For example, when receiving a PWM control signal from the microcontroller, the gate driver IC may set the corresponding transistor to be in one of an on state (i.e., on state) or an off state (i.e., off state).

The gate driver IC may be configured to receive instructions including power transistor control signals from the microcontroller and to switch the respective transistor 3 on or off in accordance with the received instructions and control signals. For example, during the turn-on process of the respective transistor 3, the gate driver IC may be used to provide (source) a gate current to the gate of the respective transistor 3 in order to charge the gate. Conversely, during the turn-off process, the gate driver IC may be used to draw (draw) the gate current from the gate of the transistor 3 in order to discharge the gate.

The inverter control unit 2 or the controller and driver unit 5 itself may include a PWM controller, ADC, DSP and/or clock source (i.e., timer or counter) used in implementing a PWM scheme for controlling the state of each transistor and, ultimately, each phase current provided on the respective load current paths U, V and W.

In particular, the microcontroller 6 of the controller and driver unit 5 may use a motor control algorithm, such as a Field Oriented Control (FOC) algorithm, to provide current control in real time for each phase current output to a multiphase load, such as a multiphase motor. Thus, the field-oriented control loop may be referred to as a current control loop. The motor speed can be further controlled by adding a speed constant control loop over the FOC control that provides speed constant control. Thus, a FOC (i.e., a current control loop) may be considered an inner control loop, while a constant speed control loop may be considered an outer control loop. In addition, motor power, and in turn motor speed, may be further controlled by a constant power control loop over the constant speed control loop. Thus, the power constant control loop may be considered to be the outermost control loop, at least with respect to the current control loop and the speed constant control loop. In other words, the current control loop may be considered an inner control loop, the speed constant control loop may be considered an intermediate control loop, and the power constant control loop may be considered an outer control loop.

The current control loop and the constant speed control loop remain active or enabled at all times during motor control (i.e., during operation of the motor). Likewise, the power constant control loop may remain active or enabled during motor control. However, the power-constant control loop may also be switchably activated/deactivated (enabled/disabled) during motor control. In the example when the power constant control loop is activated, the controller and driver unit 5 is considered to be in the power constant control mode, even if the speed constant control loop is also activated. In the situation when the power constant control loop is deactivated, the controller and driver unit 5 is considered to be in the speed constant control mode.

In some cases, a fourth control loop (e.g., a position control loop), also outside of the constant speed control loop, may be used to control motor position.

For example, during an FOC, motor phase currents should be measured so that the exact rotor position can be determined in real time. To enable determination of motor phase currents, the microcontroller 6 may employ an algorithm using single-shunt current sensing (e.g., Space Vector Modulation (SVM), which is also known as Space Vector Pulse Width Modulation (SVPWM)).

Furthermore, the switches 3 (i.e. transistors) of the power inverter 1 are controlled such that at any time both switches in the same inverter branch will not be conducting, otherwise the dc power supply will be short-circuited. This requirement can be met by complementary operation of the switches 3 in the inverter branch according to the motor control algorithm.

Fig. 1B is a schematic diagram illustrating a power inverter 1 with single-tap current sensing in accordance with one or more embodiments. In particular, the power inverter 1 comprises a shunt resistor Rs placed on the negative DC link of the power inverter 1. Transistor 3_u+、3_u-、3_v+、3_v-、3_w+And 3_w-Represented as switches and the motor M is shown with one winding for each of its phases. Here, UO denotes the line-to-neutral voltage from the bridge midpoint U to the motor neutral point O; UN represents the U-bridge voltage from the bridge midpoint U to the negative bus supply rail N; UV denotes line-to-line voltage from U phase to V phase; VW denotes the line-to-line voltage from the V phase to the W phase; and WV denotes a line-to-line voltage from the W phase to the V phase.

The microcontroller 6 in fig. 1A may receive samples of the current drawn from the shunt resistor Rs and then use an algorithm (i.e., software) to reconstruct the three-phase current in real-time. For example, SVPWM is a vector control based algorithm that requires the sensing of three motor phase currents. The DC link current pulses are sampled at precise time intervals by using a single shunt resistor Rs. The voltage drop across the shunt resistor Rs may be amplified by an operational amplifier inside the inverter control unit 2 and shifted up by, for example, 1.65V. The resulting voltage may be converted by an ADC inside the inverter control unit 2. Based on the actual combination of switches, the SVPWM algorithm is used to reconstruct the three-phase currents of the motor M. The ADC may measure the DC link current during the active vector of the PWM period. Two-phase current measurements are available in each sector. Since the sum of the three winding currents is zero, the third phase current value can be calculated.

SVPWM itself is an algorithm for controlling PWM in real time. It is used for the creation of AC waveforms and can be used to drive a three-phase AC powered motor at variable speeds from a DC power source using a plurality of switching transistors. Although the examples herein are described in the context of a three-phase motor, the examples are not so limited and may be applied to any load scheme.

Additionally, it will be understood that other implementations besides a single shunt resistor may be used for current sensing, and other motor control algorithms may be used to control the load, and the embodiments described herein are not limited thereto.

FIG. 2A is a diagram illustrating a constant speed control feature in accordance with one or more embodiments. Specifically, the graph shows the graph for the air flow (m) with respect to the inlet opening degree (%)³H), motor speed (rpm/s) and current (A), where 0% corresponds to the inlet being closed or blocked and 100% corresponds to the inlet being fully open or unblocked. The constant speed control maintains the motor speed at a constant value regardless of changes in the load, which may change in response to changes in the air intake.

Fig. 2B is a diagram illustrating power constant control features in accordance with one or more embodiments. Specifically, the graph shows the graph for the air flow (m) with respect to the inlet opening degree (%)³H), motor speed (rpm/s), current (A) and workAny value of the rate (W), where 0% corresponds to the inlet being closed or blocked and 100% corresponds to the inlet being fully open or unblocked. The motor power is also the same as the output power of the inverter 1. The constant power control maintains the motor power at a constant value regardless of load changes, which may vary in response to changes in the air intake.

In accordance with one or more embodiments, in applications with fans with constant power control, the airflow does not change much. When the air intake is reduced, the load current will change by a smaller amount as compared to the speed constant control. Here, the power constant controller increases the output current in order to maintain a constant power, then the electronic torque will increase and the motor speed will also increase.

The power constant control of the described embodiment can make the target power as the input power of the inverter. Therefore, the power constant controller controls the input power of the inverter and protects the battery. Further, by combining constant speed control and constant power control, in the manner described herein, the constant speed control can be used to achieve a quick start of the motor when the motor is started, and once the motor reaches a predetermined "normal" speed at a selected target power, the control mode can be switched to constant power control by the motor controller.

3A-1 and 3A-2 are schematic block diagrams of a motor control algorithm 300 in accordance with one or more embodiments. FIG. 3B is a schematic block diagram of a fast start logic module included in the motor control algorithm 300 and used to implement a speed constant control mode and a power constant control mode in accordance with one or more embodiments. Fig. 3C is a schematic block diagram of a fail-safe logic module included in motor control algorithm 300 and used to implement a fail-safe state in a power-constant control mode in accordance with one or more embodiments.

In particular, FIG. 3A consists of two parts, FIG. 3A-1 and FIG. 3A-2, coupled together at respective boundaries (A) to form a complete motor control algorithm 300. The motor control algorithm 300 may be implemented as firmware programmed into the motor controller 6 or by a combination of firmware and circuit components. The motor controller 6 itself may comprise one or more controllers.

In particular, the motor control algorithm 300 includes a power constant control loop 11, a speed constant control loop 12 (i.e., sensorless FOC), a current control loop 13, and a power control enable block 14 implemented by the motor controller 6 shown in FIG. 1A. Thus, the motor controller 6 includes a power controller, a speed controller, and a current controller, which implement their respective control loops. Therefore, the power constant control loop 11 may be used interchangeably with the power controller 11, the speed constant control loop 12 may be used interchangeably with the speed controller 12, the current control loop 13 may be used interchangeably with the current controller 13, and the power control enable block 14 may be used interchangeably with the power constant control enable signal 14 s. The power control enable block 14 is itself a controller device configured to switch the motor controller 6 into one of two operating modes including a power control mode or a speed control mode.

As used herein, Vq and Vd denote the stator q-axis voltage and d-axis voltage of the motor in the dq coordinate system, respectively. That is, Vq is a motor voltage component on the q-axis of the DQ coordinate system, and Vd is a motor voltage component on the d-axis of the DQ coordinate system. Similarly, Iq and Id represent stator q-axis current and d-axis current, respectively, of the motor in the DQ coordinate system. That is, Iq is a motor current component on the q-axis of the DQ coordinate system, and Id is a motor current component on the d-axis of the DQ coordinate system. In addition, each proportional-integral (PI) controller receives a proportional gain KP and an integral gain KI.

PI output ═ KP Δ + KI ═ Δ dt, equation 1,

where Δ is the error or deviation of the actual measured value (PV) from the setpoint value (SP).

Δ ═ SP-PV equation 2.

Sensorless FOC software supports driving two types of Permanent Magnet Synchronous Motors (PMSM), i.e., a constant air gap, surface mounted magnet motor and an internally mounted magnet motor with variable reluctance. The sensorless FOC algorithm structure is illustrated in fig. 3A and follows a cascaded control structure having an outer power constant control loop, an intermediate speed constant control loop, and an inner current control loop, each of which is operative in varying the motor winding voltage to drive the motor to a target power or target speed. If the output of the power control enable signal 14s is, for example, logic high (i.e., "1"), the input TargetSpeed of the speed invariant control loop is connected to the output of the power invariant control loop 11 via the switch 63, and the power invariant control loop implements power invariant control. If the output of the power control enable signal 14s is logic low (i.e., "0"), the input TargetSpeed of the speed constant control loop is connected to an external digital or analog signal 15s (universal asynchronous receiver/transmitter (UART), Variable Speed Pump (VSP), frequency and duty cycle) via a switch 63. In other words, the input of the speed ramp rate SpdRampRate block 21 is switchably connected to one of the two alternative paths via the switch 63 in accordance with the control of the power control enable block 14. An external digital or analog signal 15s is generated by an external signal generator 15, which external signal generator 15 is configured to generate the external digital or analog signal 15s based on one or more input parameters related to setting a target speed of the motor.

The speed controller 12 calculates the motor torque required to follow the target speed (TargetSpeed). TargetSpeed is a variable that sets the target speed of the motor. When the power control enable signal 14s is 1, the target speed (TargetSpeed) is from the output of the power constant control loop 11. When the power control enable signal 14s is 0, the target speed (TargetSpeed) is from an external digital or analog signal 15 s. The target speed is a constant value; the target speed is changed to the ramped boost value SpdRef of the ramp by a speed ramp rate SpdRampRate block 21 in accordance with the speed ramp rate. The error generator 22 receives the SpdRef signal and the actual (measured) motor speed value MotorSpeed (i.e., estimated speed) from a flux estimator and Phase Locked Loop (PLL) unit 43 and generates a speed error ErrSpeed, which is the deviation between the SpdRef signal and the actual (estimated) motor speed.

The PI compensator 23 acts on the error ErrSpeed. The integral term forces the velocity steady state error to zero, while the proportional term improves the high frequency response. The PI compensator gains KP and KI are adjusted depending on the motor and load characteristics to meet the target dynamic performance. The output of the PI compensator 23 is a torque current TrqRef that is capable of maintaining the motor speed SpdRef. The limit function block 24 applies one or more limit functions on the output of the PI compensator 23. For example, the limit function block 24 performs a motor limit function MotorLim on the output of the PI compensator 23 to prevent integral saturation (windup) and to maintain the motor current within the motor maximum current. The limit function block 24 performs a low speed limit function LowSpeedLim on the output of the PI compensator 23 to limit the motor current at a low speed. The limit function block 24 executes a regenerative current limit function RegenLim on the output of the PI compensator 23 to limit the regenerative current of the motor.

The motor current is required to generate the torque current TrqRef when the current loop of the current controller 13 drives the motor current. An Interior Permanent Magnet (IPM) controller 31 is configured to split the torque current TrqRef into IdRef and IqRef based on the difference between Ld and Lq for an interior mounted magnet motor having variable reluctance. For a constant air gap Surface Mount Magnetic (SMM) motor, IqRef equals TrqRef, and IdRef equals 0. IqRef is the current command (i.e., reference current value) on the q-axis. In other words, IqRef is the value of the target current for the Iq current component. Similarly, IdRef is the value of the target current (i.e., the reference current value) for the Id current component. The IPM controller 31 also receives a field weakening current IdFwk, which is limited by block 45 based on FwkCurrRatio. The flux weakening current IdFwk is calculated by the field weakening block 44 based on FwkVoltLvl (which indicates the field weakening level) and Vdq (which is the square root of Vd and Vq), which sets the field weakening level. For both Surface Magnetic Motors (SMM) and Interior Permanent Magnets (IPM), the field-weakening current IdFwk is added to IdRef in IPM controller 31.

The current Iq loop PI compensator 34 acts on the error ErrIq between IqRef and Iq, the current Iq loop PI compensator 34 also being referred to as Iq controller 34. The integral term forces the steady state error to zero, while the proportional term improves the high frequency response. The PI compensator gains KP and KI are adjusted depending on the motor and load characteristics to meet the target dynamic performance. The limit function block 36 applies one or more limit functions on the output of the PI compensator 34 to prevent integral saturation and maintain the inverter output voltage based on VdqLim (limit on Vdq).

Similarly, a current Id loop PI compensator 35 acts on the error ErrId between IdRef and Id, the current Id loop PI compensator 35 also being referred to as Id controller 35. The PI compensator gains KP and KI are adjusted depending on the motor and load characteristics to meet the target dynamic performance, but they are typically the same as the current Iq loop PI 34. The limit function block 37 applies one or more limit functions on the output of the PI compensator 35 to prevent integral saturation and maintain the inverter output voltage based on VdqLim.

The forward vector rotation unit 38 applies forward vector rotation to the current loop output voltages Vd and Vq, and transforms the current loop output voltages Vd and Vq into two-phase AC voltage components V α and V β based on the rotor angle calculated by the flux estimator and PLL unit 43. The space vector pulse width modulator 39 receives the two-phase AC voltage components V α and V β, and generates inverter switching signals (i.e., six-path PWM control signals output from the motor controller 6) based on the V α and V β voltage inputs and SVPWM. Then, the gate driver 7 turns on/off the corresponding power transistor 3 based on the PWM control signal.

The current loop of the current controller 13 calculates the inverter voltage to drive the motor current required to generate the desired torque. The phase current reconstruction circuit 40 reconstructs each of the phase currents Iu, Iv, and Iw using a single-split reconstruction, the phase currents Iu, Iv, and Iw for each respective phase U, V and W. In particular, the phase current reconstruction circuit 40 measures the DC link current in the shunt resistor during the active vector of the PWM period. In each PWM period, there are two different active vectors, and the DC link current in each active vector represents the current on one motor phase. Since the sum of all three winding currents is zero in the balanced condition, the third phase current value can be calculated.

Field Oriented Control (FOC) applies an alpha-beta transformation to the three-phase currents using a Clarke transformation at a Clarke transformation unit 41 to derive alpha and beta currents I alpha and I beta. The FOC also uses vector rotation (i.e., coordinate (cordic) rotation) at the vector rotation unit 42 to transform the motor winding currents using the alpha and beta currents I α and I β into two quasi-DC current components, i.e., an Id current component that boosts or weakens the rotor field, and an Iq current component that generates the motor torque.

Two error generators (e.g., subtractors) 32 and 33 generate error values ErrIq and ErrId, respectively. Specifically, the error generator 32 receives the reference current value IqRef as a set value (SP) from the IPM control block 31 and the Iq current value as an actual measurement value (PV) from the vector rotation unit 42, and generates an error value ErrIq. Similarly, the error generator 33 receives the reference current value IdRef (i.e., the reference current value on the d-axis) as a set value (SP) from the IPM control block 31, and the Id current value as an actual measurement value (PV) from the vector rotation unit 42, and generates an error value ErrId.

Typically, the torque reference current from the speed controller is split into Iqref and Idref by IPM control block 31 based on the difference in motor inductances Ld, Lq. Normally, for SMM motors, IdRef is zero; or for IPM motors, IdRef is a negative value scaled with the torque current TrqRef. Above a certain speed (referred to as the base speed), however, the inverter output voltage becomes limited by the DC bus voltage. In this case, field-weakening controller 44 generates a negative Id, plus Id separately from the torque reference current, to oppose the rotor field reducing the back-wound electromotive force (EMF). This enables operation at higher speeds but lower torque output. The field reduction block 44 is used to adjust the Id current to keep the motor voltage magnitude within the bus voltage limits.

The rotor magnet position estimator includes a flux estimator and a PLL 43. The flux estimator and flux PLL operate to detect rotor position and measure motor speed of the operating motor. The flux is calculated based on the feedback currents (i.e., using the alpha and beta currents I alpha and I beta), the estimated voltages V alpha and V beta (based on the DC bus feedback voltage and modulation index), and the motor parameters (inductance and resistance). The output of the flux estimator represents the rotor flux in the two phasors α - β (stationary orthogonal frame, u phase aligned with α).

The flux estimator and the angle and frequency Phase Locked Loop (PLL) of PLL 43 estimate the flux angle (i.e., the estimated rotor angle) and the motor speed from the rotor flux vector in the alpha-beta component. The vector rotation of the PLL calculates the error between the rotor flux angle and the estimated angle. The PI compensator and integrator of the PLL in the closed loop path force the angle and frequency estimates to track the angle and frequency of the rotor flux. The motor speed is derived from the rotor frequency based on the number of rotor poles.

When driving an Interior Permanent Magnet (IPM) motor, rotor salient poles generate a reluctance torque component to increase torque generated by rotor magnets. When driving a Surface Magnet Motor (SMM), the saliency is zero (Ld ═ Lq), and Id is set to zero for maximum efficiency. In the case of an IPM motor with negative salient poles (Ld < Lq), a negative Id will produce a positive reluctance torque. The most efficient operating point is when the total torque is maximized for a given current magnitude. The most efficient operating points of the Surface Magnet Motor (SMM) and the Interior Permanent Magnet (IPM) are calculated by the IPM control block 31.

It should be understood that the illustrated speed constant controller 12 and current controller 13 illustrate only one example configuration and are not limited thereto. For example, normally, the speed constant controller 12 is configured with a speed control loop that outputs a torque current TrqRef based on the target speed TargetSpeed. In addition, the current controller 13 is configured to calculate voltage information and current information for driving the motor based on the torque current TrqRef output from the speed constant controller 12. Specifically, the current controller 13 determines a stator q-axis voltage Vq and a d-axis voltage Vd, and a stator q-axis current Iq and a d-axis current Id. The voltage and current information Vd, Vq, Iq, and Id are supplied to the power constant controller 11, and specifically to the power feedback estimator 51 of the power constant control loop 11.

The power feedback estimator 51 receives the voltage and current information Vd, Vq, Iq, and Id from the current control loop 13, and calculates the feedback power of the motor M. Specifically, the Vd, Vq, Id, Iq inputs of block 51 are supplied by blocks 42, 36, and 37, respectively. As indicated above, the motor power is also the output power of the inverter 1. Therefore, the feedback power calculated by the power feedback estimator 51 is the motor power, or the output power of the inverter 1 shown in fig. 1A.

The feedback power Pfb is calculated by the following DQ coordinate equation, equation 3:

here, equation 3 is based on the principle of a normal coordinate formula by constant amplitude transformation represented by equation 4:

power u_ai_a+u_bi_b+u_ci_cIn the case of the equation 4,

wherein u is_a,u_b,u_cRepresents instantaneous values of input three-phase voltages of the motor, and_a,i_b,i_crepresenting the instantaneous values of the motor input three-phase currents.

The first power sealer, i.e. the power normalization block 52, scales the feedback power Pfb to derive a scaled (normalized) feedback power value Pscfb. Thus, the first power scaler 52 normalizes the feedback power Pfb, and the power scaler, power normalization block 52 output is represented by the following equation 5:

the scaled (normalized) feedback power value Pscfb is then provided to a second sealer, i.e. a power adjustment block 53, which power adjustment block 53 converges (scales) the scaled feedback power value Pscfb, i.e. the output power, to the input power Inputpower based on an adjustment parameter. The input power value Inputpower is then provided to an error generator (e.g., subtractor) 54. The error generator 54 receives the target power value (TargetPower) as a reference value, and the input power value Inputpower from the second scaler 54, and thereby generates a power error value ErrPower. The power error value ErrPower is the deviation or difference between the target power and the actual (estimated) input power. In other words, if the target power and the actual (estimated) input power are equal, the power error value ErrPower is zero; if the target power and the actual (estimated) input power are not equal, the power error value ErrPower is not zero and therefore represents the difference between the two.

The error generator 54 supplies the power error value ErrPower to the power control loop PI controller 55, and the PI controller 55 generates a target speed (TargetSpeed) based on the power error value ErrPower. The target power TargetPower is from an external signal generator 16, which external signal generator 16 is configured to generate an external digital or analog signal 16s (UART, VSP, frequency and duty cycle) providing the target power TargetPower. The external signal generators 15 and 16 may be separate devices or may be combined into a single device.

The target speed (TargetSpeed) output from the power control loop PI controller 55 is limited by the limit function block 56. For example, when the target speed (TargetSpeed) exceeds a threshold maxseedlim (maxseed), where maxseed is the maximum speed of the motor, then the target speed will be limited to maxseedlim (maximum speed limit of the motor).

The switch 61 is used to achieve a quick start operation using the speed constant control mode before the power constant control mode. In particular, the motor controller 6 will be set in a constant speed control mode in order to perform a quick start operation of the motor, and then the motor controller 6 will switch to a constant power control mode after one or more motor start conditions have been met. The timing and relationship of these two control modes is further illustrated in fig. 5.

In the first step of the fast start (i.e., the speed constant control mode), the motor speed MotorSpeed of the motor, as monitored by the fast start logic module 58, is less than 5% maxseed. Further, the start flag output from the rapid start logic module 58 has been initialized to 0 in response to the motor stop state (e.g., the controller 6 in fig. 1A received a motor stop command or did not receive a start command after power-on). This means that the start flag is set to 0 before the controller 6 enters the motor running state, and the start flag is kept to 0 after the controller 6 enters the running state while the motor speed MotorSpeed is less than 5% maxseed. Accordingly, the fast start logic module 58 receives the measured motor speed MotorSpeed from block 43, compares the motor speed MotorSpeed to a mode threshold (e.g., 5% maxseed), and sets the value of the start flag based on the comparison. The mode threshold represents a switching point of the two control modes and may be configurable based on the respective application.

The enable flag output from the fast enable logic module 58 is a control signal for the switch 61 to switchably control its switching state. As a result of the enable flag being set to 0, the target speed of the speed controller 12 is connected to a signal generation block 57 (10% maxseed) which sets the target speed TargetSpeed of the TargetSpeed signal path to 10% maxseed. It will be appreciated that the fraction or percentage of the maxseed value applied by the signal generation block 57 is configurable based on the respective application.

The TargetSpeed signal path may include a control mode switch 61 for fast start control and/or may include a fail-safe switch 62 for fail-safe control. In other words, the quick start control and the fail-safe control may be implemented independently of each other, or may be implemented in combination as shown. If fail-safe control is not implemented, switch 62, block 59, and block 60 may be removed and the output of switch 61 will be coupled to switch 63. If quick start control is not implemented, the switch 61, block 57 and block 58 may be removed and the switch 62 will be coupled to the block 56 when the fail-safe flag is 0. Accordingly, the TargetSpeed signal path may be varied according to the switching states of the switches to provide TargetSpeed to block 21. The TargetSpeed signal path may also include blocks 57, 58, and 59, each of which generates a value for the target speed TargetSpeed provided to block 21.

Normally, the fail-safe function will only be run after the quick start has been completed. During the quick start, the fail-safe function is not enabled, then the fail-safe flag is equal to 0 and switch 62 is connected to switch 61. After the quick start process, the start flag output from block 58 is set to 1, then the switch 61 is connected to block 56, and the fail-safe function is enabled. Switch 62 will be connected to switch 61 or block 59 depending on the fail-safe state.

In a first step, the initialization power loop integral is also set by the fast start logic module 58 when the start flag is 0. Then, the PI controller 55 is initialized to a normal speed of the target power, which is a motor speed corresponding to the target power of the motor. This means that KI ^ Δ dt is initialized to the normal speed of the target power in the PI 55 controller expression as follows.

PI output ═ KP Δ + KI ═ Δ dt, equation 6,

in the second step of the rapid start (i.e., power-constant control mode), the motor speed as monitored by the rapid start logic module 58 is equal to or greater than 5% maxseed. As a result of this threshold being met, the enable flag output from the fast enable logic module 58 is set to 1. In response to the enable flag being set to 1, the target speed of the speed controller 12 is connected to block 56, block 56 being the output of the power control loop. As a result, the control mode of the motor controller 6 is changed from the speed constant control mode to the power constant control mode.

Fig. 3B is a schematic block diagram of the fast boot logic module 58 according to fig. 3A. The quick start logic module 58 includes a threshold generator 58a, a comparator 58b, and a motor status detector 58 c. The threshold generator 58a sets the mode threshold used by the comparator 58 b. The motor state detector 58c detects the state of the motor, more specifically, detects when the motor is stopped. When the motor is stopped, the motor state detector 58c sets the start flag to 0, which also initializes the PI controller 55.

Therefore, first, the start flag is set to 0 by the motor state detector 58c, so that when the controller 6 enters the running state, the start flag has been initialized to 0. As a result, the switch 61 is connected to the block 57 that realizes the constant speed control, and the integration of the power loop PI 55 is initialized to the normal speed of the target power in this process. For example, the mode threshold may be set based on a preset fraction or percentage of the maximum speed of the motor, maxseed. Comparator 58b receives the mode threshold from threshold generator 58a, and the measured motor speed MotorSpeed from block 43, and performs the comparison. If the measured motor speed MotorSpeed is equal to or greater than the mode threshold, the comparator 58b sets the enable flag to 1. The state of switch 61 is controlled by an enable flag generated by fast enable logic module 58.

The power calculation method utilized by the power controller 11 is affected only by the accuracy of the current sampling derived from the stator q-axis current Iq and the d-axis current Id, and the accuracy of the stator q-axis voltage Vq and the d-axis voltage Vd output from the PI controllers 34 and 35. As a result, the method can work well over all motor speed ranges including the flux reduction zone. However, it does not work well under overmodulation conditions.

Note that the second sealer 53 performs input power adjustment to derive the measured input power from the scaled feedback power Pscfb (i.e. the scaled output power). As shown in fig. 3A, in order to control the input power of the inverter 1 for battery protection, an adjustment parameter is set on a signal path of the feedback power. The adjustment parameters are set according to the following procedure.

First, the adjustment value at the second scaler 53 is set to 1, the target power is changed at a plurality of values from the low power to the maximum power, and the input power value output by the second scaler 53 is recorded in the memory for each value of the changed target power.

Second, a ratio between the input power and the target power is calculated based on the recorded values. The determined ratio is set as an adjustment parameter (i.e., a scaler value) for the second scaler 53. Thus, the adjustment parameter is set such that the scaled output power from the first sealer 52 converges to the input power. In other words, the scaled output power is converted into the estimated input power Inputpower.

After the above steps, the input power of the inverter may be matched to the target power using the error generator 54 and the PI controller 55. The output of the PI controller 45 is a target speed (TargetSpeed) provided by the power controller 11.

The quick start function is now described using the motor control algorithm 300.

Normal motor speed (rpm/s) was tested at some target power point. These target power points are arranged at equal intervals from small to large. For example, the target power point may be a percentage set in 10% increments from 0% to 100%, where 100% represents the maximum target power for the motor. Curve fitting is then performed to build a curve for the target power and normal motor speed. Fig. 4 illustrates a fitted curve (i.e., a normal speed fitted curve) in which the normal motor speed is fitted to the target power.

The equation for fitting the curve is set forth in equation 7 below:

normal speed ═ a × target power ^2+ B × target power + C equation 7

(NormalSpeed＝A*TargetPower^2+B*TargetPower+C)

Where A, B and C are the fitting parameters for fitting the curve.

FIG. 5 is a diagram illustrating a motor rapid start-up flow in accordance with one or more embodiments as compared to a conventional power constant start-up flow. According to the start-up procedure implemented using the motor control algorithm 300, the motor M is first started with constant speed control. That is, the motor controller 6 is initially set to the speed constant control mode at the time of startup (i.e., at the time of turning on the power switch of the motor). The target speed TargetSpeed is set to 10% of the maximum speed maxseed by the signal generator 57. When the motor speed MotorSpeed determined by the flux estimator and PLL unit 43 reaches a predetermined percentage of the maximum speed maxseed, the fast start logic sets the start flag to 1 and then the switch 61 responds by coupling to the block 56 to effect a transition from speed constancy control to power constancy control. The predetermined percentage is set to be greater than zero but less than the percentage or amount of the fast start target speed TargetSpeed (e.g., 10% of the maximum speed maxseed) in the fast start first step. In this example, the predetermined percentage is set to 5%, but is not limited thereto.

The predetermined percentage corresponds to a switching threshold or switching point of the motor speed at which the operation mode of the motor controller 6 is switched from the speed constant control mode to the power constant control mode. Therefore, when the motor speed in this example reaches 5% of the maximum speed, the control mode is switched to the power constant control mode. After start-up, the control mode will remain in the constant power control mode, or during motor operation after the start-up period, the control mode may be disabled to meet other system requirements. The start cycle is defined as increasing from zero motor speed to a normal speed corresponding to the target power.

In the first step of the rapid start process, the integral of the power controller 55 is initialized to the normal speed NormalSpeed at this point using the fitted curve illustrated in fig. 4. Here, the normal speed corresponds to the target power TargetPower. Fig. 5 shows a diagram of a motor rapid start-up procedure according to one or more embodiments compared to a conventional power constant start-up procedure, the normal speed corresponding to the target power _ a being the normal speed _ a.

The target speed, TargetSpeed, for the speed control mode will maintain normal speed while the power control loop is initialized to normal speed. This is because the frequency width of the power controller 55 is smaller than that of the speed controller 23, and the integration of the power controller 55 is affected by the speed controller 23. Therefore, a quick start of the motor can be achieved by using these features.

In view of the above, power constant control is configured to maintain power consumption at a constant level, which may, for example, maintain airflow in a fan application, and may be implemented using currents Iq and Id and voltages Vd and Vq as their feedback inputs. In addition, the measured output power of inverter 1 is adjusted (i.e., converted) to the input power of inverter 1 by power control loop 11 (e.g., scalers 52 and 53), which provides battery-powered protection. Furthermore, during motor start-up, the speed constant control mode may be used first to achieve a faster start-up before switching to the power constant control mode. By checking the motor speed when the inlet of the fan is blocked, it is possible to see in which mode the motor controller 6 is. When the motor speed changes and the inlet of the fan is blocked, the motor controller 6 is in power constant control. When the motor speed is constant and the inlet of the fan is blocked, the motor controller 6 is in a constant speed control mode.

Returning to fig. 3A, the power controller 11 may also include a fail-safe switch 62 controlled by the fail-safe logic module 60. The switch 62 is used to implement a fail-safe function while the motor controller 6 is in the constant power control mode. The fail-safe logic module 60 is configured to receive the measured motor speed MotorSpeed from block 43 and perform a comparison thereon.

The initialized value of the fail-safe flag is 0, so if the measured motor speed MotorSpeed is less than the fail speed threshold vacfaultfeed while the motor is operating in the power constancy control mode, the fail-safe logic module 60 maintains the fail-safe flag at 0 and thus controls the fail-safe switch 62 to be connected to the switch 61, which switch 61 is connected to the block 56 in the power constancy control mode. As a result, the motor controller 6 continues to operate in the constant power control mode.

If the measured motor speed MotorSpeed is equal to or greater than the fault speed threshold vacfaultfaulspeed when the motor is operating in the power constant control mode, the fail-safe logic module 60 sets the fail-safe flag to 1 and thus controls the fail-safe switch 62 to be connected to the signal generation block 59. The signal generation block 59 sets the target speed TargetSpeed to the minimum power speed MinPowerSpeed. As a result, the target speed supplied to the block 21 will be set to the MinPowerSpeed value, and the control mode of the motor controller 6 switches to the speed-constant control mode in accordance with the entry into the fail-safe state. That is, by coupling the switch 61 to the signal generator 59, the motor controller 6 switches out of the power constant control mode and into the speed constant control mode.

The fail-safe logic module 60, and thus the motor controller 6, remains in a fail-safe state until certain conditions are met. For example, a safe operating condition is detected, in response to which the fail-safe logic module 60 reconnects the switch 62 to the switch 61 by setting the fail-safe flag to 0. Alternatively, the unsafe condition persists, and the fail-safe logic module 60 then shuts down the motor.

In the fail-safe state, the fail-safe logic module 60 continues to monitor the measured motor speed MotorSpeed and, based on an evaluation of the measured motor speed MotorSpeed, switchably couples the fail-safe switch 62 to the switch 61 (which is set in its power-invariant control mode in a position coupled to the block 56 and represents the fail-safe power-invariant control mode) or to the signal generator 59 (which represents the fail-safe speed-invariant control mode). The fault speed threshold VacFaultSpeed may be set to, for example, 1.08 normalseed. The minimum power speed MinPowerSpeed may be set to, for example, 0.7 × NormalSpeed. However, these thresholds are configurable based on the application.

In a fail-safe state, the measured motor speed MotorSpeed should be reduced when the fail-safe switch 62 is coupled to the signal generator 59 (i.e., fail-safe flag is 1). If the measured motor speed MotorSpeed reaches (i.e., decreases to) the minimum power speed threshold MinPowerSpeed, the fail-safe logic module 60 sets the fail-safe flag to 0 and connects the fail-safe switch 62 to block 56, block 56 being the output of the power control loop. As a result, the mode of the motor controller 6 is changed from the fail-safe speed constant control mode to the fail-safe power constant control mode.

This process may be repeated one or more times. For example, the measured motor speed MotorSpeed may again start increasing towards the fault speed threshold vacfaultfaulspeed while still in the fail-safe state. If the measured motor speed MotorSpeed meets or exceeds the fail speed threshold vacfaultfaultspeed, the fail-safe logic module 60 enters into the fail-safe speed constant control mode by setting the fail-safe flag to 1, which causes the fail-safe switch 62 to be connected to the signal generator 59.

Fig. 3C is a schematic block diagram of the fail-safe logic module 60 according to fig. 3A. The fail-safe logic module 60 includes a first threshold generator 60a, a second threshold generator 60b, a first comparator 60c, and a second comparator 60 d. The first threshold generator 60a sets a fault speed threshold vacfaultfeeded which is used by the first comparator 60 c. For example, the fault speed threshold vacfaultfaulspeed may be set to be greater than the normal speed of the motor and may be a preset fraction or percentage based on the normal speed of the motor.

The second threshold generator 60b sets the minimum power speed threshold MinPowerSpeed used by the second comparator 60 d. The minimum power speed threshold MinPowerSpeed, for example, may be set to be less than the normal speed of the motor and may be a preset fraction or percentage based on the normal speed of the motor.

Both comparators 60c and 60d receive their respective thresholds and both receive the measured motor speed MotorSpeed from block 43 and both perform the respective comparisons. If the measured motor speed MotorSpeed is equal to or greater than the fault speed threshold vacfaultfaulspeed, the comparator 60c sets the fault-safety flag to 1, causes the motor controller 6 to enter into a fault-safe speed constant control mode, and causes the fault-safety switch 62 to be coupled to the signal generator 59.

If the measured motor speed MotorSpeed is equal to or less than the minimum power speed threshold MinPowerSpeed, the comparator 60d sets the fail-safe flag to 0, causing the motor controller 6 to enter a fail-safe power constant control mode, and causing the fail-safe switch 62 to be coupled to block 56.

As will be described in more detail below, while in the constant power control mode, the fail-safe state enables the vacuum cleaner or other suction apparatus to automatically (i.e., without user intervention) remove the inlet blockage during operation of the motor.

Fig. 6 is a graph illustrating motor speed and input power of the inverter 1 according to a fail-safe state of the motor controller 6 in accordance with one or more embodiments. As noted above, while in the power-constant control mode (i.e., switch 61 is coupled to block 56), the motor controller 6 may repeatedly switch between the fail-safe speed-constant control mode (i.e., switch 62 is coupled to block 59) and the fail-safe power-constant control mode (i.e., switch 62 is coupled to block 56). As described above, the fail-safe logic module 60 monitors the measured motor speed MotorSpeed and controls the switch 62 accordingly.

While in the power-constant control mode, the motor speed is maintained at normal speed NormalSpeed by the power control loop 11 based on the target power, TargetPower. In the event of an inlet blockage, the motor speed will automatically increase. If the motor speed increases to the fault speed threshold VacFaultSpeed, the fail-safe logic module 60 will enter a fail-safe state and couple a switch 62 to the signal generator 59 to reduce the motor speed, which helps prevent temperature increases. If the motor speed decreases to the minimum power speed threshold MinPowerSpeed, the fail-safe logic module 60 will couple the switch 62 back to the limit function block 56 to allow the motor speed to be increased based on the target speed TargetSpeed output by the PI controller 55 and the limit function block 56. If there is still an inlet blockage, the motor speed will automatically increase back to the fault speed threshold VacFaultSpeed again, causing the process to repeat. If the inlet blockage is cleared (i.e., removed), the motor speed will likely not increase back to the fault speed threshold VacFaultSpeed, and the fail-safe logic module 60 may decide to exit the fail-safe state and resume normal operation in the constant power control mode. If the motor speed reaches the fault speed threshold vacfaultfaulspecification a threshold number of consecutive times (e.g., 5 times) while in the fail-safe state, the fail-safe logic module 60 may stop the motor function and stop the motor.

Switching between the fail-safe speed constant control mode and the fail-safe power constant control mode allows the motor speed to change rapidly. As can be seen from the input power of the inverter 1 in fig. 7, the force of the evacuation in the process changes sharply. A change in airflow (i.e., suction) is applied to debris blocking the inlet in order to cause the debris to be released or sucked in by the vacuum as a result of the changed force. If the jam cannot be removed by this process, the motor is stopped after a number of intervals so that the jam can be manually removed and to prevent damage to the motor and/or circuit board.

Fig. 7 is a flow diagram of a fail-safe logic flow implemented by fail-safe logic module 60 in accordance with one or more embodiments. The fail-safe logic flow 700 includes a normal run state 701, a block and speed increase state 702, a block and speed decrease state 703, and a stop state 704, all of which operate in a main power constant control mode of the motor controller 6 during which the switch 61 is coupled to the limit function block 56.

With switch 61 coupled to restriction function block 56 and switch 62 coupled to switch 61, motor controller 6 operates in normal run state 701 until the inlet is blocked. When the inlet is blocked, the normal operation state 701 switches to the blocking and rate increasing state 702. The blocked entry is detected by the fault protection logic module 60 in response to the measured motor speed increasing from the normal speed. For example, the fail-safe logic module 60 may compare the measured motor speed to a predetermined threshold (e.g., 1.01 × NormalSpeed) for detecting a blocked inlet.

If the motor is in the blocking and increasing state 702 via the fail-safe logic module 60 for a predetermined time interval (e.g., six seconds) and the motor speed has not reached vacfaultfeed, the fail-safe logic module 60 returns to the normal run state 701. This may indicate that the inlet blockage is partial or temporary and is not dangerous to the motor or circuit board. In the block and speed increase state 702, the control mode is still master power constant control and the target speed is equal to the output of the PI controller 55 and limit function 56. The fail-safe state has not yet been triggered.

Conversely, if the motor speed reaches VacfaultSpeed in the block and speed up state 702, the fault-and-safety logic module 60 will switch to the block and speed down state 703 and the VacFault count value is incremented by 1. By entering the blocking and derating state 703, the fail-safe logic module 60 has triggered the fail-safe state. Here, the target speed is set to MinPowerSpeed, and the fail-safe speed constant control mode is implemented as a sub-state. When the motor speed decreases to MinPowerSpeed, the fail-safe logic module 60 switches to the block and ramp up state 702 again and reverts to the power-constant control mode.

The blocking and speed increasing state 702 is repeated and the fail-safe state may be exited by the satisfaction of the criteria for switching to the normal running state 701 or may remain in the fail-safe state by switching to the blocking and speed decreasing state 703 after the criteria are met. In the latter case, the block and ramp down state 703 is repeated.

Each time the fail-safe logic module 60 switches to the blocking and de-speeding state 703, the VacFault count value stored in the counter of the fail-safe logic module 60 is incremented by 1 and compared to a count threshold (e.g., the fail value of 5). If the VacFault count value is equal to the count threshold, the fail-safe logic module 60 switches to the stopped state 704 and stops the motor by setting the VacFaultFlag (fail count flag) to 1. On a condition that the VacFault count value is less than the count threshold, the fail-safe logic module 60 monitors the motor speed by comparing the motor speed to MinPowerSpeed and triggers the speed-up state 702 on a condition that the motor speed is equal to or less than MinPowerSpeed, causing the motor speed to increase.

The fail-safe logic module 60 results in the following advantages. The fail-safe state extends the life of the permanent magnet synchronous motor, preventing burnout. With sensorless Field Oriented Control (FOC), the cost of achieving a fail-safe state is low. The constant power control may maintain the airflow when the inlet is not completely blocked. Repeated rapid reductions and increases in motor speed can automatically clear debris.

While various embodiments have been described, it will be apparent to those of ordinary skill in the art that many more embodiments and implementations are possible within the scope of the disclosure. Accordingly, the invention is not to be restricted except in light of the attached claims and their equivalents. In regard to the various functions performed by the above described components or structures (assemblies, devices, circuits, systems, etc.), the terms (including a reference to a "means") used to describe such components are intended to correspond, unless otherwise indicated, to any component or structure which performs the specified function of the described component (i.e., that is functionally equivalent), even though not structurally equivalent to the disclosed structure which performs the function in the herein illustrated exemplary implementations of the invention.

Furthermore, the following claims are hereby incorporated into the detailed description, with each claim standing on its own as a separate example embodiment. Although each claim may stand alone as an independent example embodiment, it is noted that-although a dependent claim may refer in a claim to a specific combination with one or more other claims-other example embodiments may also include a combination of a dependent claim with the subject matter of each other dependent or independent claim. Such combinations are presented herein unless it is stated that the particular combination is not intended. Furthermore, it is intended that features of a claim are also included in any other independent claim, even if that claim is not directly dependent on that independent claim.

It is also noted that the methods disclosed in the specification or claims may be implemented by an apparatus having means for performing each of the respective actions of these methods.

Further, it should be understood that the disclosure of multiple acts or functions disclosed in the specification or claims may not be construed as limited to a particular sequence. Thus, the disclosure of multiple acts or functions will not limit them to a particular order unless such acts or functions are not interchangeable for technical reasons. Further, in some embodiments, a single action may include or may be divided into multiple sub-actions. Unless expressly excluded, such sub-actions can be included and are part of the disclosure of the single action.

Embodiments provided herein may be implemented in hardware or in software, depending on certain implementation requirements. An implementation may be performed using a digital storage medium (e.g., a floppy disk, a DVD, a blu-ray, a CD, a RAM, a ROM, a PROM, an EPROM, an EEPROM or a FLASH memory) having electronically readable control signals stored thereon, which cooperate (or are capable of cooperating) with a programmable computer system such that the corresponding method is performed. Accordingly, the digital storage medium may be computer-readable.

The instructions may be executed by one or more processors, such as one or more Central Processing Units (CPUs), Digital Signal Processors (DSPs), general purpose microprocessors, Application Specific Integrated Circuits (ASICs), field programmable logic arrays (FPGAs), or other equivalent integrated or discrete logic circuitry. Thus, as used herein, the term "processor" refers to any of the foregoing structure or any other structure suitable for implementing the techniques described herein. In addition, in some aspects, the functionality described herein may be provided within dedicated hardware and/or software modules. Also, the techniques may be fully implemented in one or more circuits or logic elements.

Thus, the techniques described in this disclosure may be implemented, at least in part, in hardware, software, firmware, or any combination thereof. For example, various aspects of the described techniques may be implemented within one or more processors, including one or more microprocessors, DSPs, ASICs, or any other equivalent integrated or discrete logic circuitry, as well as any combinations of such components.

The control unit, including hardware, may also perform one or more of the techniques described in this disclosure. Such hardware, software, and firmware may be implemented within the same device or within separate devices to support the various techniques described in this disclosure. The software may be stored on a non-transitory computer readable medium such that the non-transitory computer readable medium comprises program code or program algorithms stored thereon that, when executed, cause a computer program to perform the steps of the method.

Although various exemplary embodiments have been disclosed, it will be apparent to those skilled in the art that various changes and modifications can be made which will achieve some of the advantages of the concepts disclosed herein without departing from the spirit and scope of the invention. It is obvious to those skilled in the art that other components performing the same function may be appropriately replaced. It is to be understood that other embodiments may be utilized and structural or logical changes may be made without departing from the scope of the present invention. It should be mentioned that features explained with reference to a particular figure can be combined with features of other figures, even features in those figures which are not explicitly mentioned. Such modifications to the general inventive concept are intended to be covered by the appended claims and their legal equivalents.