阅读说明：本技术 永磁同步电机单电流传感器预测控制和参数扰动抑制方法 (Single current sensor prediction control and parameter disturbance suppression method for permanent magnet synchronous motor ) 是由 张硕 李雪荣 张承宁 董岳林 于 2020-06-02 设计创作，主要内容包括：本发明提供了一种永磁同步电机单电流传感器预测控制和参数扰动抑制方法,将基于单电流传感器的重构相电流应用于改进无差拍预测控制,降低了控制系统的成本,减少了控制系统的体积,避免了电流传感器之间的差异带来的测量误差。此外,实现了良好的控制特性。针对提出的重构策略,利用改进的滑模控制有效的抑制了电机参数变化带来影响。(The invention provides a single current sensor prediction control and parameter disturbance suppression method for a permanent magnet synchronous motor, which applies reconstructed phase current based on a single current sensor to improve dead-beat prediction control, reduces the cost of a control system, reduces the volume of the control system and avoids measurement errors caused by differences among current sensors. Furthermore, good control characteristics are achieved. Aiming at the proposed reconstruction strategy, the influence brought by the change of the motor parameters is effectively inhibited by utilizing the improved sliding mode control.)

1. The single current sensor predictive control and parameter disturbance suppression method for the permanent magnet synchronous motor is characterized by comprising the following steps: the method specifically comprises the following steps:

acquiring online data, namely acquiring a bus current i, a rotor rotating speed w and a rotor position angle theta of an inverter in real time;

step two, reconstructing three-phase current i by utilizing real-time switching state of the inverter and acquired bus current i of the inverter_a、i_b、i_c；

Establishing mathematical models of the permanent magnet synchronous motor in an alpha-beta coordinate system and a d-q coordinate system, establishing a deadbeat current prediction control model, and improving the deadbeat current prediction control model based on a reconstruction strategy for three-phase current in the second step; calculating reference voltage within the SVPWM output voltage range at the next moment in real time by combining the data collected in the first step and the three-phase current reconstructed in the second step;

and step four, improving sliding mode prediction control based on a reconstruction strategy of a sampling time point and three-phase current.

2. The method of claim 1, wherein: the mathematical model of the permanent magnet synchronous motor in the third step under the alpha-beta coordinate system is as follows:

u_α＝R_si_α+L_spi_α-w_eψ_rsinθ

u_β＝R_si_β+L_spi_β+w_eψ_rcosθ

ψ_α＝L_si_α+ψ_rcosθ

ψ_β＝L_si_β+ψ_rsinθ

T_e＝1.5p_mψ_r(i_βcosθ-i_αsinθ)

in the formula u_α、u_βIs stator voltage under α - β coordinate system i_α、i_βStator current phi of α - β coordinate system_rIs a rotor flux linkage; r_sIs a stator resistor; l is_sIs a stator inductance; w is a_e、w_mThe electrical angular velocity of the rotor and the mechanical angular velocity of the rotor, respectively; theta is a rotor position angle; p is a differential operator; t is_eIs an electromagnetic torque; t is_LIs the load torque; b is a viscosity coefficient; p is a radical of_mThe number of pole pairs of the motor is shown; psi_α、ψ_βIs a stator flux linkage under a α - β coordinate system, t is a time variable, J is a load moment of inertia;

the mathematical model of the permanent magnet synchronous motor under a d-q coordinate system is as follows:

u_d＝R_si_d+pψ_d-w_eψ_q

u_q＝R_si_q+pψ_q+w_eψ_d

ψ_d＝L_di_d+ψ_r

ψ_q＝L_qi_q

T_e＝1.5p_m(ψ_ri_q+(L_d-L_q)i_di_q)

in the formula u_d、u_qIs the stator voltage under a d-q coordinate system; i.e. i_d、i_qIs the stator current under a d-q coordinate system; psi_d、ψ_qIs a stator flux linkage under a d-q coordinate system; l is_d、L_qArmature inductances of d and q axes, respectively;

the established deadbeat current prediction control model comprises the following steps:

i_d-p(k+1)＝i_d(k)×(1-T_s×R_s/L_s)+i_q(k)×T_s×w_e+T_s/L_s×u_d(k)

i_q-p(k+1)＝i_q(k)×(1-T_s×R_s/L_s)-i_d(k)×T_s×w_e+T_s/L_s×u_q(k)-T_s×w_e×ψ_r/L_s

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

u_q-p(k+1)＝L_s/T_s×(i_q-ref-(1-T_s×R_s/L_s)×i_q-p(k+1)+T_s×i_d-p(k+1)×w_e+T_s×w_e/L_s×ψ_r)

in the formula i_d-p(k +1) d-axis predicted current, i, at time k +1_q-p(k +1) is the predicted current of the q-axis at the time k +1, i_d-refFor d-axis reference current, i_q-refFor q-axis reference current, u_d-p(k +1) is the predicted voltage at time k +1, u_q-p(k +1) is the predicted voltage of the q-axis at time k +1, T_sIs a switching cycle;

the improved deadbeat current prediction control model is as follows:

i_d-p(k)＝i_d(k)×(1-(T_s-t₂)×R_s/L_s)+i_q(k)×(T_s-t₂)×w_e+(T_s-t₂)/L_s×u_d(k-1)

i_q-p(k)＝i_q(k)×(1-(T_s-t₂)×R_s/L_s)-i_d(k)×(T_s-t₂)×w_e+(T_s-t₂)/L_s×u_q(k-1)-(T_s-t₂)×w_e×ψ_f/L_s

i_d-p(k+1)＝i_d-p(k)×(1-T_s×R_s/L_s)+i_q-p(k)×T_s×w_e+T_s/L_s×u_d(k)

i_q-p(k+1)＝i_q-p(k)×(1-T_s×R_s/L_s)-i_d-p(k)×T_s×w_e+T_s/L_s×u_q(k)-T_s×w_e×ψ_f/L_s

wherein, t₂The current update time in one switching cycle.

3. The method of claim 2, wherein: the fourth step is specifically as follows:

predicting current and kT at the time of last switching cycle current update_sThe current at the moment of time is,

i_d-p(t₂)＝i_d-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_d(k-1)+i_q-p(k-1)×t₂×w_e

i_q-p(t₂)＝i_q-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_q(k-1)-i_d-p(k-1)×t₂×w_e-t₂×w_e×ψ_f/L_s

i_d-p(k)＝i_d-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_d(k-1)+i_q×(T_s-t₂)×w_e-(T_s-t₂)/L_s×f_d(k-1)-(T_s-t₂)/L_s×u_dsmo(k-1)

i_q-p(k)＝i_q-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_q(k-1)-i_d×(T_s-t₂)×w_e-(T_s-t₂)×w_e×ψ_f/L_s-(T_s-t₂)/L_s×f_q(k-1)-(T_s-t₂)/L_s×u_qsmo(k-1)

in the formula i_d-p(t₂) Is t₂Time d-axis prediction of current i_q-p(t₂) Is t₂Time q-axis prediction of current, i_d-p(k) Predicting the current, i, for the d-axis at time k_q-p(k) Predicting the current, i, for the q-axis at time k_d-p(k-1) d-axis predicted current, i, at time k-1_q-p(k-1) predicted current, u, for the q-axis at time k-1_d(k-1) d-axis applied voltage at the time of k-1, u_q(k-1) applying a voltage to the q-axis at the time of k-1;

wherein the content of the first and second substances,

u_dsmo＝(L_s×λ-R_s)×e₁+k₁×L_s×sign(e₁)

u_qsmo＝(L_s×λ-R_s)×e₃+k₁×L_s×sign(e₃)

f_d(k+1)＝f_d(k)+T_s×g_d×u_dsmo

f_q(k+1)＝f_q(k)+T_s×g_q×u_qsmo

in the formula of U_dsmoIs a d-axis sliding mode control law, U_qsmoSliding form for q axisControl law, f_dD-axis parameter disturbance, fq q-axis parameter disturbance, lambda, k₁、g_d、g_qIs an adjustable parameter; e.g. of the type₁＝i_d-p(t₂)-i_d；e₃＝i_q-p(t₂)-i_q；i_d、i_qAnd performing Clark and Park conversion on the reconstructed three-phase current to obtain the three-phase current.

Technical Field

The invention relates to the technical field of permanent magnet synchronous motor control, in particular to a single current sensor prediction control and parameter disturbance suppression method of a permanent magnet synchronous motor based on a bus current sensor.

Background

The main control method of the permanent magnet synchronous motor at present adopts vector control and direct torque control, and the control methods need to realize closed-loop control and need to acquire three-phase current of the motor for feedback to form a closed loop. Accurate acquisition of three-phase current is particularly important for the control strategies, the traditional three-phase current information acquisition is mainly completed through 2-4 current sensors, and two-phase or three-phase current of a stator is directly acquired to complete control. However, the high-precision current sensor is expensive, the cost of the motor control system is increased, the volume of the electric control system is also increased, and errors are brought to current detection due to the difference between the sensors, so that the further application of the current sensor is limited to a certain extent. Traditional control strategies such as PI control and model prediction control need to depend on system models and motor parameters, when the parameters change, the control effect is rapidly reduced, and the control strategies depending on the motor models cannot meet the requirements of high-performance control of special occasions such as aerospace vehicles and the like, so that the application range of the control strategies is limited. PI control and model predictive control both need to utilize motor parameters to realize the control requirements, have the problem of inherent parameter robustness poor, control performance reduction brought to parameter disturbance, scholars at home and abroad propose some control strategies for suppressing disturbance, common suppression strategies include sliding mode variable structure control, self-adaptive control, fuzzy control, neural network and the like, wherein the sliding mode variable structure control has the characteristics of simple structure, low requirement on model precision, and strong robustness to internal parameter disturbance and external disturbance, and is widely applied to an electric control system.

Disclosure of Invention

In view of this, the invention provides a method for prediction control and parameter disturbance suppression of a single current sensor of a permanent magnet synchronous motor, which specifically includes the following steps:

acquiring online data, namely acquiring a bus current i, a rotor rotating speed w and a rotor position angle theta of an inverter in real time;

step two, utilizing the real-time switch state of the inverter (S)_a、S_b、S_c) Reconstructing three-phase current i with collected bus current i of inverter_a、i_b、i_c；

Establishing mathematical models of the permanent magnet synchronous motor in an alpha-beta coordinate system and a d-q coordinate system, establishing a deadbeat current prediction control model, and improving the deadbeat current prediction control model based on a reconstruction strategy for three-phase current in the second step; calculating reference voltage within the SVPWM output voltage range at the next moment in real time by combining the data collected in the first step and the three-phase current reconstructed in the second step;

and step four, improving sliding mode prediction control based on a reconstruction strategy of a sampling time point and three-phase current.

In the second step, the three-phase current is reconstructed mainly according to the relationship between the bus current and the three-phase current obtained by the on-off state of the inverter, which is shown in the following table:

TABLE 1 relationship between switching state, bus current, and three-phase current

S_a、S_b、S_cThe switching states of a phase, b phase and c phase are represented respectively, 1 represents that the upper switching tube is conducted and the lower switching tube is disconnected; 0 represents that the lower switch tube is conducted and the upper switch tube is disconnected. i.e. i_dcRepresenting the collected bus current i_a、i_b、i_cRepresenting the three-phase current of the stator. And an SVPWM seven-segment modulation mode is adopted, and bus current is collected twice at different voltage vector action moments in the first half period of a switching period. The specific acquisition method comprises the following steps: the current value of the moment when the switch state is changed from 0 to 1 is collected, and the current value of a certain phase (assumed to be i) is obtained according to the corresponding relation in the table by the bus current collected for the first time_a) And obtaining the current value of the other phase (assumed as-i) according to the corresponding relation in the table by the bus current acquired for the second time_c) Then according to i_a+i_b+i_c0 to yield i_b. The second bus current moment (t) is collected₂) And updating the values of the three-phase currents, wherein the updated values of the three-phase currents are kept unchanged until the next updating moment.

Further, the mathematical model of the permanent magnet synchronous motor in the third step under the α - β coordinate system is as follows:

u_α＝R_si_α+L_spi_α-w_eψ_rsinθ

u_β＝R_si_β+L_spi_β+w_eψ_rcosθ

ψ_α＝L_si_α+ψ_rcosθ

ψ_β＝L_si_β+ψ_rsinθ

T_e＝1.5p_mψ_r(i_βcosθ-i_αsinθ)

in the formula u_α、u_βIs stator voltage under α - β coordinate system i_α、i_βStator current phi of α - β coordinate system_rIs a rotor flux linkage; r_sIs a stator resistor; l is_sIs a stator inductance; w is a_e、w_mThe electrical angular velocity of the rotor and the mechanical angular velocity of the rotor, respectively; theta is a rotor position angle; p is a differential operator; t is_eIs an electromagnetic torque; t is_LIs the load torque; b is a viscosity coefficient; p is a radical of_mThe number of pole pairs of the motor is shown; psi_α、ψ_βIs a stator flux linkage under a α - β coordinate system, t is a time variable, J is a load moment of inertia;

the mathematical model of the permanent magnet synchronous motor under a d-q coordinate system is as follows:

u_d＝R_si_d+pψ_d-w_eψ_q

u_q＝R_si_q+pψ_q+w_eψ_d

ψ_d＝L_di_d+ψ_r

ψ_q＝L_qi_q

T_e＝1.5p_m(ψ_ri_q+(L_d-L_q)i_di_q)

in the formula u_d、u_qIs the stator voltage under a d-q coordinate system; i.e. i_d、i_qIs the stator current under a d-q coordinate system; psi_d、ψ_qIs a stator flux linkage under a d-q coordinate system; l is_d、L_qArmature inductances of d and q axes, respectively;

the established deadbeat current prediction control model comprises the following steps:

i_d-p(k+1)＝i_d(k)×(1-T_s×R_s/L_s)+i_q(k)×T_s×w_e+T_s/L_s×u_d(k)

i_q-p(k+1)＝i_q(k)×(1-T_s×R_s/L_s)-i_d(k)×T_s×w_e+T_s/L_s×u_q(k)-T_s×w_e×ψ_r/L_s

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

u_q-p(k+1)＝L_s/T_s×(i_q-ref-(1-T_s×R_s/L_s)×i_q-p(k+1)+T_s×i_d-p(k+1)×w_e+T_s×w_e/L_s×ψ_r)

in the formula i_d-p(k +1) d-axis predicted current, i, at time k +1_q-p(k +1) is the predicted current of the q-axis at the time k +1, i_d-refFor d-axis reference current, i_q-refFor q-axis reference current, u_d-p(k +1) is the predicted voltage at time k +1, u_q-p(k +1) is the predicted voltage of the q-axis at time k +1, T_sIs a switching cycle;

the improved deadbeat current prediction control model is as follows:

i_d-p(k)＝i_d(k)×(1-(T_s-t₂)×R_s/L_s)+i_q(k)×(T_s-t₂)×w_e+(T_s-t₂)/L_s×u_d(k-1)

i_q-p(k)＝i_q(k)×(1-(T_s-t₂)×R_s/L_s)-i_d(k)×(T_s-t₂)×w_e+(T_s-t₂)/L_s×u_q(k-1)-(T_s-t₂)×w_e×ψ_f/L_s

i_d-p(k+1)＝i_d-p(k)×(1-T_s×R_s/L_s)+i_q-p(k)×T_s×w_e+T_s/L_s×u_d(k)

i_q-p(k+1)＝i_q-p(k)×(1-T_s×R_s/L_s)-i_d-p(k)×T_s×w_e+T_s/L_s×u_q(k)-T_s×w_e×ψ_f/L_s

wherein, t₂The current update time in one switching cycle.

Further, the fourth step is specifically:

predicting current and kT at the time of last switching cycle current update_sThe current at the moment of time is,

i_d-p(t₂)＝i_d-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_d(k-1)+i_q-p(k-1)×t₂×w_e

i_q-p(t₂)＝i_q-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_q(k-1)-i_d-p(k-1)×t₂×w_e-t₂×w_e×ψ_f/L_s

i_d-p(k)＝i_d-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_d(k-1)+i_q×(T_s-t₂)×w_e

-(T_s-t₂)/L_s×f_d(k-1)-(T_s-t₂)/L_s×u_dsmo(k-1)

i_q-p(k)＝i_q-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_q(k-1)-i_d×(T_s-t₂)×w_e

-(T_s-t₂)×w_e×ψ_f/L_s-(T_s-t₂)/L_s×f_q(k-1)-(T_s-t₂)/L_s×u_qsmo(k-1)

in the formula i_d-p(t₂) Is t₂Time d-axis prediction of current i_q-p(t₂) Is t₂Time q-axis prediction of current, i_d-p(k) Predicting the current, i, for the d-axis at time k_q-p(k) Predicting the current, i, for the q-axis at time k_d-p(k-1) d-axis predicted current, i, at time k-1_q-p(k-1) predicted current, u, for the q-axis at time k-1_d(k-1) d-axis applied voltage at the time of k-1, u_q(k-1) applying a voltage to the q-axis at the time of k-1;

wherein the content of the first and second substances,

u_dsmo＝(L_s×λ-R_s)×e₁+k₁×L_s×sign(e₁)

u_qsmo＝(L_s×λ-R_s)×e₃+k₁×L_s×sign(e₃)

f_d(k+1)＝f_d(k)+T_s×g_d×u_dsmo

f_q(k+1)＝f_q(k)+T_s×g_q×u_qsmo

in the formula of U_dsmoIs a d-axis sliding mode control law, U_qsmoIs a q-axis sliding mode control law, f_dD-axis parameter disturbance, fq q-axis parameter disturbance, lambda, k₁、g_d、g_qIs an adjustable parameter, wherein e₁＝i_d-p(t₂)-i_d；e₃＝i_q-p(t₂)-i_q；i_d、i_qAnd performing Clark and Park conversion on the reconstructed three-phase current to obtain the three-phase current.

In the sliding mode control process, a means of reconstructing phase current based on a bus current sensor is adopted, so that the three-phase current of the motor can be completely replaced for prediction control; the invention provides improved dead-beat prediction control aiming at the reconstruction strategy, and obtains good control effect; the improved sliding mode prediction control provided by the invention has the characteristic of good disturbance suppression. It can be seen that the method of the present invention has at least the following beneficial effects compared to the prior art:

(1) the method utilizes the single current sensor to reconstruct the phase current, thereby reducing the cost of the control system, reducing the volume of the control system and avoiding the measurement error caused by the difference between the current sensors.

(2) Aiming at the provided reconstruction strategy, the method effectively inhibits the influence caused by the change of the motor parameters by utilizing the improved sliding mode control, so that the system has good control characteristics.

Drawings

FIG. 1 is a block diagram of a system model corresponding to the method of the present invention.

FIG. 2 flux linkage mismatch (R)_f＝1.5R_f0) Improving the d-axis and q-axis current and reference current contrast of the dead-beat prediction control under the condition;

FIG. 3 flux linkage mismatch (R)_f＝1.5R_f0) Comparing d-axis current and q-axis current of the traditional sliding mode control with reference current under the condition;

FIG. 4 flux linkage mismatch (R)_f＝0.5R_f0) Improving the d-axis and q-axis current and reference current contrast of the dead-beat prediction control under the condition;

FIG. 5 flux linkage mismatch (R)_f＝0.5R_f0) The d and q axis currents of the sliding mode control are compared with the reference current.

Detailed Description

The technical solutions of the present invention will be described clearly and completely with reference to the accompanying drawings, and it should be understood that the described embodiments are some, but not all embodiments of the present invention. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.

The method for the predictive control and the parameter disturbance suppression of the single current sensor of the permanent magnet synchronous motor, disclosed by the invention, as shown in figure 1, specifically comprises the following steps:

acquiring online data, namely acquiring a bus current i, a rotor rotating speed w and a rotor position angle theta of an inverter in real time;

step two, utilizing the real-time switch state of the inverter (S)_a、S_b、S_c) Reconstructing three-phase current i with collected bus current i of inverter_a、i_b、i_c；

Establishing mathematical models of the permanent magnet synchronous motor in an alpha-beta coordinate system and a d-q coordinate system, establishing a deadbeat current prediction control model, and improving the deadbeat current prediction control model based on a reconstruction strategy for three-phase current in the second step; calculating reference voltage within the SVPWM output voltage range at the next moment in real time by combining the data collected in the first step and the three-phase current reconstructed in the second step;

and step four, improving sliding mode prediction control based on a reconstruction strategy of a sampling time point and three-phase current.

The conventional deadbeat current prediction control outputs a motor reference voltage at the next time, i.e., u (k +1), according to a voltage vector, i.e., u (k), applied to the motor at the current time and motor parameters. The calculation formula of u (k +1) calculated at the k-th time is as follows:

u_d(k+1)＝(2T_sR_s-2L_s)wi_q(k)-(L_s/T_s+T_sR_sR_s/L_s-2R_s)i_d(k)+T_sL_swwi_d(k)

-T_swu_q(k)-(1-T_sR_s/L_s)u_d(k)+wwT_sψ_f

u_q(k+1)＝L_s/T_s×i_q-ref-(L_s/T_s+T_sR_sR_s/L_s-2R_s)i_q(k)-(2T_sR_s-2L_s)i_d(k)i_q(k)

+T_sL_swwi_q(k)+T_swu_d(k)-(1-T_sR_s/L_s)u_q(k)+w(2ψ_f-T_sR_sψ_f/L_s)

in the formula T_sIs a control period; i.e. i_q-refIs a q-axis reference current.

When the calculated reference voltage exceeds the maximum output voltage limit of the SVPWM, the output reference voltage needs to be adjusted to obtain the reference voltage within the SVPWM output range:

in the formula u_d-j、u_q-jThe calculated stator reference voltage under the d-q coordinate system; u. of_d-x、u_q-xThe reference voltage within the corrected SVPWM output voltage range under the d-q coordinate system is obtained; u shape_dcIs the dc bus voltage.

The invention changes the traditional dead-beat prediction control aiming at the proposed phase current reconstruction strategy, and according to the phase current reconstruction strategy provided by the invention, the current updating time is t₂At the moment, the reconstructed three-phase current is at t₂The error between the time and the three-phase current of the actual motor is the minimum, so the current collected at the time k cannot be mistaken as the current at the time k, and the current at the time k +1 can be directly predicted, and thus certain error is brought, the prediction is inaccurate, and the performance of the motor is reduced. The invention provides an improved deadbeat current prediction controlThe current collected at time k is actually t in the last switching period₂The exact current at the moment, so that t is passed first₂The current at the moment (namely, the current collected at the moment k) is predicted firstly, and then the current at the moment k +1 is predicted according to the predicted current at the moment k. The reference voltage calculated in this way is more accurate, and the performance of the motor is improved.

In the second step, the three-phase current is reconstructed mainly according to the relationship between the bus current and the three-phase current obtained by the on-off state of the inverter, which is shown in the following table:

TABLE 1 relationship between switching state, bus current, and three-phase current

S_a、S_b、S_cThe switching states of a phase, b phase and c phase are represented respectively, 1 represents that the upper switching tube is conducted and the lower switching tube is disconnected; 0 represents that the lower switch tube is conducted and the upper switch tube is disconnected. i.e. i_dcRepresenting the collected bus current i_a、i_b、i_cRepresenting the three-phase current of the stator. And an SVPWM seven-segment modulation mode is adopted, and bus current is collected twice at different voltage vector action moments in the first half period of a switching period. The specific acquisition method comprises the following steps: the current value of the moment when the switch state is changed from 0 to 1 is collected, and the current value of a certain phase (assumed to be i) is obtained according to the corresponding relation in the table by the bus current collected for the first time_a) And obtaining the current value of the other phase (assumed as-i) according to the corresponding relation in the table by the bus current acquired for the second time_c) Then according to i_a+i_b+i_c0 to yield i_b. The second bus current moment (t) is collected₂) And updating the values of the three-phase currents, wherein the updated values of the three-phase currents are kept unchanged until the next updating moment.

In a preferred embodiment of the present invention, the mathematical model of the permanent magnet synchronous motor in step three under the α - β coordinate system is:

u_α＝R_si_α+L_spi_α-w_eψ_rsinθ

u_β＝R_si_β+L_spi_β+w_eψ_rcosθ

ψ_α＝L_si_α+ψ_rcosθ

ψ_β＝L_si_β+ψ_rsinθ

T_e＝1_.5p_mψ_r(i_βcosθ-i_αsinθ)

in the formula u_α、u_βIs stator voltage under α - β coordinate system i_α、i_βStator current phi of α - β coordinate system_rIs a rotor flux linkage; r_sIs a stator resistor; l is_sIs a stator inductance; w is a_e、w_mThe electrical angular velocity of the rotor and the mechanical angular velocity of the rotor, respectively; theta is a rotor position angle; p is a differential operator; t is_eIs an electromagnetic torque; t is_LIs the load torque; b is a viscosity coefficient; p is a radical of_mThe number of pole pairs of the motor is shown; psi_α、ψ_βIs a stator flux linkage under a α - β coordinate system, t is a time variable, J is a load moment of inertia;

the mathematical model of the permanent magnet synchronous motor under a d-q coordinate system is as follows:

u_d＝R_si_d+pψ_d-w_eψ_q

u_q＝R_si_q+pψ_q+w_eψ_d

ψ_d＝L_di_d+ψ_r

ψ_q＝L_qi_q

T_e＝1.5p_m(ψ_ri_q+(L_d-L_q)i_di_q)

in the formula u_d、u_qIs the stator voltage under a d-q coordinate system; i.e. i_d、i_qIs the stator current under a d-q coordinate system; psi_d、ψ_qIs a stator flux linkage under a d-q coordinate system; l is_d、L_qArmature inductances of d and q axes, respectively;

the established deadbeat current prediction control model comprises the following steps:

i_d-p(k+1)＝i_d(k)×(1-T_s×R_s/L_s)+i_q(k)×T_s×w_e+T_s/L_s×u_d(k)

i_q-p(k+1)＝i_q(k)×(1-T_s×R_s/L_s)-i_d(k)×T_s×w_e+T_s/L_s×u_q(k)-T_s×w_e×ψ_r/L_s

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

u_q-p(k+1)＝L_s/T_s×(i_q-ref-(1-T_s×R_s/L_s)×i_q-p(k+1)+T_s×i_d-p(k+1)×w_e+T_s×w_e/L_s×ψ_r)

in the formula i_d-p(k +1) d-axis predicted current, i, at time k +1_q-p(k +1) is the predicted current of the q-axis at the time k +1, i_d-refFor d-axis reference current, i_q-refFor q-axis reference current, u_d-p(k +1) is the predicted voltage at time k +1, u_q-p(k +1) is the predicted voltage of the q-axis at time k +1, T_sIs a switching cycle;

the improved deadbeat current prediction control model is as follows:

i_d-p(k)＝i_d(k)×(1-(T_s-t₂)×R_s/L_s)+i_q(k)×(T_s-t₂)×w_e+(T_s-t₂)/L_s×u_d(k-1)

i_q-p(k)＝i_q(k)×(1-(T_s-t₂)×R_s/L_s)-i_d(k)×(T_s-t₂)×w_e+(T_s-t₂)/L_s×u_q(k-1)-(T_s-t₂)×w_e×ψ_f/L_s

i_d-p(k+1)＝i_d-p(k)×(1-T_s×R_s/L_s)+i_q-p(k)×T_s×w_e+T_s/L_s×u_d(k)

i_q-p(k+1)＝i_q-p(k)×(1-T_s×R_s/L_s)-i_d-p(k)×T_s×w_e+T_s/L_s×u_q(k)-T_s×w_e×ψ_f/L_s

wherein, t₂The current update time in one switching cycle.

The fourth step is specifically as follows: predicting current and kT at the time of last switching cycle current update_sThe current at the moment of time is,

i_d-p(t₂)＝i_d-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_d(k-1)+i_q-p(k-1)×t₂×w_e

i_q-p(t₂)＝i_q-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_q(k-1)-i_d-p(k-1)×t₂×w_e-t₂×w_e×ψ_f/L_s

i_d-p(k)＝i_d-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_d(k-1)+i_q×(T_s-t₂)×w_e

-(T_s-t₂)/L_s×f_d(k-1)-(T_s-t₂)/L_s×u_dsmo(k-1)

i_q-p(k)＝i_q-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_q(k-1)-i_d×(T_s-t₂)×w_e

-(T_s-t₂)×w_e×ψ_f/L_s-(T_s-t₂)/L_s×f_q(k-1)-(T_s-t₂)/L_s×u_qsmo(k-1)

in the formula i_d-p(t₂) Is t₂Time d-axis prediction of current i_q-p(t₂) Is t₂Time q-axis prediction of current, i_d-p(k) Predicting the current, i, for the d-axis at time k_q-p(k) Predicting the current, i, for the q-axis at time k_d-p(k-1) d-axis predicted current, i, at time k-1_q-p(k-1) predicted current, u, for the q-axis at time k-1_d(k-1) d-axis applied voltage at the time of k-1, u_q(k-1) applying a voltage to the q-axis at the time of k-1;

wherein the content of the first and second substances,

u_dsmo＝(L_s×λ-R_s)×e₁+k₁×L_s×sign(e₁)

u_qsmo＝(L_s×λ-R_s)×e₃+k₁×L_s×sign(e₃)

f_d(k+1)＝f_d(k)+T_s×g_d×u_dsmo

f_q(k+1)＝f_q(k)+T_s×g_q×u_qsmo

in the formula of U_dsmoIs a d-axis sliding mode control law, U_qsmoIs a q-axis sliding mode control law, f_dD-axis parameter disturbance, fq q-axis parameter disturbance, lambda, k₁、g_d、g_qIs an adjustable parameter, wherein e₁＝i_d-p(t₂)-i_d；e₃＝i_q-p(t₂)-i_q；i_d、i_qAnd performing Clark and Park conversion on the reconstructed three-phase current to obtain the three-phase current.

It should be understood that, the sequence numbers of the steps in the embodiments of the present invention do not mean the execution sequence, and the execution sequence of each process should be determined by the function and the inherent logic of the process, and should not constitute any limitation on the implementation process of the embodiments of the present invention.

Although embodiments of the present invention have been shown and described, it will be appreciated by those skilled in the art that changes, modifications, substitutions and alterations can be made in these embodiments without departing from the principles and spirit of the invention, the scope of which is defined in the appended claims and their equivalents.