Control device for power conversion device and motor drive system
阅读说明:本技术 用于电力转换装置的控制装置以及电动机驱动系统 (Control device for power conversion device and motor drive system ) 是由 许扬 P·比克塞尔 森藤力 于 2019-09-25 设计创作,主要内容包括:根据实施方式的用于电力转换装置(3)的控制装置(10)包括:控制流过电动机(2)的定子绕组的电流的驱动控制单元(16);计算规定电动机的驱动量的驱动量指令值的驱动量调节单元(14);计算电动机的定子磁通的第一估计值和电动机的转子磁通的第一估计值的磁通观测器(22);计算流过电动机的定子绕组的电流的第一估计值的电流观测器(21);和对电动机的定子磁通的第一估计值进行计算的平均校正单元(30)。驱动量调节单元(14)基于电动机的定子磁通的第一估计值来计算规定电动机的驱动量的控制量。(A control device (10) for a power conversion device (3) according to an embodiment includes: a drive control unit (16) that controls a current flowing through a stator winding of the motor (2); a drive amount adjusting means (14) for calculating a drive amount command value for specifying the drive amount of the motor; a flux observer (22) that calculates a first estimated value of a stator flux of the electric motor and a first estimated value of a rotor flux of the electric motor; a current observer (21) that calculates a first estimated value of a current flowing through a stator winding of the motor; and an average correction unit (30) that calculates a first estimated value of the stator magnetic flux of the motor. A drive amount adjusting unit (14) calculates a control amount that defines a drive amount of the motor based on a first estimated value of a stator magnetic flux of the motor.)
1. A control device of a power conversion device for driving a motor, comprising:
a drive control unit that controls a current that the power conversion device causes to flow through a stator winding of the motor;
a drive amount adjusting unit that calculates a drive amount command value that specifies a drive amount of the motor, and controls the power conversion device based on the calculated drive amount command value;
a flux observer that calculates a first estimated value of a stator flux of the electric motor and a first estimated value of a rotor flux of the electric motor based on at least the calculated drive amount command value;
a current observer that calculates a first estimated value of a current flowing through a stator winding of the motor based on at least the calculated first estimated value of the stator magnetic flux; and
an average correction unit including a first conversion processing unit, a second conversion processing unit, and a third conversion processing unit,
the first conversion processing unit calculates a second estimated value of the stator magnetic flux of the motor with respect to the first estimated value of the stator magnetic flux of the motor according to a first predetermined conversion rule based on a synchronization/excitation angular frequency of the motor,
the second conversion processing unit calculates a second estimated value of the rotor magnetic flux of the electric motor with respect to the first estimated value of the rotor magnetic flux of the electric motor according to a second predetermined conversion rule based on the synchronous/excitation angular frequency of the electric motor,
the third conversion processing unit calculates a second estimated value of the stator current of the motor for the first estimated value of the stator current of the motor according to a third prescribed conversion rule based on the synchronous/excitation angular frequency of the motor, and
the drive amount adjusting means calculates a control amount that defines a drive amount of the motor based on at least a torque command for the motor, the second estimated value of the stator magnetic flux of the motor, the second estimated value of the rotor magnetic flux of the motor, and the second estimated value of the current flowing through the stator winding of the motor.
2. The control device for the power conversion device according to claim 1,
the drive amount adjustment unit includes a flux circle calculation processing unit that determines a radius of a flux circle based on a command value of a rotor magnetic flux of the motor, determines a center position of the flux circle based on at least the first estimated value of a stator magnetic flux of the motor, and adjusts the center position of the flux circle based on the second estimated value of a current flowing through a stator winding of the motor and a resistance value of the stator winding.
3. The control device for the power conversion device according to claim 1,
the drive amount adjusting unit includes a torque line processing unit that calculates a torque estimated value of the motor based on the first estimated value of stator magnetic flux of the motor and the first estimated value of rotor magnetic flux of the motor,
the torque line processing unit calculates a voltage-time product corresponding to a product of an average voltage and a control period based on at least a torque command for the motor, a torque estimation value of the motor, the second estimation value of a stator magnetic flux of the motor, the second estimation value of a rotor magnetic flux of the motor, and a rotation speed of the rotor magnetic flux of the motor.
4. The control device for the power conversion device according to claim 1,
the average correction unit adjusts a correction coefficient according to any one of the first to third predetermined conversion rules, based on the magnitude of the synchronous/excitation angular frequency of the motor.
5. The control device for the power conversion device according to claim 3,
in a calculation cycle that is distributed from a first time point of a discrete-time system as a starting point to a second time point after the first time point, the torque line processing unit calculates the torque estimation value of the motor based on the first estimation value of the stator magnetic flux of the motor and the first estimation value of the rotor magnetic flux of the motor at the first time point.
6. The control device for the power conversion device according to claim 5,
the torque line processing unit calculates a voltage-time product corresponding to a product of the average voltage and the control period based on at least a torque command for the motor acquired at the first time point, a torque estimation value of the motor at the first time point, the second estimation value of stator magnetic flux of the motor, the second estimation value of rotor magnetic flux of the motor, and a rotation speed of rotor magnetic flux of the motor at the first time point.
7. The control device for the power conversion device according to claim 6,
the torque line processing unit calculates the voltage-time product based on a result of coordinate transformation with reference to the second estimated value of the stator magnetic flux of the motor.
8. The control device for the power conversion device according to claim 2,
the magnetic flux circle calculation processing unit determines the center position of the magnetic flux circle based on a result of coordinate transformation with the first estimated value of the stator magnetic flux of the motor as a reference.
9. An electric motor drive system comprising:
a power conversion device that drives the motor; and
a control device for the power conversion device,
the control device includes:
a drive control unit that controls a current that the power conversion device causes to flow through a stator winding of the motor;
a drive amount adjusting unit that calculates a drive amount command value that specifies a drive amount of the motor, and controls the power conversion device based on the calculated drive amount command value;
a flux observer that calculates a first estimated value of a stator flux of the electric motor and a first estimated value of a rotor flux of the electric motor based on at least the calculated drive amount command value;
a current observer that calculates a first estimated value of a current flowing through a stator winding of the motor based on at least the calculated first estimated value of the stator magnetic flux; and
an average correction unit including a first conversion processing unit, a second conversion processing unit, and a third conversion processing unit,
the first conversion processing unit calculates a second estimated value of the stator magnetic flux of the motor with respect to the first estimated value of the stator magnetic flux of the motor according to a first predetermined conversion rule based on a synchronization/excitation angular frequency of the motor,
the second conversion processing unit calculates a second estimated value of the rotor magnetic flux of the electric motor with respect to the first estimated value of the rotor magnetic flux of the electric motor according to a second predetermined conversion rule based on the synchronous/excitation angular frequency of the electric motor,
the third conversion processing unit calculates a second estimated value of the stator current of the motor for the first estimated value of the stator current of the motor according to a third prescribed conversion rule based on the synchronous/excitation angular frequency of the motor, and
the drive amount adjusting means calculates a control amount that defines a drive amount of the motor based on at least a torque command for the motor, the second estimated value of the stator magnetic flux of the motor, the second estimated value of the rotor magnetic flux of the motor, and the second estimated value of the current flowing through the stator winding of the motor.
10. A control device for a power conversion device, comprising:
a drive control unit that controls a current that the power conversion device causes to flow through a stator winding of the motor;
a drive amount adjusting means that calculates a drive amount command value that specifies a drive amount of the motor and controls the power conversion device based on the calculated drive amount command value;
a flux observer that calculates a first estimated value of a stator flux of the electric motor and a first estimated value of a rotor flux of the electric motor based on at least the calculated drive amount command value;
a current observer that calculates a first estimated value of a current flowing through a stator winding of the motor based on at least the calculated first estimated value of the stator magnetic flux; and
an average correction unit that receives the first estimated value of the stator flux of the motor, the first estimated value of the rotor flux of the motor, and the first estimated value of the current flowing through the stator winding of the motor, and calculates a second estimated value of the stator flux of the motor, a second estimated value of the rotor flux of the motor, and a second estimated value of the current flowing through the stator winding of the motor by advancing phases of the first estimated value of the stator flux of the motor, the first estimated value of the rotor flux of the motor, and the first estimated value of the current flowing through the stator winding of the motor by a predetermined angle corresponding to a synchronous/excitation angular frequency of the motor.
11. The control device for the power conversion device according to claim 10,
the predetermined angle corresponding to the synchronization/excitation angular frequency of the motor is specified based on the cycle of the driving amount command value calculated by the driving amount adjustment unit and the synchronization/excitation angular frequency.
12. An electric motor drive system comprising:
a power conversion device; and
a control device for the power conversion device,
the control device includes:
a drive control unit that controls a current that the power conversion device causes to flow through a stator winding of the motor;
a drive amount adjusting means that calculates a drive amount command value that specifies a drive amount of the motor and controls the power conversion device based on the calculated drive amount command value;
a flux observer that calculates a first estimated value of a stator flux of the electric motor and a first estimated value of a rotor flux of the electric motor based on at least the calculated drive amount command value;
a current observer that calculates a first estimated value of a current flowing through a stator winding of the motor based on at least the calculated first estimated value of the stator magnetic flux; and
an average correction unit that receives the first estimated value of the stator flux of the motor, the first estimated value of the rotor flux of the motor, and the first estimated value of the current flowing through the stator winding of the motor, and calculates a second estimated value of the stator flux of the motor, a second estimated value of the rotor flux of the motor, and a second estimated value of the current flowing through the stator winding of the motor by advancing phases of the first estimated value of the stator flux of the motor, the first estimated value of the rotor flux of the motor, and the first estimated value of the current flowing through the stator winding of the motor by a predetermined angle corresponding to a synchronous/excitation angular frequency of the motor.
13. A control device for a power conversion device, comprising:
a drive control unit that controls a current that the power conversion device causes to flow through a stator winding of the motor;
a drive amount adjusting means for calculating a drive amount command value that specifies a drive amount of the motor by feedback control and controlling the power conversion device based on the calculated drive amount command value;
a flux observer that calculates a first estimated value of a stator flux of the electric motor and a first estimated value of a rotor flux of the electric motor based on at least the calculated drive amount command value; and
and an average correction unit that adjusts the first estimated value of the stator magnetic flux of the motor and the first estimated value of the rotor magnetic flux of the motor based on an estimated value of a control state of the motor, and calculates a feedback amount of the feedback control.
14. An electric motor drive system comprising:
a power conversion device; and
a control device for the power conversion device,
the control device includes:
a drive control unit that controls a current that the power conversion device causes to flow through a stator winding of the motor;
a drive amount adjusting means for calculating a drive amount command value that specifies a drive amount of the motor by feedback control and controlling the power conversion device based on the calculated drive amount command value;
a flux observer that calculates a first estimated value of a stator flux of the electric motor and a first estimated value of a rotor flux of the electric motor based on at least the calculated drive amount command value; and
and an average correction unit that adjusts the first estimated value of the stator magnetic flux of the motor and the first estimated value of the rotor magnetic flux of the motor based on an estimated value of a control state of the motor, and calculates a feedback amount of the feedback control.
Technical Field
Embodiments described herein relate generally to a control device for a power conversion device and a motor drive system.
This application is based on and claims priority from us provisional patent application 62/737,125 filed on 27.9.2018 and us
Background
As a control system for directly controlling the magnetic flux and torque of the motor, a deadbeat direct torque and magnetic flux control (DB-DTFC) system is known. In the DB-DTFC system, in the two-dimensional flux space, a range that can be specified for the motor flux is represented by a circle, and an expression of the rate of change of the required torque is represented as a straight line. The control device using the DB-DTFC system controls the power conversion device by generating command values for controlling the magnetic flux and torque of the motor using the relationship between circles and straight lines in the magnetic flux space. In a control device using a DB-DTFC system, it is desirable to control an induction motor with high accuracy.
Reference list
Patent document
[ patent document 1] U.S. Pat. No.9281772
Disclosure of Invention
Technical problem
An object of the present invention is to provide a control device and a motor drive system for a power conversion device to improve the control accuracy of a motor.
Means for solving the problems
According to an embodiment, a control device of a power conversion device for driving an electric motor includes a drive control unit, a driving amount adjustment unit, a magnetic flux observer, a current observer, and an average correction unit. The drive control unit controls a current that the power conversion device causes to flow through a stator winding of the motor. The drive amount adjusting means calculates a drive amount command value that specifies a drive amount of the motor, and controls the power conversion device based on the calculated drive amount command value. The flux observer calculates a first estimated value of a stator flux of the electric motor and a first estimated value of a rotor flux of the electric motor based on at least the calculated drive amount command value. The current observer calculates a first estimated value of a current flowing through a stator winding of the motor based on at least the calculated first estimated value of the stator magnetic flux. The average correction unit includes a first conversion processing unit, a second conversion processing unit, and a third conversion processing unit. The first conversion processing unit calculates a second estimated value of the stator magnetic flux of the motor with respect to the first estimated value of the stator magnetic flux of the motor according to a first predetermined conversion rule based on a synchronization/excitation angular frequency of the motor. The second conversion processing unit calculates a second estimated value of the rotor magnetic flux of the electric motor with respect to the first estimated value of the rotor magnetic flux of the electric motor according to a second predetermined conversion rule based on the synchronous/excitation angular frequency of the electric motor. The third conversion processing unit calculates a second estimated value of the stator current of the motor for the first estimated value of the stator current of the motor according to a third predetermined conversion rule based on the synchronous/excitation angular frequency of the motor. The drive amount adjusting means calculates a control amount that defines a drive amount of the motor based on at least a torque command for the motor, the second estimated value of the stator magnetic flux of the motor, the second estimated value of the rotor magnetic flux of the motor, and the second estimated value of the current flowing through the stator winding of the motor.
Drawings
Fig. 1 is a block diagram showing a motor drive system according to a first embodiment;
fig. 2 is a block diagram showing a current magnetic flux estimation unit according to the first embodiment;
fig. 3 is a diagram showing a prescribed conversion rule according to the first embodiment;
fig. 4 is a diagram showing an advantage of applying a prescribed conversion rule based on the synchronous angular frequency of the motor according to the first embodiment;
FIG. 5 is a timing chart showing DB-DTFC according to the first embodiment;
fig. 6A is a diagram showing voltage/torque control according to the first embodiment.
Fig. 6B is a diagram showing voltage/torque control according to the first embodiment.
Fig. 7 is a block diagram showing a DB-DTFC calculation unit according to the first embodiment;
fig. 8 is a diagram showing voltage/torque control according to a first modification of the first embodiment;
fig. 9 is a block diagram showing a current magnetic flux estimation unit according to a second modification of the first embodiment;
fig. 10 is a block diagram showing a motor drive system according to a second embodiment;
fig. 11 is a block diagram showing a DB-DTFC calculation unit according to the second embodiment;
fig. 12 is a block diagram showing a motor drive system according to a third embodiment;
fig. 13 is a block diagram showing a control apparatus according to an embodiment;
fig. 14 is a diagram showing the evaluation results of the motor drive system according to the first embodiment;
fig. 15 is a diagram showing the evaluation results of the motor drive system according to the second embodiment;
fig. 16 is a diagram showing the evaluation results of the motor drive system according to the comparative example;
fig. 17 is a diagram showing the evaluation results of the step response test of torque.
Fig. 18 is a diagram showing the evaluation result of the current observer.
Fig. 19 is a diagram showing a complex vector variable among variables according to the embodiment;
fig. 20 is a diagram showing scalar variables among variables according to the embodiment;
fig. 21 is a diagram showing scalar variables among variables according to the embodiment; and
fig. 22 is a diagram illustrating an advantage of compensating for a hysteresis amount in calculation of a current observer according to an embodiment.
Detailed Description
Hereinafter, a control device and a motor drive system for a power conversion device according to embodiments will be described with reference to the drawings. The power conversion device and the motor drive system described below supply a prescribed AC power to the motor.
In the following description, the motor drive system according to the embodiment is identified as a discrete-time system model. As variables representing the time history for identifying the calculation period, k, (k +1) and (k +2) will be used. With respect to the time point k as the start point of the calculation period, a future time point of the unit time is represented as a time point (k +1), and a time point of the future time increased by one unit time is represented as a time point (k + 2). Any given time point (third time point) between the time point (k +1) and the time point (k +2) may be expressed as a time point (k + α). Here, α may be a real number of 1 to 2, and 1.5 is a representative value thereof. In this embodiment, in a calculation cycle allocated from a point of time (k +1) (first point of time) as a starting point to a point of time (k +2) (second point of time) after the point of time (k +1), an estimated value of the predicted state at the point of time (k + α) may be used. The calculation period starting from the time point (k +1) is referred to as a current period, the calculation period starting from the time point k is referred to as a previous period, and the period starting from the time point (k +2) is referred to as a next period.
For example, the state quantity based on the operation state of the
(first embodiment)
Next, an example of the configuration of the
The
The
The
The DC input of the
The
The
The
Here, a coordinate system used by the
The control performed by the
The first coordinate system is a three-phase coordinate system. The three-phase coordinate system includes three-phase components based on the voltage of the stator winding of the motor 2 (stator voltage). For example, the stator voltage of the
The second coordinate system is a dqs-axis coordinate system. The dqs-axis coordinate system includes a ds axis and a qs axis that are orthogonal to each other. For example, the three-phase coordinate system and the dqs-axis coordinate system may be arranged on a predetermined plane such that the direction of the qs-axis of the dqs-axis coordinate system coincides with the direction of the u-phase voltage vector of the stator with reference to the origin of the dqs-axis coordinate system. The arithmetic operation of converting the three-phase signal component of the three-phase coordinate system into the two-phase signal component of the ds axis and the qs axis of the dqs axis coordinate system is referred to as "dqs axis conversion". The three-phase signal components are transformed into two-phase signal components of the ds and qs axes according to "dqs-axis transformation". The arithmetic operation of converting the two-phase signal components of the ds axis and the qs axis of the dqs axis coordinate system into the three-phase signal components of the three-phase coordinate system is called "dqs axis inverse transform". According to the "dqs-axis inverse transform", the two-phase signal components of the ds-axis and qs-axis are transformed into three-phase signal components. For example, the origin of the dqs-axis coordinate system is defined based on stator flux.
The third coordinate system is a re-aligned coordinate system. Similar to the second coordinate system (stator-side coordinate system), the realigned coordinate system includes a ds axis and a qs axis that are orthogonal to each other. The arithmetic operation of converting the two-phase signal components of the ds axis and the qs axis of the stator-side coordinate system into two-phase signal components of the ds axis and the qs axis of the realigned coordinate system is called "ras-axis transformation". According to the "ras-axis transformation", the two-phase signal components of the ds-axis and qs-axis of the stator-side coordinate system are transformed into two-phase signal components of the ds-axis and qs-axis of the realigned coordinate system. The arithmetic operation of converting the two-phase signal components of the ds axis and the qs axis of the realigned coordinate system into the two-phase signal components of the ds axis and the qs axis of the stator-side coordinate system is called "ras-axis inverse transformation". According to the "ras-axis inverse transformation", the two-phase signal components of the ds-axis and qs-axis of the realigned coordinate system are converted into two-phase signal components of the ds-axis and qs-axis of the stator-side coordinate system. The realigned coordinate system is used in the DB-DTFC, which will be described later. In this embodiment, the method of defining the direction of the axes of the realigned coordinate system includes two techniques: using a technique based on stator flux; and a technique based on the rotor magnetic flux. Details of the realigned coordinate system will be described later. For example, the origin of the realigned coordinate system is defined based on the stator flux.
In fig. 19 to 21, variables used in equations and drawings illustrating embodiments will be described. Fig. 19 is a diagram illustrating a complex vector variable among variables according to an embodiment. Fig. 20 and 21 are diagrams exemplarily showing scalar variables among the variables according to the embodiment.
For example, in this embodiment, the estimated value of the stator flux in the dqs-axis coordinate system is represented as the stator qds axis flux estimated value λ qds _ s _ est. Here, "λ" represents a magnetic flux. "qds" in the first part of its suffix represents the qs-axis component and the ds-axis component of the dqs-axis coordinate. "s" in the second suffix indicates a stator-side stationary coordinate system (hereinafter referred to as a stator-side coordinate system). The stator qds axis magnetic flux λ qds _ s collectively represents the two-phase component of the dqs axis coordinate. In the above case, the two-phase component includes two components, i.e., the stator qs-axis magnetic flux λ qs _ s and the stator ds-axis magnetic flux λ ds _ s. The stator qs-axis magnetic flux λ qs _ s represents the q-axis component in the stator-side dqs-axis coordinate system of the stator magnetic flux. The stator ds-axis magnetic flux λ ds _ s represents the d-axis component in the stator-side dqs-axis coordinate system of the stator flux. In some cases, the information represented by the two-phase components may be collectively treated as vector values in a complex vector space. "est" of the third suffix part represents the estimate. Information for identifying the time sequence information is written in parentheses after the third section. In addition to those indicated in the third section above, there are command values (com), differential values (points), detection values (det), average values (ave), and the like.
In the following equations and drawings, symbols different from those used in the present specification may be used. For example, the stator qds axis magnetic flux estimated value λ qds _ s _ est may be as shown in equation (1).
[ mathematical formula 1]
The subscript "qds" of "λ" shown in the above-shown formula (1) represents information of two-phase components of the dqs-axis coordinate. The superscript "s" of "λ" represents information of the stator-side coordinate system. In addition, "^" above "λ" represents the estimated value. In addition to the above, the symbol above the character includes "." representing a differential value. Instruction values are denoted in superscript with an "x". The variables representing the complex vector include the magnetic flux λ, voltage V, and current i described above. For further details, please refer to fig. 19 to 21.
Turning now to fig. 1, the
For example, the
The
The speed/
For example, the speed/
For example, the speed/
It should be noted that any one of the phase and the angular velocity may be input to the above-described motion observer. When a phase sensor is used as the
When a physical sensor such as the
The DB-DTFC calculation unit 14 (shown as DB-DTFC in the drawing) is a controller that controls the
For example, the DB-
When the DB-
The DB-
The first coordinate
The
The second coordinate
For example, the dqs-axis transformation is performed using the following equation. The stator current Ius is calculated based on the stator currents Ivs and Iws. The relationship between the three-phase stator currents iu, Ivs, and Iws obtained by the two-phase transformation and the stator currents Iqs _ s and Ids _ s is represented by the following formula (2). The transformation represented by the following formula (2) is different from the Clarke transformation which is generally used. Note that the dqs-axis inverse transform is an inverse of the transform represented in equation (2).
[ mathematical formula 2]
Ius+Ivs+Iws=0
The slip
The
The current-magnetic-
The
The
For example, the
The stator qds shaft magnetic flux estimated value λ qds _ s _ est (k +1) described above is an example of a first estimated value of the stator magnetic flux. Stator qds axis flux estimate λ qds _ s _ est (k) is an example of a previous cycle estimate of the stator flux. In the previous period calculation process corresponding to the past time from the calculation process of the present period corresponding to the time point (k +1), the
The
For example, the
The
The
The
The
Here, an outline of the
For example, by using the current magnetic
The 1:
The
The current magnetic
For example, by using information on the estimated state quantity of the
A time point different from the sampling time point in the time axis direction may be defined, and information at the defined time point is given to the DB-
Hereinafter, the details thereof will be sequentially described.
Fig. 2 is a block diagram illustrating the current magnetic
The current-magnetic-
The
The calculation block 211 calculates a moving average value using the time-course data of the stator qds shaft magnetic flux estimated value λ qds _ s _ est, thereby smoothing the stator qds shaft magnetic flux estimated value λ qds _ s _ est. For example, the calculation block 211 is a sampler of the
The calculation block 211 calculates a moving time average value λ qds _ s _ ave (k) (simply referred to as a moving time average value λ ave) which is an average value of the stator qds axis magnetic flux estimated value λ qds _ s _ est (k) and the stator qds axis magnetic flux estimated value λ qds _ s _ est (k +1), based on the stator qds axis magnetic flux estimated value λ qds _ s _ est (k) (the previous period estimated value of the stator magnetic flux) and the stator qds axis magnetic flux estimated value (the first estimated value of the stator magnetic flux) stored in the storage unit. The stator qds shaft magnetic flux estimated value λ qds _ s _ est (k) and the stator qds shaft magnetic flux estimated value λ qds _ s _ est (k +1) are examples of estimated values of the rotor magnetic flux of the
As described above, the calculation block 211 preferably uses two samples, i.e., the stator qds axis flux estimate λ qds _ s _ est (k) and the stator qds axis flux estimate λ qds _ s _ est (k +1), for the moving average calculation. The reason why the calculation block 211 uses two samples is to obtain an accurate estimate of the moving time average value λ ave of the physical system without delay. For example, when the stator qds shaft magnetic flux estimated value λ qds _ s _ est (k +1) is used without moving average calculation, this will cause a phase advance. This will lead to a phase lag when the number of samples used for the moving average calculation exceeds two samples. Therefore, the calculation block 211 preferably performs a moving average calculation using two samples. The mathematical model formed by the
Limit cycles may occur in a discrete time system. The limit cycle is a phenomenon in which a periodic oscillation of the output value occurs in synchronization with the sampling frequency fs. The limit cycles tend to have larger amplitudes as the ratio of the sampling frequency fs to the fundamental frequency f1 (fs/f1) decreases. For example, when the calculation period (sampling period ts) increases or when the fundamental frequency f1 increases, the ratio (fs/f1) decreases.
The calculation block 211 has the effect of suppressing a frequency component of half the sampling frequency fs by taking a moving time average of two consecutive samples. Even when the above-described ratio (fs/f1) becomes high and the limit cycle appears in the signal of the sampling target, the calculation block 211 reduces the influence of noise having vibration characteristics by calculating a moving time average.
The calculation block 212 calculates a voltage correction value Vqds _ s _ comp1(k) (simply referred to as a voltage correction value Vcomp1) based on the moving time average value λ ave and the rotor speed ω r. For example, the calculation block 212 calculates the voltage correction value Vcomp1 by multiplying the moving time average value λ ave by a transfer function represented by equation (3) having variables including the rotor resistance Rr, the rotor winding inductance Lr, and the rotor speed ω r.
[ mathematical formula 3]
Equation (3) shown above is an approximate equation in the case where the value of the rotor speed ω r is constant. Alternatively, for example, the value of the rotor speed ω r described above may be replaced with any one of: the rotor angular velocity estimation value ω r _ est (k), the rotor angular velocity estimation value ω r _ est (k +1), a value obtained by performing conversion using a prescribed conversion rule based on any one of the above-described values, and a value defined based on both the values of the rotor angular velocity estimation value ω r _ est (k) and the rotor angular velocity estimation value ω r _ est (k + 1). In the above case, the value of the rotor speed ω r may be updated.
The calculation block 213 is an adder. The calculation block 213 adds the voltage correction value Vcomp1 calculated by the calculation block 212, the stator qds-axis voltage command value Vqds _ s _ com, the calculation result obtained by the
For example, the calculation block 213 may add at least the voltage correction value Vcomp1 calculated by the calculation block 212 and the stator qds axis voltage command value Vqds _ s _ com (k), and set the added sum as a voltage sum value Vqds _ s _ tot (simply referred to as a voltage sum value Vtot). The voltage correction value Vcomp1 calculated by the calculation block 212 is an example of a first value calculated based on the moving time average. The stator qds shaft voltage command value Vqds _ s _ com is an example of a driving amount command value.
The
The
The
[ mathematical formula 4]
The
[ math figure 5]
The description will be continued on the configuration of the
In the following description, the basic range of the
The
The equation of the current observer defined in the continuous-time system is represented in the following equation (6). Here, "p" denotes a differential operator.
[ mathematical formula 6]
The sum Vtot of the term (first term) of the stator voltage and the term (second term) of the stator magnetic flux in the above equation (6) is defined in the following equation (7). Equation (7) can be rewritten into equation (8) by transforming equation (7) into an equation of a discrete time system including average value calculation.
[ math figure 7]
[ mathematical formula 8]
By using the above formula (7), formula (6) is rewritten to formula (9).
[ mathematical formula 9]
By applying Laplace transform (Laplace transform) to equation (9), equation (10) can be obtained. "s" in the formula (10) represents a laplacian operator.
[ mathematical formula 10]
Here, a latch interface having the following equation (11) as an initial condition is applied.
[ mathematical formula 11]
Therefore, the above equation (10) is converted into the following equation (12).
[ mathematical formula 12]
"χ" in the above-mentioned formula (12) is defined as represented in the following formula (13).
[ mathematical formula 13]
Equation (12) in the s-domain is transformed into equation (14) in the time domain according to the inverse laplacian transform.
[ mathematical formula 14]
The expression (14) is converted into an expression (15) of a discrete time system. This equation (15) is an example of an equation representing the basic characteristics of the
[ mathematical formula 15]
The above-described
The
Since the process of moving-averaging the stator qds axis magnetic flux estimated value λ qds _ s _ est (k +1) is included and executed in the approximate calculation process of converting the stator qds axis magnetic flux estimated value λ qds _ s _ est (k +1) into the average value of the discrete-time model, the
A constant suitable for the characteristics of the
Next, the
The first magnetic
The first magnetic
For example, the first magnetic
Coordinate
The
[ mathematical formula 16]
The coordinate
The second magnetic
The second magnetic
For example, the second magnetic
The
The
The
The
The
The
The
The
For a detailed description of The
Since the current-magnetic-
The above-described average correction using the
The first
The second
The third
The above-described
For example, the
The predetermined angle corresponding to the synchronous angular frequency ω e of the
The above-described synchronous angular frequency ω e of the
A prescribed conversion rule based on the synchronous angular frequency ω e according to the implementation will be described with reference to fig. 3 and 4. Fig. 3 is a diagram illustrating a prescribed conversion rule according to an embodiment. In the dqs coordinate system shown in the drawings, a ds axis pointing downward in the drawings and a qs axis orthogonal to the ds axis are illustrated. The qs axis is located at a position rotated (2/π) (radians) counterclockwise from the ds axis relative to the intersection between the ds axis and the qs axis (i.e., the origin of the dqs coordinate system). A number of arrows starting from the origin of the dqs coordinate system are shown. Each of these arrows is a complex vector representing the magnitude and direction of the magnetic flux λ in the dqs coordinate system. For example, when the start (tail) of the complex vector of the magnetic flux is located at the origin of the dqs coordinate system, the magnetic flux sequentially rotates in the following order, for example, in the counterclockwise direction around the origin of the dqs coordinate system: magnetic flux λ (k +1), magnetic flux λ (k +1.5), and magnetic flux λ (k + 2). The magnetic flux λ (k +1) shown in the figure is an example of the stator qds shaft magnetic flux estimate λ qds _ s _ est (k +1), the rotor qds shaft magnetic flux estimate λ qdr _ s _ est (k +1), and the stator qds shaft current estimate Iqds _ s _ est (k + 1).
The magnetic flux λ (k +2) is reached by rotating around the origin (ω e × ts) with the magnetic flux λ (k +1) as the starting point. For example, the magnetic flux λ (k +1.5) is achieved by rotating (ω e × ts)/2 with the magnetic flux λ (k +1) as a starting point. It is desirable that the magnetic flux λ (k +1.5) take a value close to the average of the magnetic flux λ (k +1) and the magnetic flux λ (k + 2). The description given above is the basis of a prescribed conversion rule based on the synchronous angular frequency ω e of the
Meanwhile, although there is a calculation cycle starting from the time point (k +1) in the period from the time point (k +1) to the time point (k +2), there is no calculation cycle starting from a time point between the time point (k +1) and the time point (k + 2). Therefore, in this embodiment, the magnetic flux λ (k +1.5) is estimated based on the magnetic flux λ (k + 1). The magnetic flux λ (k +1.5) can be derived by rotating the magnetic flux λ (k +1) by a predetermined angle in the rotation direction of the rotor.
The following expression (17) represents conversion of the magnetic flux λ (k +1) into rotation (ω e × ts)/2.
[ mathematical formula 17]
Ke represented by the above formula (17) is defined in the following formula (18).
[ mathematical formula 18]
Fig. 4 is a diagram illustrating an advantage of applying a prescribed conversion rule based on the synchronous angular frequency ω e of the
In the DB-DTFC, a torque change rate during one sampling is necessary for torque control. Here, a method of calculating the rate of change of torque based on the actual air gap torque Te _ act will be described.
Although a torque sensor may be used to measure the actual air gap torque Te _ act, the cost increases and the measured torque has a delay of one sampling period. For this reason, instead of using the value measured by the torque sensor, the actual air-gap torque Te _ act may use the estimated value at the time point (k +1) as the value at the time point (k +1), and use the command value at the time point (k +1), which is the command value of the command value at the time point (k +2), as the value at the time point (k + 2).
The reason for this is that according to the deadbeat control, the torque response follows the torque reference in one sampling. The torque at time point (k +1) will be represented by the airgap torque estimation value Te _ est (k +1), and the torque at time point (k +2) will be represented by the airgap torque command value Te _ com (k + 1).
When time passes from time point (k +1) to time point (k +2), the air-gap torque Te changes from the actual air-gap torque Te _ act (k +1) to the air-gap torque Te _ act (k + 2). For simplicity of explanation, it is assumed that the actual air-gap torque Te _ act (k +1) at the time point (k +1) coincides with the air-gap torque estimated value Te _ est (k +1), and the air-gap torque Te at the time point (k +2) coincides with the air-gap torque command value Te _ cmd (k + 1). A curve Te (k) shown in fig. 4 represents the actual air gap torque Te _ act assumed at the time point (k + 1).
In this embodiment, two solutions including the first and second solutions are given as techniques for acquiring the rate of change of the air-gap torque Te. In either of these two solutions, the rate of change of the air-gap torque Te at the time point (k +1) (Te _ dot (k +1)) is approximated to the rate of change of the air-gap torque Te per unit time (Δ Te _ est (k + 1)/ts).
In the first solution, the rate of change of the air-gap torque Te in the period from the time point (k +1) to the time point (k +2) is calculated using the state quantity at the time point (k + 1). In the first solution described above, the rate of change (Te _ dot (k +1)) of the air gap torque Te at the time point (k +1) is defined as the following equation (19).
[ math figure 19]
In the case of the first solution, as shown in equation (19) above, the rate of change of the air gap torque Te (Te _ dot (k +1)) is defined as a function having variables including the air gap torque estimation value Te _ est (k +1), the air gap torque command value Te _ com (k +1), and the sampling period ts. Since the sampling period ts is constant, the above equation can be defined without depending on the state of change of the actual air gap torque Te _ act. With respect to the above equation (19), the rate of change of the air gap torque Te is calculated from the state quantity at the time (k + 1). There is an approximate Euler technique (Euler) associated with the above description. In the approximation technique of euler, the air-gap torque estimation value Te _ est (k +2) is calculated from the slope of the air-gap torque Te at the time point (k + 1). The first solution is different from the so-called euler approximation technique.
In the second solution, the rate of change in torque in the period from the time point (k +1) to the time point (k +2) is calculated using the state quantity at the time point (k + α).
For example, when the torque in the period from the time point (k +1) to the time point (k +2) linearly increases with respect to the elapsed time expressed as a linear function, the line representing the torque is a curve. In this case, it is preferable that the rate of change in torque in the period from the time point (k +1) to the time point (k +2) be calculated as the rate of change at the time point (k +1.5), that is, the center value thereof. Hereinafter, (k +1.5) is referred to as (1+ α).
In the control period from the time point (k +1) to the time point (k +2), the voltage output by the
Here, the waveforms of the magnetic field and the current are sine waves, and instantaneous values thereof change with time. In the case where the state quantities of the magnetic flux and the current at the time point (k +1) are used to estimate the torque change rate in the period from the time point (k +1) to the time point (k +2), the approximation error contained in the torque change rate increases because the value will change after the time point (k + 1).
Therefore, for the magnetic flux and the current, the estimated value of the time point (k + α) is used, instead of the value of the time point (k + 1). This is schematically illustrated in fig. 4. The straight line Tsol1 is a line representing the rate of change of the torque estimated using the state quantity at the time point (k + 1). The straight line Tsol2 is a line indicating the rate of change of torque estimated using the state quantity at the time point (k + α).
In the above-described second solution, as shown in the following equation (20), the rate of change of the air-gap torque Te at the time point (k + α) is defined.
[ mathematical formula 20]
In the following embodiments, the above-described second solution will be taken as an example, and the operation thereof will be described.
Fig. 5 is a timing diagram illustrating a DB-DTFC according to an embodiment. In the timing chart shown in the drawing, time (seconds) is assigned to the horizontal axis, the steps are divided into the first to third steps on the vertical axis, and the process of each step is represented on the vertical axis. The times k, (k +1), and (k +2) respectively represent the times at which the start of the calculation period in the discrete time control is started.
In the timing chart shown in fig. 5, the first step (step 1) of the upper row includes a sampling process, the second step (step 2) of the middle row includes a calculation process, and the third step (step 3) of the lower row includes an output control process.
Hereinafter, the calculation cycle with the start point as the time point (k +1) will be explained as an example.
In the sampling process of the first step (step 1), the speed/
In the first step described above, the
In the second step (step 2), the process is performed based on the result of the sampling process performed in the first step. In the second step, the calculation processes of the current magnetic
First, as described above, the current magnetic
Next, the
Next, the DB-
Next, the first coordinate
In the second step described above, the
In the third step (step 3), an output process of supplying a control signal based on the driving amount command value to the power conversion device is executed by the
The third step is performed based on the detection result obtained in the second step. At the calculation start time point of time point (k +1),
Also after the time point (k +2), a process similar to the calculation cycle of the time point (k +1) is repeated.
Voltage/torque control according to the embodiment will be described with reference to fig. 6A and 6B. Fig. 6A and 6B are diagrams illustrating voltage/torque control according to an embodiment.
Fig. 6A shows an example of a magnetic flux plane of the stator-side coordinate system. Fig. 6B illustrates an example of the flux planes of the realigned coordinate system aligned with respect to the rotor qds shaft flux estimate λ qdr _ s _ est (k + α). In fig. 6A and 6B, the ds axis is directed downward in the drawing, and the qs axis is directed rightward in the drawing. In the realigned coordinate system shown in fig. 6B, the direction of the vector (arrow) of the rotor qds shaft magnetic flux estimate λ qdr _ s _ est (k +1) is aligned with the direction of the ds axis and is shown as rotor qds shaft magnetic flux estimate λ qdr _ ras _ est (k + α). Each flux plane includes a torque line Te (k +2) derived from a physical model of the
First, 6A will be described.
The control variables used by the DB-DTFC include an air gap torque command value Te _ com (k +1) and a stator flux command value λ qds _ s _ com (k + 2). The radius of the magnetic flux circle λ c (k +2) is defined by the magnitude of the stator magnetic flux command value λ qds _ s _ com (k + 2).
Here, in the
The torque line is a set of points indicating a state where the torque variation amount is constant. The torque line Te (k +2) projected on the flux plane of the axial coordinate system of the stator qds is plotted as a straight line connecting the above points. The torque line Te (k +2) defines the stator qds shaft magnetic flux command value qds _ s _ com (k +2) to obtain a desired torque in the next control cycle. This torque line Te (k +2) is determined using the air gap torque Te _ com (k +1), the rotor qds shaft magnetic flux estimate λ qdr _ s _ est (k + α), and the stator qds shaft magnetic flux estimate λ qds _ s _ est (k + α).
Here, for the convenience of analysis, the magnetic flux plane of the realigned coordinate system shown in fig. 6B will be used. The coordinate system dependent variables are transformed by coordinate transformation. For example, the rotor qds axis flux estimate λ qdr _ s _ est (k + α) and the stator qds axis flux estimate λ qds _ s _ est (k + α) of the stator flux coordinate system are converted to the rotor qds axis flux estimate λ qdr _ ras _ est (k + α) and the stator qds axis flux estimate λ qds _ ras _ est (k + α) of the realigned coordinate system, respectively. As shown in fig. 6B, when the rotor qds axis magnetic flux estimate λ qdr _ ras _ est (k + α) is aligned parallel to the ds axis, the torque line Te (k +2) is parallel to the ds axis. This figure corresponds to a second solution described later.
In order to obtain a desired torque, it is necessary to provide a certain stator qds-shaft voltage command value Vqds _ ras _ com (k +1) within a period of one control cycle (sampling cycle ts) using the
As described above, the strength of the stator flux as indicated is defined as the flux circle λ c (k + 2). the intersection between the torque line Te (k +2) and the flux circle λ c (k +2) is a point representing the required torque.there are two intersections between the torque line Te (k +2) and the flux circle λ c (k + 2). in addition to these, a point located inside the hexagon and reachable from the origin by control is extracted as a point representing the required torque.DB-
The DB-
For example, the DB-
Torque line processing unit 141 calculates a voltage-time product command value Vqds _ ras _ com (k +1) × t by performing a calculation process for specifying a torque line of air gap torque command value Te _ com (k +1)sIs an example of a voltage time product command value.
For example, the torque line processing unit 141 receives input variables including the air-gap torque command value Te _ com (k +1), the stator qds shaft magnetic flux estimate value λ qds _ s _ est (k +1), the rotor qds shaft magnetic flux estimate value λ qdr _ s _ est (k +1), the stator qds shaft magnetic flux estimate value λ qds _ s _ est (k + α), the rotor qds shaft magnetic flux estimate value λ qdr _ s _ est (k + α), and the rotor angular velocity estimate value ω r _ est (k +1), and calculates a voltage-time product command value Vqds _ ras _ com (k +1) × t corresponding to the air-gap torque command value Te _ com (k +1)s。
The flux circle processing unit 142 performs calculation processing to determine a product of voltage × time, which defines the magnitude and direction of the flux of the
For example, the magnetic flux circle processing unit 142 receives a magnetic flux circle including a stator qds axis magnetic flux estimated value λ qds _ s _ est (k + α), a stator qds axis magnetic flux estimated value λ qds _ s _ est (k +1), a stator qds axis magnetic flux command value λ qds _ s _ com (k +2), and a voltage-time product command value Vqds _ ras _ com (k +1) × tsAnd a desired voltage-time product command value Vds _ ras _ com (k +1) × t is obtained based on the magnetic flux circle λ c on the magnetic flux plane of the realignment coordinate system and the above-mentioned torque line Tes。
Voltage conversion unit 143 outputs the voltage-time product command value Vqds _ ras _ com (k +1) × t calculated by torque line processing unit 141sDivided by the sampling period tsAt least the q-axis component of the voltage-time product command value Vqs _ ras _ com (k +1) is calculated.
The voltage conversion unit 144 processes the magnetic flux circle142, and a voltage-time product command value Vqds _ ras _ com (k +1) × tsDivided by the sampling period tsVoltage-time product command value Vds _ ras _ com (k +1) of the d-axis component is calculated.
RAS coordinate inverse transformation unit 145 calculates stator qds axis voltage command value Vqds _ s _ com (k +1) by performing RAS inverse transformation on voltage-time product command value Vqs _ RAS _ com (k +1) calculated by voltage conversion unit 143 and voltage-time product command value Vds _ RAS _ com (k +1) calculated by voltage conversion unit 144 based on rotor qds axis magnetic flux estimate value λ qdr _ s _ est (k + α).
For example, the RAS transform is defined using the following equation (21).
[ mathematical formula 21]
For example, an inverse RAS transform corresponding to the above description is defined using the following equation (22).
[ mathematical formula 22]
A more specific example of each component of the DB-
For example, the torque line processing unit 141 includes a calculation block 1410, a calculation block 1411A, a calculation block 1411B, a calculation block 1412, a calculation block 1413, a calculation block 1414, a calculation block 1415, a calculation block 1416, a calculation block 1417, a calculation block 1418, and a calculation block 1419.
The calculation block 1410 is an example of an estimated torque calculation unit. For example, the calculation block 1410 receives input variables including the stator qds shaft flux estimate λ qds _ s _ est (k +1) and the rotor qds shaft flux estimate λ qdr _ s _ est (k +1), and calculates the air gap torque estimate Te _ est (k +1) (torque estimate). The calculation block 1410 calculates an inner product of the two vectors, i.e., the stator qds shaft magnetic flux estimated value λ qds _ s _ est (K +1) and the rotor qds shaft magnetic flux estimated value λ qdr _ s _ est (K +1), and multiplies the inner product by a prescribed coefficient K to calculate the air gap torque estimated value Te _ est (K + 1). The above process is represented by formula (23).
[ mathematical formula 23]
The calculation block 1412 multiplies the air gap torque estimation value Te _ est (k +1) by a coefficient represented by the following equation (24).
[ mathematical formula 24]
The calculation block 1411A is a subtractor. Calculation block 1411A subtracts the airgap torque estimation value Te _ est (k +1) from the airgap torque command value Te _ com (k +1) to obtain the difference thereof as Δ Te _ est (k + 1). The above calculation is represented by equation (25).
[ mathematical formula 25]
Calculation block 1411B is an adder. The calculation block 1411B adds Δ Te _ est (k +1) calculated by the calculation block 1411A to the result calculated by the calculation block 1412.
The calculation block 1413 multiplies the result calculated by the calculation block 1411B by the following equation (26).
[ mathematical formula 26]
The calculation block 1414 normalizes the result calculated by the calculation block 1413 based on the rotor qds shaft magnetic flux estimation value λ qdr _ s _ est (k + α). For example, the calculation block 1414 divides the result obtained by the calculation block 1413 by the result calculated by the calculation block 1415. The output of the calculation block 1414 represents a torque reference in the realignment coordinate system.
The calculation block 1415 calculates an absolute value (norm) of the rotor qds shaft magnetic flux estimated value λ qdr _ ras _ est (k + α) based on the rotor qds shaft magnetic flux estimated value λ qdr _ s _ est (k + α). The realignment coordinate system may be matched with the stator-side coordinate system by using the result of the calculation block 1415.
The calculation block 1416 performs RAS coordinate transformation of the stator qds shaft magnetic flux estimated value λ qds _ s _ est (k + α) with reference to the rotor qds shaft magnetic flux estimated value λ qdr _ s _ est (k + α).
The calculation block 1417 multiplies the rotor angular velocity estimate ω r _ est (k +1) by the sampling period ts. The calculation block 1418 calculates an inner product of a vector represented by the result calculated by the calculation block 1416 and a vector represented by the result calculated by the calculation block 1417.
Calculation block 1419 adds the result calculated by calculation block 1414 and the result calculated by calculation block 1418 to generate voltage-time product command value Vqs _ ras _ com (k +1) × t corresponding to air gap torque command value Te _ com (k +1)0。
The above-described calculation block 1411A, calculation block 1411B, calculation block 1412, calculation block 1413, calculation block 1414, calculation block 1415, calculation block 1416, calculation block 1417, calculation block 1418, and calculation block 1419 are examples of torque line processing units.
As described above, the torque line processing unit 141 calculates the voltage × time product corresponding to the product of the average voltage and the control period based on at least the air-gap torque command value Te _ com (k +1) (the torque command of the motor 2), the air-gap torque estimated value Te _ est (k +1) (the torque estimated value of the motor 2), the stator qds shaft magnetic flux estimated value λ qds _ s _ est (k + α) (the second estimated value of the stator magnetic flux of the motor 2), the rotor qds shaft magnetic flux estimated value λ qdr _ s _ est (k + α) (the second estimated value of the rotor magnetic flux of the motor 2), and the rotation speed ω r of the rotor magnetic flux of the
For example, flux circle processing unit 142 includes a calculation block 1420, a calculation block 1421, a calculation block 1422, a calculation block 1423, a calculation block 1424, a calculation block 1425, and a calculation block 1426.
The calculation block 1420 calculates the stator qds axis magnetic flux estimated value λ qds _ RAS _ est (k +1) by performing RAS coordinate conversion of the stator qds axis magnetic flux estimated value λ qds _ s _ est (k +1) with reference to the rotor qds axis magnetic flux estimated value λ qdr _ s _ est (k + α).
The calculation block 1422 calculates the stator qds shaft current estimated value Iqds _ RAS _ est (k + α) by performing RAS coordinate conversion of the stator qds shaft current estimated value Iqds _ s _ est (k + α) with reference to the rotor qds shaft magnetic flux estimated value λ qdr _ s _ est (k + α). the calculation block 1423 multiplies (t + α) by the stator qds shaft magnetic flux estimated value λ qds _ RAS _ est (k + α)s× Rs.) calculation block 1421 is a subtractor calculation block 1421 subtracts stator qds axis current estimates Iqds _ ras _ est (k + α) and (t) from rotor qds axis flux estimate λ qdr _ ras _ est (k +1)s× Rs) of the magnetic flux circle, the center of the magnetic flux circle is corrected using the product of the magnitude of the stator qds axis current estimate Iqds _ ras _ est (k + α) and the stator resistance Rs, thereby improving the positional accuracy of the magnetic flux circle λ c, when the product of the magnitude of the stator qds axis current estimate Iqds _ ras _ est (k + α) and the stator resistance Rs is sufficiently small with respect to the radius of the magnetic flux circle, the process of correcting the center of the magnetic flux circle may be omitted.
By the processing with each of the above-described calculation blocks 1420 to 1423, the voltage × time product corresponding to the voltage drop according to the stator qds axis current estimation value Iqds _ ras _ est (k + α) flowing through the stator resistance Rs is subtracted from the stator qds axis magnetic flux estimation value λ qds _ ras _ est (k +1) in the realigned coordinates, thereby obtaining a stator qds axis magnetic flux estimation value λ qds _ ras _ est (k +1) β reflecting the influence of the stator resistance Rs. The stator qds shaft flux estimate λ qds _ ras _ est (k +1) β depends on the stator qds shaft current estimate Iqds _ ras _ est (k + α).
The calculation block 1424 adds the result calculated by the torque line processing unit 141 to the result calculated by the calculation block 1421.
The calculation block 1425 includes a D-axis coordinate calculation unit. For example, the calculation block 1425, which is a D-axis coordinate calculation unit, calculates the position of the intersection of the torque line and the magnetic flux circle λ c based on the stator qds axis magnetic flux command value λ qds _ s _ com (k +2) and the calculation result of the calculation block 1424. Details thereof will be described later.
The calculation block 1426 subtracts the result calculated by the calculation block 1421 from the result calculated by the calculation block 1425.
The above-described flux circle processing unit 142 determines the radius of the flux circle based on the command value of the rotor flux of the
Details of the DB-DTFC will be described.
First, equations (27-1) to (27-4) represent equivalent equations of an induction motor in an arbitrary coordinate system.
[ mathematical formula 27]
Vqds=Rsiqds+(p+jωe)λqds(27-1)
O=Rriqdr+[p+j(ωe-ωr)]λqdr(27-2)
λqds=Lsiqds+Lmiqdr(27-3)
λqdr=Lmiqds+Lriqdr(27-4)
By solving the above equations (27-3) and (27-4), the stator current and the rotor current are obtained using the following equations (28-1) and (28-2).
[ mathematical formula 28]
Here, the leakage coefficient σ is defined as shown in the following expression (29).
[ mathematical formula 29]
By substituting formula (28-1), formula (28-2) and formula (29) into formula (27-1) and formula (27-2), formula (30-1) and formula (30-2) are obtained.
[ mathematical formula 30]
The following equations (31-1) and (31-2) are calculated by converting the equations (30-1) and (30-2).
[ mathematical formula 31]
In the above equations (31-1) and (31-2), the following equations (32-1) and (32-2) are calculated by substituting "0" into ω e through RAS coordinate transformation using the characteristics of the rotating coordinate system.
[ mathematical formula 32]
The following expression (33) is an expression for calculating the torque of the
[ mathematical formula 33]
The following equation (34) representing the rate of change of torque is calculated by differentiating the equation (33).
[ mathematical formula 34]
The above-described equations (31-1) and (31-2) can be converted into equations (35-1) to (35-4) expressed as scalars.
[ math figure 35]
By substituting formulae (35-1) to (35-4) into formula (34), the following formula (36) can be represented.
[ mathematical formula 36]
Equation (36) expressed in the continuous time system is converted into equation (37) below in the discrete time system.
[ mathematical formula 37]
Equation (38) is calculated by converting equation (37) representing the rate of change of torque.
[ mathematical formula 38]
The above equation (38) is an equation representing the torque line. By defining the variable m as represented by formula (39-2) and defining the variable b as represented by formula (39-3), formula (38) can be simplified to formula (39-1).
[ math figure 39]
Vqs(k+1)ts=m×Vds(k+1)ts+b (39-1)
As described above, in the case of the second solution, m corresponds to the slope with respect to the ds axis in fig. 6B, and thus m becomes 0. When m is "0", the qs-axis component Vqs _ ras _ com (k +1) of the voltage command value in the realignment coordinate system is calculated using the following equation (40).
[ mathematical formula 40]
Vqs(k+1)=b/ts(40)
Next, processing related to the magnetic flux circle λ c will be described.
The differential value of the stator qds axis magnetic flux λ qds can be expressed by the following equation (41).
[ math figure 41]
pλqds=Vqds-Rsiqds-jωeλqds(41)
The expression (41) is converted into an expression (42) of a discrete time system.
[ mathematical formula 42]
As shown in equation (43), equation (42) is divided into a qs-axis component and a ds-axis component.
[ math figure 43]
In the above equation (43), the qs-axis component Vqs _ s _ com (k +1) of the voltage command value in the stator stationary coordinate system may be obtained using equation (42) as a known variable. By solving equation (43), ds-axis component Vds _ s _ com (k +1) of the voltage command value is calculated.
In the above description, although the stator qds shaft magnetic flux estimated value λ qds _ s _ est (k +2) has been used, it may be replaced with the stator qds shaft magnetic flux command value λ qds _ s _ com (k +1) as described above.
In the above description, although the stator stationary coordinate system is described as an example, the realignment coordinate system may be used in a similar manner. The qs-axis component Vqs _ ras _ com (k +1) of the voltage command value may be acquired using the expression of the realigned coordinate system converted from expression (42) as a known variable. By solving the equation of the realigned coordinate system converted from equation (43), ds-axis component Vds _ ras _ com (k +1) of the voltage command value can be obtained.
According to the above embodiment, the current magnetic
In some cases, the accuracy of control can be improved by compensating for the amount of delay in the calculation performed by the
The
As described above, the
In calculating the rotor magnetic flux estimated value λ qdr _ s _ est (k +1), the
The above-described delay may be included in the values of the state variables of the
The state from the time point k to the time point (k +2) changes according to the passage of time. However, when the change in the value of the synchronization angular frequency ω e is considered to be constant and the trend of the state change from the time point k to the time point (k +2) is assumed to be continuous, the change in the state is predictable. The
For example, when a time of about (ts/2) elapses from the time point of the point G22 shown in the graph in the middle of fig. 22, this will be the time point (k + 1). During this period, the state at the point in time at which the point G22 is located changes along the line of the stator current Iqds _ s, and is detected as the stator current measurement value Iqds _ s _ det (k + 1). Since the stator current measurement value Iqds _ s _ det (k +1) contains much noise, the stator current estimation value Iqds _ s _ est (k +1) calculated by the
Thus, the
The above description using the stator current Iqds _ s can be similarly applied to the case of using the above-described stator magnetic flux λ qds _ s or rotor magnetic flux λ qdr _ s. The point G11, the point G13, and the point G14 in the stator magnetic flux λ qds _ s correspond to the point G21, the point G23, and the point G24 of the stator current Iqds _ s, respectively. A point G31, a point G33, and a point G34 in the rotor magnetic flux λ qdr _ s correspond to a point G21, a point G23, and a point G24 of the stator current Iqds _ s, respectively.
As described above, the
According to the embodiment, the control accuracy performed by the DB-DTFC can be improved by reducing the influence of the voltage drop due to the resistance component of the stator resistance Rs.
When the influence of the voltage drop due to the resistance component of the stator resistance Rs is small, the second term in the above equation (42) may be omitted.
(first modification of the first embodiment)
A first modification of the first embodiment will be described with reference to the drawings.
In the first embodiment, the case related to the second solution in the DB-DTFC control has been described. Instead, in this modification, a case related to the first solution in the DB-DTFC control will be described.
Fig. 8 is a diagram illustrating a first solution in DB-DTFC control according to an embodiment. Fig. 8 shows an example of a flux plane of a flux coordinate system. The flux coordinate system shown in fig. 8 is a realigned coordinate system aligned with respect to the stator qds axis flux estimated value λ qds _ s _ est (k + α). The ds axis is directed downward in the figure and the qs axis is directed to the right in the figure.
FIG. 8 differs from FIG. 6B in that the stator qds shaft flux estimate λ qds _ s _ est (k + α) is aligned with the ds axis. In this case, the torque line Te (k +2) is not parallel to the ds axis.
Therefore, in the first solution, the intersection point between the flux circle and the torque line is derived using a method for obtaining a solution of a quadratic equation by simultaneously solving an equation representing the flux circle and an equation representing the torque line.
For simplification of explanation, expression (43) indicating the magnetic flux circle λ c is converted into expression (44). This equation (44) is simplified by neglecting the effect of the stator resistance Rs.
[ math figure 44]
By solving the equations for the flux loop and the torque line simultaneously, these equations can be converted to the following equation (45). By solving this, qs-axis component Vqs _ ras _ com (k +1) of the voltage command value can be obtained from equation (46), and ds-axis component Vds _ ras _ com (k +1) of the voltage command value can be obtained from equation (45-3).
[ mathematical formula 45]
[ mathematical formula 46]
[ math figure 47]
According to the above modification, the realignment coordinate system aligned with respect to the stator qds axis magnetic flux estimated value λ qds _ s _ est (k + α) is used. This modification can also achieve effects similar to those of the above-described embodiment.
(second modification of the first embodiment)
A second modification of the embodiment will be described with reference to the drawings.
In the first embodiment, the case where the latch interface is used for the stator flux in the process performed by the
A modification of the
Fig. 9 is a block diagram showing a current observer 21A according to a second modification of the first embodiment.
Instead of the
The
[ math figure 48]
The
The
For example, the
[ math figure 49]
In the above formula (49), a1, B0 and B1 are defined by the following formula (50). Here, χ is an arbitrary variable.
[ mathematical formula 50]
As described above, the
The above-described
The
The
The calculation block 213B is an adder. The calculation block 213B adds the calculation result calculated by the
The description will be continued on the configuration of the current observer 21A.
In the transition from a continuous system to a discrete system, the current observer 21A applies a latch interface to the voltage input and a ramp interface to the flux interface.
[ mathematical formula 51]
In the above formula (51), formula (52) is defined.
[ math 52]
Since equation (51) is based on a linear system, equation (53) below can be obtained by applying the superposition principle.
[ mathematical formula 53]
χ in the above formula (53) is defined in the following formula (54).
[ math formula 54]
As described above, by applying the latch interface to the stator voltage, applying the ramp interface to the stator flux, and converting equation (54) into an equation of a discrete time system, equation (55) and equation (56) are obtained. Z [ ] represents a Z transformation.
[ math figure 55]
[ math figure 56]
Here, by defining a1, B0, and B1 as described above and simplifying formula (55) and formula (56), the following formula (57) is obtained.
[ math formula 57]
According to the above-described modification, effects similar to those according to the embodiment are obtained even when current observer 21A receives the magnetic flux estimation value using the latch interface.
(second embodiment)
The motor drive system 1A according to the second embodiment will be described with emphasis on the DB-DTFC calculation unit 14A. For example, in the first embodiment, the estimation accuracy of the current and the magnetic flux is improved using the observer, and the value of the state variable (state quantity) in the DB-DTFC control is corrected by the average correction. In the second embodiment, for the DB-DTFC control, the accuracy of estimating the current and the magnetic flux is improved by the observer, but the effect produced by correcting the value of the state variable by the above-described average correction is not obtained.
Fig. 10 is a block diagram showing a motor drive system 1A according to the second embodiment.
For example, the control device 10A of the motor drive system 1A is different from the
The DB-DTFC calculation unit 14A receives input variables including the air gap torque command value Te _ com (k +1), the rotor qds shaft magnetic flux estimation value λ qdr _ s _ est (k +1), the stator qds shaft magnetic flux estimation value λ qds _ s _ est (k +1), and the stator qds shaft magnetic flux command value λ qds _ s _ com (k +2), and calculates the stator qds shaft voltage command value Vqds _ s _ com (k +1) based on the input variables. The stator qds shaft flux estimate λ qds _ s _ est (k +1) and the rotor qds shaft flux estimate λ qdr _ s _ est (k +1) are provided by the current
While the above-described DB-
Fig. 11 is a block diagram showing the DB-DTFC calculation unit 14A according to the embodiment.
The DB-DTFC calculation unit 14A includes a torque line processing unit 141A, a flux circle processing unit 142A, and an RAS coordinate inverse transformation unit 145A, instead of the torque line processing unit 141, the flux circle processing unit 142, and the RAS coordinate inverse transformation unit 145 of the above-described DB-
The torque line processing unit 141A includes a calculation block 1415A and a calculation block 1416A instead of the calculation block 1415 and the calculation block 1416 of the torque line processing unit 141.
The calculation block 1415A calculates the magnitude (norm) of the stator qds axis magnetic flux estimated value λ qds _ s _ est (k + 1).
The calculation block 1416A performs RAS coordinate transformation of the stator qds axis magnetic flux estimate λ qds _ s _ est (k +1) with reference to the stator qds axis magnetic flux estimate λ qds _ s _ est (k +1) to generate a stator qds axis magnetic flux estimate λ qds _ RAS _ est (k + 1). As a result of the above calculation, the stator qs-axis magnetic flux estimate λ qs _ ras _ est (k +1) is zero, and the stator ds-axis magnetic flux estimate λ ds _ ras _ est (k +1) is the magnitude of the stator qds-axis magnetic flux estimate λ qds _ s _ est (k + 1).
The magnetic flux circle processing unit 142A is different from the magnetic flux circle processing unit 142 in that the magnetic flux circle processing unit 142A does not include the calculation block 1421, the calculation block 1422, and the calculation block 1423, but includes the calculation block 1420A instead of the calculation block 1420.
The calculation block 1420A has the same configuration as the calculation block 1416A described above. The calculation block 1420A supplies the calculation result, i.e., the stator qds axial magnetic flux estimated value λ qds _ ras _ est (k +1), to the calculation block 1424 and the calculation block 1426 at the subsequent stage.
As described above, in this second embodiment, the function of a part of the
(third embodiment)
A
For example, in the above-described embodiment, the deadbeat direct torque control (DB-DTFC) system is used to generate the voltage command value. This third embodiment uses a magnetic field direction control system. Hereinafter, the magnetic field direction control system is referred to as a FOC system.
In the FOC system, a current component generating torque (rotational force) and a current component generating magnetic flux are separated, and each current component is independently controlled to a direct current amount.
Examples of FOC systems include direct field direction control (DFOC) systems. In the DFOC system, the flux vector is directly estimated by a flux sensor or flux observer and controlled.
Fig. 12 is a block diagram showing a
Examples of FOC systems include indirect magnetic field direction control (IFOC) systems, and direct magnetic field direction control (DFOC) systems. The IFOC system uses indirect vector control (also referred to as slip frequency type vector control) to control the slip of the induction motor without estimating or detecting the magnetic flux. The DFOC system controls the slip of the
The
The
The torque command value Te _ com (k +1) calculated by the
For example, the speed/
For example, the speed/
For example, the speed/
For example, the slip
The speed/
By appropriately selecting the phase of the reference signal, the seventh coordinate
Contrary to the above, sixth coordinate transforming
Eighth coordinate transformation section 17B converts three-phase stator voltage command values Vus _ com (k +1), Vvs _ com (k +1), and Vws _ com (k +1) into voltage command values Vqds _ s _ com (k +1) of the two-axis components of the dqs axis. The value converted by the eighth coordinate conversion unit 17B is supplied to the current magnetic
The rotor qds shaft magnetic flux estimated value λ qdr _ s _ est (k +1) and the stator qds shaft magnetic flux estimated value λ qds _ s _ est (k +1) estimated by the current magnetic
The
Voltage command value V γ _ com (k +1) of the γ component and voltage command value V _ com (k +1) of the component are supplied to sixth coordinate
In the third embodiment described above, although the control system is different from the first and second embodiments, the current magnetic
The
The control system according to an embodiment is a FOC. In the case of FOC, a feedback signal of the current (current FBK) is used for control by the
In the
The non-speed sensor control may also be applied to the above DB-DTFC. When the DB-DTFC performs the speed sensorless control, the control accuracy means the accuracy of the speed control, the magnetic flux control, and the torque control.
(modification of the third embodiment)
For example, in the third embodiment, the
(control device of embodiment)
The
(examples)
The evaluation results according to the embodiment will be described with reference to fig. 14 to 18.
Fig. 14 and 15 are diagrams showing the evaluation results of the
On the other hand, fig. 16 is a diagram showing the evaluation results of the motor drive system according to the comparative example. In a comparative example shown in fig. 16, a current observer according to a comparative example is applied to the control device 10A described in the first embodiment. The current observer of the comparative example differs from the
The evaluation result shown in each of fig. 14 to 16 is an evaluation result when the switching frequency fs is set to 240 hertz (Hz), and a step response test of torque is performed when the
From the evaluation results of the first embodiment shown in fig. 14, it can be seen that the command value, the measured value, and the estimated value of the torque coincide with each other within a desired range according to the control performed using the
From the evaluation results of the second embodiment shown in fig. 15, it can be seen that the measured value and the estimated value are different from the command value of the torque in the control performed by the control device 10A. Here, although the measured value and the estimated value are different from the torque command value described above, the measured value and the estimated value respond to a step change in torque. The above evaluation result acquired by the control device 10A may be regarded as a value that evaluates the actual torque.
In the evaluation results according to the comparative example shown in fig. 16, the command value, the measured value, and the estimated value of the torque are independent of each other. Although it has been confirmed that the stepwise changes of the values are synchronized with each other, no correlation is found between the values as found in the first and second embodiments.
Therefore, under the above test conditions, it was confirmed that the comparative example may be difficult to apply, and the configuration of the first embodiment showed effective results.
The results of the step response test of the torque will be described. Fig. 17 is a diagram showing the evaluation results of the step response test of torque. The graph in fig. 17 analytically shows the results of the step response test of the torque. The vertical axis represents the magnitude of the torque error, the horizontal axis represents the speed of the rotor, and the depth represents the magnitude of the torque.
The step response test of the torque was performed under two conditions, i.e., 2000 hertz (Hz) and 240 hertz (Hz) as the sampling frequency fs. As a state where the test is performed, the
In any of the cases of 200 hertz (Hz) and 240 hertz (Hz), the results obtained were such that the
In order to verify the above evaluation results, the current magnetic
Fig. 18 is a diagram showing the evaluation result of the current magnetic
In fig. 18, the result of comparison of stator flux λ qs _ s and stator flux estimate λ qs _ s _ est is shown in the first chart at the top, the result of comparison of stator current Iqs _ s and stator current estimate Iqs _ s _ est is shown in the second chart at the top, the result of comparison of rotor flux λ qr _ s and rotor flux estimate λ qr _ s _ est is shown in the third chart at the top, and the three results of comparison of torque command value Te _ com, torque measurement value Te _ det, and torque estimate Te _ est are shown in the chart at the bottom. The conditions tested were as follows: the sampling frequency fs is set to 500 hertz (Hz), the load of the
Here, the evaluation results shown in fig. 18 will be described in comparison.
In the case of the second embodiment, it can be determined that in any one of the stator magnetic flux, the stator current, the rotor magnetic flux, and the torque, stepwise quantization distortion occurs in the estimated value, but the reproducibility of the input waveform is high and follows the input waveform.
On the other hand, in the case of the comparative example, the reproducibility of the input waveforms of the stator current and the rotor magnetic flux is low. As for the torque, a result is obtained such that torque measurement value Te _ det and torque estimation value Te _ est are different from torque command value Te _ com. In the case of the comparative example, there is a value that does not follow the input waveform.
According to at least one of the above embodiments, the
While certain embodiments have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the invention. Indeed, the novel embodiments described herein may be embodied in a variety of other forms; also, various omissions, substitutions, and changes in the form of the embodiments described herein may be made without departing from the spirit of the inventions. The accompanying claims and their equivalents are intended to cover such forms or modifications as would fall within the scope and spirit of the inventions.
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