阅读说明：本技术 具有ac正激电桥和改进的dc/dc拓扑的逆变器 (Inverter with AC forward bridge and improved DC/DC topology ) 是由 法布里斯·法雷贝尔 蒂埃里·琼尼斯 奥利弗·考博 保罗·布莱尤斯 于 2018-11-07 设计创作，主要内容包括：本发明涉及一种具有主DC输入(1)和主单相AC输出(4)的DC到AC功率变换器,该DC到AC功率变换器包括：单个DC到DC变换器(5)；以及首先根据直接路径,与该DC到DC变换器(5)级联的双向电压型DC到AC变换器(6),所述双向电压型DC到AC变换器(6)具有并联连接到该DC输出(10)的DC输入-输出(11)和连接到所述主AC输出(4)的AC输出-输入(12)；以及其次根据旁路路径,并且与所述双向电压型DC到AC变换器(6)和所述低频二极管(2)并联,具有DC输入和AC输出的电流型低频全开关H电桥(7),下文称为AC正激电桥,所述DC输入连接到该单个DC到DC变换器(5)的所述DC输出(10),并且所述AC输出与所述主AC输出(4)并联连接,所述AC正激电桥的工作频率小于1kHz,从而,当所述主AC输出(4)的端子之间的瞬时电压达到预定电平时,该低频正激AC电桥(7)接通,该低频二极管(2)被反向偏置并且不导通,并且该DC到DC变换器(5)将恒定功率直接提供给该负载。(The invention relates to a DC-to-AC power converter having a main DC input (1) and a main single-phase AC output (4), the DC-to-AC power converter comprising: a single DC-to-DC converter (5); and a bidirectional voltage-type DC-to-AC converter (6) cascaded with the DC-to-DC converter (5) first according to a direct path, said bidirectional voltage-type DC-to-AC converter (6) having a DC input-output (11) connected in parallel to the DC output (10) and an AC output-input (12) connected to said main AC output (4); and secondly a current-mode low-frequency full-switching H-bridge (7), hereinafter AC-forward bridge, according to a bypass path and in parallel with said bidirectional voltage-mode DC-to-AC converter (6) and said low-frequency diode (2), said DC input being connected to said DC output (10) of the single DC-to-DC converter (5) and said AC output being connected in parallel with said main AC output (4), the operating frequency of said AC-forward bridge being less than 1kHz, so that, when the instantaneous voltage between the terminals of said main AC output (4) reaches a predetermined level, the low-frequency forward AC bridge (7) is switched on, the low-frequency diode (2) is reverse biased and non-conductive, and the DC-to-DC converter (5) supplies a constant power directly to the load.)

1. A DC-to-AC power converter having a main DC input (1) and a main single-phase AC output (4), capable of converting and adapting a DC voltage at the main DC input (1) to a fundamental frequency f at the main AC output (4)₀And is capable of delivering a rated power at the main AC output (4) to a load, the DC to AC power converter comprising:

-a single DC-to-DC converter (5) having said main DC input (1) as an input and having a DC output (10) and a tank capacitor (3) connected to said DC output (10), two low frequency diodes (2) biased so as to be able to transfer current from the DC output (10) to the tank capacitor (3), respectively from the tank capacitor (3) to the DC output (10);

-a bidirectional voltage-type DC-to-AC converter (6) cascaded with the DC-to-DC converter (5) according to a direct path, said bidirectional voltage-type DC-to-AC converter (6) having a DC input-output (11) connected to the DC output (10) and an AC output-input (12) connected to said main AC output (4);

-a current-mode low-frequency fully-switched H-bridge (7), hereinafter referred to as AC forward bridge, having a DC input connected to the DC output (10) of the single DC-to-DC converter (5) and an AC output connected in parallel with the main AC output (4), according to a bypass path and in parallel with the bidirectional voltage-mode DC-to-AC converter (6) and the low-frequency diode (2), the operating frequency of the AC forward bridge being less than 1kHz and preferably 400Hz or 50/60 Hz;

-control means for controlling said bidirectional voltage-type DC-to-AC converter (6) to deliver a sinusoidal AC voltage at said first AC output-input (12) and for controlling said AC forward bridge to deliver a quasi-square AC forward current in phase with said sinusoidal AC voltage, said control means being able to control said bidirectional voltage-type DC-to-AC converter (6) and said AC forward bridge so that the latter can operate simultaneously;

thus, when the instantaneous voltage between the terminals of the main AC output (4) reaches a predetermined level, the low frequency forward AC bridge (7) is switched on, the low frequency diode (2) is reverse biased and non-conducting, and the DC-to-DC converter (5) provides constant power directly to the load.

2. The DC-to-AC power converter as claimed in claim 1, wherein the two closed switches (TPH, TNL; TPL, TNH) of the low frequency AC forward bridge (7) are selected according to the polarity of the output AC voltage.

3. The DC to AC power converter as defined in claim 1, wherein the DC to DC converter (5) is designed to support a variable output voltage while delivering a nearly constant power.

4. The DC to AC power converter according to claim 1, wherein the DC to DC converter (5) is isolated and comprises an active clamp on the primary side of the transformer, the active clamp being formed by a main mosfet (mp) connecting the primary winding (TFO-P) of the transformer to a primary power supply (VIN +, VIN-) providing the main DC input (1), a resonant Capacitor (CRP) in parallel with the primary mosfet (mp) to allow the primary mosfet (mp) to operate at ZVT, and a second mosfet (maux) arranged to provide a voltage clamp on the main mosfet (mp) to protect the main mosfet (mp) from overvoltage.

5. The DC to AC power converter according to claim 1, wherein the DC to DC converter (5) is isolated and comprises on the primary side of the transformer: a two-transistor forward converter primary stage consisting of two MOSFETs (MP1, MP2), a respective one end of each MOSFET being directly connected to one end of the primary winding (TFO-P) of the transformer and to a respective primary power supply terminal (VIN-, VIN +) providing the main DC input (1); -a resonant capacitor (CRP1, CRP2) connected in parallel with each MOSFET (MP1, MP2) respectively to allow said MOSFET (MP1, MP2) to operate at ZVT; and diodes (DAUX1, DAUX2) arranged to connect each main MOSFET (MP1, MP2) to a primary power supply terminal (VIN +, VIN-) that is not directly connected to the primary power supply terminal (VIN-, VIN +) of the corresponding MOSFET (MP1, MP2), respectively.

6. The DC to AC power converter as claimed in claim 4 or 5, wherein the DC to DC converter (5) further comprises on the secondary side of the transformer: at least a first capacitor (CS1, CS2) establishing an AC connection to the secondary winding (TFO-S) of the transformer; a resonant inductance (LR) that can be reduced to a leakage inductance of the transformer; a rectifier Diode (DR) and a freewheel Diode (DLR) for rectifying a voltage generated at the secondary side of the transformer; and a secondary resonance Capacitor (CRS) connected in parallel with the rectifier Diode (DR); and a decoupling capacitor (CS3) connected in parallel with a terminal of the output DC voltage (10).

7. The DC to AC power converter according to claim 6, wherein the first capacitor (CS1), the resonant inductance (LR), the rectifier Diode (DR), and the freewheeling Diode (DLR) are arranged such that during a magnetization phase of the transformer, the converter input voltage VIN reflected to the secondary of the transformer as NVIN charges the first capacitor (CS1) through the rectifier Diode (DR) and creates a resonance between the first capacitor (CS1) and the resonant inductance (LR), where N is the transformer turns ratio, the freewheeling Diode (DRL) is non-conductive, and there is no overvoltage at the junction between the rectifier Diode (DR) and the freewheeling Diode (DRL).

8. The DC to AC power converter according to claim 6, wherein the first capacitor (CS1), the resonant inductance (LR), the rectifier Diode (DR) and the freewheeling Diode (DLR) are arranged such that during a demagnetization phase of the transformer, current flows from the charged first capacitor (CS1) through the resonant inductance (LR) and the freewheeling Diode (DRL) to the load, said current not only transferring the transformer magnetization energy, but also simultaneously transferring the energy stored in the first capacitor (CS1) during the magnetization phase.

9. The DC to AC power converter as recited in claim 8, wherein the output power (P) of the DC/DC converter_O) The output power (P) of the equivalent flyback converter is obtained by the following equation_FB) And (3) correlation:

P_O＝P_FBm, wherein,

wherein, V_INAnd V_OBeing the input voltage and the output voltage, respectively, of the DC/DC converter, M is a multiplication factor larger than 1, preferably larger than 2.

10. The DC to AC power converter as recited in claim 1, wherein the DC to AC power converter is bidirectional.

11. The DC to AC power converter according to claim 1, wherein the low frequency diodes (2) are replaced by controlled switches.

12. The DC to AC power converter according to claim 11, wherein the controlled switches are MOSFETs, IGBTs, or relays.

13. The DC to AC power converter as recited in claim 1, wherein the DC to DC converter is non-isolated.

14. An isolated DC-to-DC converter (5) adapted for use in a DC-to-AC power converter according to claim 1 having a main DC input (1) and a main single-phase AC output (4), the isolated DC-to-DC converter comprising: on the primary side of the transformer, an active clamp consisting of a main mosfet (mp) connecting the primary winding (TFO-P) of the transformer to a primary power supply (VIN +, VIN-) providing the main DC input (1), a resonant Capacitor (CRP) in parallel with the primary mosfet (mp) to allow the primary mosfet (mp) to operate at ZVT, and a Capacitor (CAUX) and a second mosfet (maux) arranged to provide a voltage clamp over the main mosfet (mp) to protect the main mosfet (mp) from overvoltage.

Technical Field

The present invention relates to a DC/AC power converter with improved efficiency comprising a main path consisting of a DC/DC converter followed by a bi-directional voltage type DC/AC converter and a bypass path consisting of a current type DC/AC converter, wherein power peaks are carried by the bypass converter to minimize losses.

In particular, the invention is an improvement of the DC/AC power converter described in document WO 2016/083143 a 1.

Background

Fig. 1 shows a classical inverter topology. The DC/DC input stage, whether (potential) isolated or not, transforms the input DC voltage into a different DC voltage well suited to power an output non-isolated DC/AC inverter bridge, which typically operates in the frequency range of a few kHz to a few MHz and produces a 50Hz or 60Hz sine wave output by typically using high frequency PWM modulation.

For efficiency, document US 7,710,752B 2 shows a parallel configuration in which one path is built around the boost bridge converter and the other direct path is built around a simple bridge converter.

The problem with this topology is that the most efficient direct path can only be activated when the instantaneous voltage of the output is sufficiently low. When the instantaneous power demand is maximum, it cannot be activated. Efficiency improvements are not optimal for increased part count.

Document EP 2270624 a1 proposes another solution that optimizes efficiency by alternately connecting three current generators (with different characteristics) to three output phases based on which connection is optimal to maximize efficiency. This topology can only be applied to three-phase output systems and does not solve the insulation problem.

Document WO 2016/083143 a1 proposes an isolated DC to AC converter which provides efficiency improvements by increasing the direct path from the DC input to the AC output. This approach improves efficiency, but has the disadvantage of requiring an additional switching converter or an additional winding in the isolated DC/DC converter transformer.

Document US 7,778,046B1 proposes an interesting solution to improve the efficiency of a DC/DC converter that can be used in an inverter to perform the required insulation. Fig. 35a is particularly representative thereof. Although a very interesting mode of operation, this invention has several drawbacks:

the converter cannot fully operate in soft switching operation, which prevents the use of high operating frequencies;

switch S has no overvoltage protection (no clamping effect);

diode CR2 has no overvoltage protection. When switch S is closed (or switched on), capacitor C2 continues to be charged by the current flowing in the leakage inductance of the transformer, diode CR2 is blocked due to the occurrence of an overvoltage at the terminals of diode CR2 (after its reverse recovery phase), which may be several times the diode blocking voltage, and may damage diode CR 2. In addition, the possible clamping effect of diode CR1 is counteracted by the series inductance Lr, since an overvoltage is generated across diode CR1 when switch S is closed. This problem makes it difficult to increase the power as needed.

Document EP 1852964 a1 discloses a maximum single-phase inverter 3B-INV which uses, as its DC power supply, a DC voltage V3B boosted from a solar photovoltaic voltage by a step-up chopper circuit arranged at the center, and single-phase inverters 2B-INV and 1B-INV which use, as their inputs, DC power supplies V1B and V2B supplied from the maximum DC power supply V3B are arranged on both sides of the maximum single-phase inverter 3B-INV. The AC sides of the respective single-phase inverters are connected in series. Thus, the power regulator is configured to provide an output voltage by using the sum of the voltages generated by the respective single-phase inverters. The chopper circuit is connected between the maximum DC power source V3B and the DC power sources V1B and V2B, and power is supplied from the maximum DC power source V3B to the DC power sources V1B and V2B via the switching devices in the single-phase inverter.

Document US 2017/025962 a1 discloses two versions of an isolated single-stage converter AC/DC Power Factor Correction (PFC) converter topology. One version with a full bridge rectifier at its input and the other version is a true bridgeless version. These two topology versions are characterized by their new configurations and circuits including simplified damper circuits and clamp capacitor flipping circuits, their control methods allowing them to implement improved single-stage isolated power factor converters suitable for high power operation, with zero voltage switching features to maximize conversion efficiency and minimize electromagnetic interference generation, no additional circuitry required to limit inrush current, achieve reasonably low input current Total Harmonic Distortion (THD), and are easy to control. The second version provides a truly bridgeless single-stage isolated power factor converter with even higher efficiency and lower input current THD.

Document US 2013/223106 a1 discloses a switching circuit for a power converter comprising a first active switch coupled between a first terminal of an input of the power converter and a first terminal of a primary winding of a transformer. The second active switch is coupled between the second terminal of the input terminal and the second terminal of the primary winding. The output capacitance of the first active switch is greater than the output capacitance of the second active switch. The first passive switch is coupled between the second terminal of the primary winding and the first terminal of the input terminal. The second passive switch is coupled between the second terminal of the input terminal and the first terminal of the primary winding. The reverse recovery time of the first passive switch is greater than the reverse recovery time of the second passive switch.

Object of the Invention

The present invention aims to propose an inverter that converts a wide range of DC input voltages to AC output voltages with as high an efficiency as possible while maintaining strict requirements on the shape of the input DC current.

In particular, one of the requirements is to maintain a constant DC input current, and thus a constant input power, even in case of fluctuations in the AC output power during a 50Hz/60Hz AC cycle.

Another object of the invention is to make the inverter as compact as possible.

Disclosure of Invention

A first aspect of the invention relates to a DC-to-AC power converter having a main DC input and a main single-phase AC output, capable of converting and adapting a DC voltage at the main DC input to a fundamental frequency f at the main AC output₀And is capable of delivering a rated power at the main AC output to a load, the DC to AC power converter comprising:

-a single DC-to-DC converter having said main DC input as an input and having a DC output and a tank capacitor connected to said DC output, two low frequency diodes being biased so as to be able to transfer current from the DC output to the tank capacitor and, correspondingly, from the tank capacitor to the DC output;

-a bidirectional voltage-type DC-to-AC converter cascaded with the DC-to-DC converter according to a direct path, the bidirectional voltage-type DC-to-AC converter having a DC input-output connected to the DC output and an AC output-input connected to the main AC output;

-a current-mode low-frequency full-switching H-bridge, hereinafter referred to as AC forward bridge, having a DC input connected to the DC output of the single DC-to-DC converter and an AC output connected in parallel with the main AC output, according to a bypass path and in parallel with the bidirectional voltage-mode DC-to-AC converter and the low-frequency diode, the operating frequency of the AC forward bridge being less than 1kHz and preferably 400Hz or 50/60 Hz;

-control means for controlling said bidirectional voltage source DC-to-AC converter to deliver a sinusoidal AC voltage at said first AC output-input and for controlling said AC forward bridge to deliver a quasi-square AC forward current in phase with said sinusoidal AC voltage, said control means being capable of controlling said bidirectional voltage source DC-to-AC converter and said AC forward bridge such that the latter can operate simultaneously;

thus, when the instantaneous voltage between the terminals of the main AC output reaches a predetermined level, the low frequency forward AC bridge is switched on, the low frequency diode is reverse biased and non-conductive, and the DC-to-DC converter provides constant power directly to the load.

According to a preferred embodiment, the DC-to-AC power converter of the present invention is such that:

-selecting two closed switches (TPH, TNL; TPL, TNH) of the low frequency AC forward bridge in dependence on the polarity of the output AC voltage;

the DC-to-DC converter is designed to support a variable output voltage while delivering an almost constant power;

-the DC-to-DC converter is isolated and comprises, on the primary side of the transformer, an active clamp consisting of a main mosfet (mp) connecting the primary winding (TFO-P) of the transformer to a primary power supply (VIN +, VIN-) providing the main DC input, a resonant Capacitor (CRP) connected in parallel with the primary mosfet (mp) to allow the primary mosfet (mp) to operate at ZVT, and a second mosfet (maux) arranged to provide a voltage clamp over the main mosfet (mp) to protect the main mosfet (mp) from overvoltage;

-the DC-to-DC converter is isolated and comprises on the primary side of the transformer: a two-transistor forward converter primary stage consisting of two MOSFETs (MP1, MP2), a respective one end of each MOSFET being directly connected to one end of the primary winding (TFO-P) of the transformer and to a respective primary power supply terminal (VIN-, VIN +) providing the main DC input; -a resonant capacitor (CRP1, CRP2) connected in parallel with each MOSFET (MP1, MP2) respectively to allow said MOSFET (MP1, MP2) to operate at ZVT; and diodes (DAUX1, DAUX2) arranged to connect each MOSFET (MP1, MP2) to a primary power supply terminal (VIN +, VIN-) that is not directly connected to the primary power supply terminal (VIN-, VIN +) of the corresponding MOSFET (MP1, MP2), respectively;

-the DC-to-DC converter further comprises on the secondary side of the transformer: at least a first capacitor (CS1, CS2) establishing an AC connection to the secondary winding (TFO-S) of the transformer; a resonant inductance (LR) that can be reduced to a leakage inductance of the transformer; a rectifier Diode (DR) and a freewheel Diode (DLR) for rectifying a voltage generated at the secondary side of the transformer; and a secondary resonance Capacitor (CRS) connected in parallel with the rectifier Diode (DR); and a decoupling capacitor (CS3) connected in parallel with the terminal of the output DC voltage;

-the first capacitor (CS1), the resonant inductance (LR), the rectifier Diode (DR) and the freewheel Diode (DLR) are arranged such that during a magnetization phase of the transformer, the input voltage VIN of the converter, reflected to the secondary of the transformer as NVIN, charges the first capacitor (CS1) through the rectifier Diode (DR) and creates a resonance between the first capacitor (CS1) and the resonant inductance (LR), where N is the transformer turns ratio, the freewheel Diode (DRL) is non-conductive, and there is no overvoltage at the junction between the rectifier Diode (DR) and the freewheel Diode (DRL);

-the first capacitor (CS1), the resonant inductance (LR), the rectifier Diode (DR) and the freewheeling Diode (DLR) are arranged such that during a demagnetization phase of the transformer, a current flows from the charged first capacitor (CS1) through the resonant inductance (LR) and the freewheeling Diode (DRL) to the load, said current not only transferring the transformer magnetization energy, but at the same time transferring the energy stored in the first capacitor (CS1) during the magnetization phase;

-the output power (P) of the DC/DC converter_O) The output power (P) of the equivalent flyback converter is obtained by the following equation_FB) And (3) correlation:

P_O＝P_FBm, wherein,

wherein, V_INAnd V_OAn input voltage and an output voltage, respectively, of the DC/DC converter, M being a multiplication factor greater than 1, preferably greater than 2;

-the DC to AC power converter is bidirectional;

these low-frequency diodes are replaced by controlled switches;

-the controlled switches are MOSFETs, IGBTs or relays;

-the DC-to-DC converter is non-isolated.

A second aspect of the invention relates to an isolated DC-to-DC converter suitable for use in the above-described DC-to-AC power converter herein having a main DC input and a main single-phase AC output, the isolated DC-to-DC converter comprising: on the primary side of the transformer, an active clamp consisting of a main mosfet (mp) connecting the primary winding (TFO-P) of the transformer to a primary power supply (VIN +, VIN-) providing the main DC input, a resonant Capacitor (CRP) in parallel with the primary mosfet (mp) to allow the primary mosfet (mp) to operate at ZVT, and a Capacitor (CAUX) and a second mosfet (maux) arranged to provide a voltage clamp over the main mosfet (mp) to protect the main mosfet (mp) from overvoltage.

Drawings

Fig. 1 schematically represents an inverter topology according to the prior art.

Fig. 2 schematically shows the principle of an efficiency improved DC/AC power converter according to the invention.

Fig. 3 shows typical waveforms in the case of the power converter of fig. 2.

Fig. 4 illustrates a circuit topology for use in accordance with certain embodiments of the present invention.

Fig. 5 is the circuit topology of fig. 4 with variables that allow operational analysis.

Fig. 6 shows characteristic voltage and current waveforms in the three-phase operation of the circuit depicted in fig. 5.

Fig. 7 shows a topology according to another embodiment of the invention, wherein the input stage of the topology of fig. 5 has been replaced by the input stage of a two-transistor forward converter.

Detailed Description

The solution proposed by the present invention is presented in fig. 2.

The isolated DC/DC converter 5 transfers power from the DC input (typically 48Vdc) to the tank voltage (typically 400Vdc) in a constant power mode. The low frequency diode 2 allows current to flow to the tank capacitor 3. The DC/AC converter 6 (which is typically a full bridge converter) converts the DC voltage into an AC sine wave, typically a 50Hz or 60Hz sine wave.

During the time evolution of the output sine wave voltage, as long as the instantaneous output voltage between nodes LOUT and NOUT is sufficiently high (i.e. typically 200V), the low frequency "AC forward bridge" 7 can switch on and the D1 and D2 diodes 2 are automatically reverse biased (non-conducting) and then provide the constant power provided by the DC/DC converter 5 directly to the load.

The DC/AC converter only needs to provide the remaining/supplemental portion of the power and therefore operates at a lower power level and with lower losses.

It should be noted that the two switches (TPH/TNL or TNH/TPL) in the low frequency "AC forward bridge" 7 that are turned on are selected according to the polarity of the output voltage.

The advantage of this topology is that the efficiency of the DC/DC + AC forward bridge is much higher than the efficiency of the DC/DC + DC/AC path, since the AC forward bridge 7 operates at low frequency and does not require an output inductor. Because of the low switching frequency (typically 100Hz), the commutation losses are low and the bridge efficiency is only limited by the conduction losses in the switches and can be improved by placing the switches in parallel. Since the effective resistance is the resistance of one switch divided by the number of parallel switches, increasing the number of parallel switches can actually reduce conduction losses.

Working principle of AC forward bridge

Fig. 3 shows typical voltage and current waveforms of the converter presented in fig. 2.

The output voltage vout is set by the DC/AC converter 6 by methods well known to those skilled in the art of power electronics and inverters.

From t0 to t1, the AC forward bridge is disconnected and the converters 5 and 6 operate as in the prior art. Diodes D1 and D2 (diode 2) conduct.

At t1, the output voltage vout, equal to v (lout) -v (nout), is high enough to allow the AC forward bridge 7 to transfer power directly to the output; thus, the AC forward bridge 7 is turned on. When vout is positive, transistors TPH and TNL turn on during interval t1 to t 2.

During the interval t1 to t2, there are some relationships:

-vout is lower than Vtank and TPH and TNL are on, so diodes D1 and D2 are off and idc is equal to 0;

the inverter is directly generated by an isolated DC/DC converter that must maintain a constant power (demand) at its input. Thus, iforward vout represents constant power. This law determines the shape of iforward;

the output current must be supplied to the load, and therefore iac iout-inverter. This law determines the shape of iac.

As can be seen on fig. 3, during this time interval iac is much lower than iout, explaining why the total losses will decrease.

At t2, transistors TPH and TNL are turned off and the converter reverts to "normal operation", i.e. its operation is similar to that during t0 and t 1. During this interval, diodes D1 and D2 (diode 2) conduct again.

At T₀After/2, the next half cycle begins and vout is negative, except that transistors TPL and TNH are turned on instead of transistors TPH and TNL, the sequence of operation is similar to that described above.

DC/DC converter

Description of the invention

As explained above, the efficiency of the AC forward bridge 7 is limited only by its conduction losses and is therefore very high. Therefore, the overall inverter efficiency is mainly affected by the DC/DC converter efficiency. To support the architecture proposed in fig. 2, we have to implement a high efficiency DC/DC converter capable of supporting variable output voltages while delivering almost constant power. The present invention proposes a high efficiency converter that meets the above requirements. An example of such a transformer is described in US 7,778,046B1, but has some disadvantages as explained above. It is an object of the present invention to solve these disadvantages in a simple manner.

One embodiment of the present invention is a circuit topology as shown in fig. 4.

TFO-P and TFO-S are the primary and secondary windings of the transformer, respectively.

MP is the main primary MOSFET connecting the transformer to the primary source. CRP is a resonant capacitance placed in parallel with MP and allowing the primary MOSFET MP to operate at ZVT (zero voltage transition). CAUX and MAUX each provide a voltage clamp and protect MP from over-voltage. This part of the circuit is known in the art as "active clamping". Finally, CP is the primary side decoupling capacitor.

In this application, the term ZVT denoting "zero voltage switching" will be used instead of ZVS denoting "zero voltage switching", since switching is performed over the entire cycle (in order to reverse the magnetization current), rather than in short time intervals.

On the secondary side, capacitors CS1 and CS2 establish an AC connection to the transformer. The secondary transformer winding is TFO-S. LR is the resonant inductance which usually represents the leakage inductance of the transformer. The voltage generated at the secondary side of the transformer is rectified by DR (rectifying diode) and DRL (freewheeling diode). CRS is a secondary resonant capacitor connected directly across the diode DR. CS3 is an output decoupling capacitor.

It should be noted that the arrangement of CS1, CS2 and CS3 may be modified without modifying the principle of the proposed circuit.

The DC/AC converter of the invention has only one DC/DC converter, which is not the case in the prior art (the transformer of the DC/DC converter has e.g. several secondary stages). Therefore, the present invention has an advantage of reducing the number of electronic components.

Principle of operation

Fig. 5 shows a circuit, the operation of which will be further analyzed. LM is an inductance which simulates the magnetizing inductance of the transformer as seen from the primary side. Fig. 6 shows the corresponding waveforms.

Note that the values of CP (typically 10uF), CAUX (typically 2uF), and CS3 (typically 1uF) are very high, and the corresponding voltages across these capacitors (i.e., VIN, VOUT/N, and VOUT) are approximately constant. It is also contemplated that each MOSFET (MP, MAUX) has an intrinsic reverse-biased diode (not shown). When MP is turned off, the potential at D will increase, but is limited by the voltage VOUT/N on CAUX. Above this value, the intrinsic diode of MAUX will conduct.

The circuit operation is basically divided into 3 stages (see fig. 6).

-Stage 1:

in phase 1, the primary MOSFET voltage v (d) is zero. At the beginning of this phase, v (d) reaches zero naturally and relatively slowly, and thus the primary MOSFET MP can be turned on at zero voltage. Note that the gate voltages Vg and Vga of the MOSFETs are arbitrarily vertically shifted in fig. 6 for clarity. During this phase, the magnetizing current i (lm) rises linearly. It should be noted that the supply voltage VIN is reflected to the secondary of the transformer. This voltage is applied across CS1 and resonates with LR, which ensures that the voltage across CS1 is fixed equal to VIN x N at CMID, where N is the transformer turns ratio. For the flyback converter, the energy of the transformer is also stored during this phase, but charging capacitor CS1 allows some additional energy transfer and the available power density increases. Fig. 6 shows that the voltage v (cmid) on the CS1 capacitor is nearly constant during phase 1, with a small positive slope ripple. The charging current of CS1 is exactly i (dr). The current in the diode DR is part of the resonant sine wave and returns to 0 before the end of phase 1 (i.e., before the main primary MOSFET MP turns off). Thus, the diode DR is turned off at zero current. The diode DRL is not conducting in phase 1.

-Stage 2

The main primary MOSFET MP is switched off, the voltage v (d) rises rapidly according to the required power level, and the internal diode of the auxiliary MOSFET MAUX clamps the voltage v (d), which drops a little during phase 2. During this stage, the magnetizing current i (lm) decreases linearly to 0. It should be noted that capacitor CRP across MP limits the slew rate dV/dt of voltage v (d) and helps MP to turn off gently. Since capacitor CAUX clamps v (d), the current in diode DRL does not follow the magnetizing current at the beginning of phase 2. This takes time until the current in the resonant inductor LR increases. This corresponds to the time required for the current to flow completely to the secondary. Finally, I (DRL) decreases with the magnetizing current. During phase 2, the auxiliary mosfet maux may be turned on to ensure charge balance of capacitor CAUX. It should be remembered that the gate voltage v (ga) must be negative in order for MAUX to turn on, since it is a P-type MOSFET (according to fig. 5). Thus, both MOSFET drivers can be commanded with respect to the same potential (ground). At the end of this phase, the magnetizing current i (lm) reaches 0. At the same time, the current in the DRL also reaches 0, resulting in zero current switching of the DRL.

-Stage 3

The magnetizing current is reversed and resonance occurs between the magnetizing inductance LM and the equivalent resonant capacitor. The equivalent capacitor is a combination of CRP and CRS connected in parallel by a transformer. During this resonance, the voltage v (d) gently drops and reaches 0. This allows the next phase 1 to be started by switching the primary MOSFET MP at zero voltage.

According to an alternative embodiment of the invention shown in fig. 7, the input stage of the topology of fig. 5 is replaced by the input stage of a two-transistor forward (or two-transistor flyback) converter, which topology is known in the art. The main primary MOSFET MP is replaced by two primary MOSFETs MP1, MP2, and the active clamping component consisting of a clamping capacitor CAUX and an auxiliary MOSFET MAUX is replaced by two corresponding diodes DAUX1, DAUX 2. The resonant capacitance CRP placed in parallel with MP is replaced by resonant capacitances CRP1, CRP2 placed in parallel with primary MOSFETs MP1 and MP2, respectively. The operation of the above topology is very similar to that of the above topology (fig. 5).

Two-transistor forward converters are typically used to handle higher input voltages.

THE ADVANTAGES OF THE PRESENT INVENTION

Soft switching and compactness

Those skilled in the art recognize that the use of high frequencies in operating a DC/DC converter is a key factor in achieving compactness. However, the use of higher frequencies also means an increase in switching losses. From the description of the above phase, the circuit appears to be optimal in terms of switching losses, since both MOSFETs switch at zero voltage and both diodes are turned off (or reverse polarized) at zero current. The proposed circuit is therefore very suitable for constructing very compact converters.

Efficient power control and transmission

As schematically shown in fig. 2, the DC/DC converter 5 must maintain a constant current, which is an initial requirement, but its output switches between different voltages. Flyback converters, well known to those skilled in the art, can operate in discontinuous conduction mode and are therefore ideal circuits for this application because the output current is naturally controlled. A problem with flyback converters, however, is that they are limited to operating at several hundred watts, since the transformer must store the total transferred energy during the first portion of the switching cycle to restore it to the output over the second portion of the switching cycle.

In the present invention, during phase 2, the magnetizing current reflected to the secondary side of the transformer (which flows through diode DRL) causes the stored transformer magnetizing energy to be transferred to the output. This behavior is very similar to that of a flyback converter. However, the difference of the proposed converter compared to a flyback converter is that the output winding of the transformer is not connected in parallel to the output through a DRL, but here in series with CS 1. This means that both the energy stored in the magnetizing inductance of the transformer and the energy stored in CS1 are transferred to the output simultaneously. It can be seen that the output power (P) of the proposed DC/DC converter_O) The output power (P) of the equivalent flyback converter is obtained by the following equation_FB) And (3) correlation:

P_O＝P_FBm, wherein,

wherein, V_INAnd V_ORespectively an input voltage and an output voltage of the DC/DC converter. The multiplication factor M is greater than 1. Typical values for M are even greater than 2. Thus, for this application, the proposed converter has the same advantages as a flyback converter, but is able to deliver at least twice the power under the same conditions.

Constant voltage limitation for semiconductors

The proposed circuit has very special and interesting properties in terms of semiconductor maximum stress voltage:

operating peak voltage of DRL and DR V_O。

Operating peak voltages of-MP and MAUX are V_O/N。

Thus, the peak operating voltages and V of all semiconductors_IN Is irrelevant. For a large input voltage range this is the ideal case, since the switches are optimally used independently of the input voltage.

This is a very difficult property for a DC/DC converter.

Two-way operation

It should be noted that DR and DRL may be replaced by controlled switches such as MOSFETs, IGBTs, relays, etc. that are controlled simultaneously with MP and MAUX, respectively. In this case, the converter may operate in a bidirectional mode and may transfer power from right to left.