阅读说明：本技术 改进效率的包络跟踪调制器的线性放大器 (Linear amplifier for envelope tracking modulator with improved efficiency ) 是由 杰勒德·温彭尼 于 2014-01-31 设计创作，主要内容包括：本申请涉及改进效率的包络跟踪调制器的线性放大器。本发明揭示一种包络跟踪电源,其经布置以产生依赖于参考信号的调制供电电压,所述包络跟踪电源包括用于跟踪所述参考信号的低频变化的第一路径及用于跟踪所述参考信号的高频变化的第二路径,所述第二路径包含线性放大器,其中所述线性放大器的输出包括连接到高频输出的电流源及电流吸收器,在所述高频输出处进一步提供DC补偿电流。(The present application relates to a linear amplifier of an envelope tracking modulator with improved efficiency. An envelope tracking power supply arranged to generate a modulated supply voltage dependent on a reference signal, the envelope tracking power supply comprising a first path for tracking low frequency variations of the reference signal and a second path for tracking high frequency variations of the reference signal, the second path comprising a linear amplifier, wherein an output of the linear amplifier comprises a current source and a current sink connected to a high frequency output, a DC compensation current being further provided at the high frequency output.)

1. An envelope tracking power supply arranged to generate a modulated supply voltage dependent on a reference signal, the envelope tracking power supply comprising:

a first path for tracking low frequency variations of the reference signal; and

a second path for tracking high frequency variations of the reference signal, the second path comprising:

a current source (250) and a current sink (252) connected to the high frequency output,

a further voltage supply for generating a second supply voltage for generating a compensation current provided at the high frequency output, an

A power difference between the power dissipated in the current source and the current sink is sensed (820), and the sensed power difference is integrated (816) to control the further voltage supply.

2. The envelope tracking power supply of claim 1 wherein the compensation current is determined in dependence on an input voltage waveform.

3. The envelope tracking power supply of claim 2 wherein the compensation current is determined in dependence on a difference between the input voltage waveform and an average of the input voltage waveform.

4. The envelope tracking power supply of claim 1 wherein the further voltage supply comprises a switched mode converter (810).

5. The envelope tracking power supply of claim 1 wherein the compensation current is selected to minimize the power dissipated in the current source (250) and the current sink (252), or wherein the compensation current is provided by an inductor connected between the further voltage supply and the high frequency output.

6. The envelope tracking power supply of claim 1 wherein sensing the power difference comprises measuring a supply voltage used to generate the compensation current, the output voltage, the source current and the sink current.

7. An envelope tracking power supply arranged to generate a modulated supply voltage dependent on a reference signal, the envelope tracking power supply comprising a first path for tracking low frequency variations of the reference signal and a second path for tracking high frequency variations of the reference signal, the second path comprising a current source (250) and a current sink (252) connected to the high frequency output, a compensation current being further provided at the high frequency output, wherein a target compensation current (808) is determined dependent on a difference between an input voltage waveform and an average of the input voltage waveform.

8. The envelope tracking power supply of claim 7 wherein the error between the target compensation current and the measured compensation current is integrated and used to control a switch mode converter to generate the second supply voltage to generate the compensation current.

9. The envelope tracking power supply of claim 7 wherein the second path further comprises a further voltage supply to generate a second supply voltage to generate the compensation current provided at the high frequency output.

10. An RF amplifier or a mobile device for a mobile communication system, or an infrastructure element for a mobile communication system, comprising an envelope tracking power supply according to any of claims 1 to 9.

Technical Field

The present invention relates to an envelope tracking modulated power supply suitable for radio frequency power amplifier applications. The present disclosure relates to such power supplies, among other things, where a reference signal is used as an input to a low frequency path and a high frequency path, and where each path produces a separate output that is combined to form a supply voltage.

Background

Envelope tracking power supplies for radio frequency power amplifiers are well known in the art. The reference signal is typically generated based on the envelope of the input signal to be amplified. The envelope tracking power supply generates a supply voltage for the power amplifier that tracks the reference signal.

Fig. 1 shows a prior art Envelope Tracking (ET) modulator architecture in which a frequency divider 12 is used to divide an incoming envelope reference signal on line 10 into a High Frequency (HF) path signal on line 14 and a Low Frequency (LF) path signal on line 16. Divider 12 may include a low pass filter 18 in the low frequency path and a high pass filter 20 in the high frequency path. The signal in the LF path on line 16 is amplified by a high efficiency switch mode amplifier 22 and the signal in the HF path on line 14 is amplified by a broadband linear amplifier 24. A frequency selective combiner 26 is used to combine the amplified signals in the LF and HF paths. In fig. 1, the combiner 26 is illustrated as including a low frequency combining element (and high frequency blocking element) 28 in the low frequency path, and a high frequency combining element (and low frequency blocking element) 30 in the high frequency path. The combined signal on line 32 from combiner 26 provides a feed to load 34, load 34 being illustrated as a resistor for purposes of example. In a typical application, the load is a Power Amplifier (PA) and the reference signal is derived from an input signal to be amplified by the PA.

An example of a Power amplifier system incorporating a Power supply architecture such as that illustrated in fig. 1 may be found in "band separation and Efficiency optimization in Linear-Assisted Switching Power amplifiers" (band separation and Efficiency optimization) in madura (Yousefzadeh) et al, united states, IEEE Power Electronics conference (IEEE Power Electronics specialities conference), 2006.

Fig. 2 shows an alternative prior art arrangement in which the frequency selective combiner 26 is an inductor-capacitor (LC) combiner. The low frequency combining element is an inductor 28a and the high frequency combining element is a capacitor 30 a. In this arrangement, a feedback path 36 carries the signal on line 32 from the combiner (or modulator) output to the input of the linear amplifier 24. The signal on feedback path 36 is subtracted from the signal in the high frequency path on line 14 by subtractor 38 to provide an input to linear amplifier 24. The inclusion of the feedback path 36 will achieve improved tracking accuracy compared to the arrangement of figure 1.

An example of a Power amplifier system incorporating a Power supply architecture such as that illustrated in fig. 2 may be found in "Efficiency optimization of Linear auxiliary Switching Power converters for Envelope Tracking in RF Power Amplifiers" (Efficiency optimization in Linear-Assisted Switching Power converters) "by yoosefzadeh et al, [ IEEE Symposium on Circuits and Systems for Circuits and Systems, 2005.

It is an object of the present invention to provide an envelope tracking modulated power supply which is improved compared to the prior art, e.g. the arrangements of figures 1 and 2.

Disclosure of Invention

The present invention provides an envelope tracking power supply arranged to generate a modulated supply voltage dependent on a reference signal, the envelope tracking power supply comprising a first path for tracking low frequency variations of the reference signal and a second path for tracking high frequency variations of the reference signal, the second path comprising a linear amplifier, wherein an output stage of the linear amplifier comprises a current source and a current sink connected to the high frequency output, a DC compensation current being further provided at the high frequency output.

The DC offset current may be selected to minimize power dissipated in the output stage of the linear amplifier.

The DC compensation current may originate from a further voltage supply which is lower than the output stage voltage supply.

The DC compensation current may be provided via an inductor connected between the further power supply and the high frequency output.

The envelope tracking power supply may further comprise sensing a power difference in the output, and integrating the sensed power difference to control the switch mode converter to generate the second supply voltage to generate the DC compensation current. Sensing the power difference may include measuring a supply voltage used to generate the DC offset current, an output voltage, a source current, and a sink current.

The target DC compensation current may be determined in dependence on the difference between the input voltage waveform and the halved sum of the maximum and minimum voltage levels of the input waveform voltage. An error between the target DC compensation current and a measured DC compensation current may be integrated and used to control a switch mode converter to generate a second supply voltage to generate the DC compensation current.

An RF amplifier may include an envelope tracking power supply.

A mobile device for a mobile communication system may include an envelope tracking power supply.

An infrastructure element for a mobile communication system may include an envelope tracking power supply.

The invention further provides a method for an envelope tracking power supply arranged to generate a modulated supply voltage dependent on a reference signal, the method comprising providing a first path for tracking low frequency variations of the reference signal and providing a second path for tracking high frequency variations of the reference signal, the second path comprising a linear amplifier, wherein an output stage of the linear amplifier comprises a current source and a current sink connected to the high frequency output, the method further comprising providing a DC compensation current at the high frequency output.

Drawings

The invention will now be described, by way of example, with reference to the accompanying drawings, in which:

FIG. 1 illustrates a prior art envelope tracking modulated power supply having a high frequency path and a low frequency path;

FIG. 2 illustrates a prior art envelope tracking modulated power supply incorporating feedback in the high frequency path;

FIG. 3 illustrates a modified implementation of the output of a linear amplifier according to the arrangement of FIG. 1 or FIG. 2;

fig. 4(a) to 4(c) illustrate the current flow in the arrangement of fig. 3;

FIG. 5 illustrates an implementation of the output of a linear amplifier in the arrangement of FIG. 1 or FIG. 2, according to an embodiment of the invention;

6(a) to 6(c) illustrate the current flow in the arrangement of FIG. 5;

FIGS. 7(a) and 7(b) illustrate waveform plots in the arrangements of FIGS. 3 and 5;

FIG. 8 illustrates an implementation of the output of a linear amplifier in the arrangement of FIG. 1 or FIG. 2 according to an exemplary embodiment of the invention; and

fig. 9 illustrates an implementation of the output of a linear amplifier in the arrangement of fig. 1 or 2 according to another exemplary embodiment of the invention.

Detailed Description

In the following description, the invention is described with reference to exemplary embodiments and implementations. The invention is not limited to the specific details of any arrangement as set forth which are provided for the purpose of understanding the invention.

Embodiments of the present invention are applicable to different feedback architectures of linear amplifiers in the high frequency path. The invention and its embodiments are not limited to a particular feedback arrangement in the high frequency path. For example, in the foregoing description of fig. 2, an arrangement is illustrated in which a linear amplifier receives feedback from the output of a combiner. For example, the invention is also applicable to arrangements in which a linear amplifier receives feedback from the output of the linear amplifier at the input to the combiner and in which the path containing the linear amplifier does not include a high frequency filter (e.g., filter 20 in fig. 2), the linear amplifier path receiving a full spectrum reference signal.

In general, in a hybrid envelope tracking modulator (i.e., an architecture that uses a switched mode amplifier and a linear amplifier) as illustrated in fig. 2, a significant proportion of the total modulator power dissipation occurs in the output stage of the linear amplifier.

This may be understood with reference to fig. 3, which illustrates an exemplary implementation of the linear amplifier 24 class B output stage. As illustrated, the current source 250 is connected at the supply voltage V_SUPPLYAnd a common node 254, and a current sink 252 connected between the common node 254 and electrical ground V_GNDIn the meantime. Instantaneous source current I_SRCFlows in the current source element 250, and the instantaneous current I_SNKFlows in the current sink element 252. At any given moment, current flows in the source device 250 or sink device 252, and the current in the inactive devices is zero. Forming an output voltage V at node 254_EA. The combiner capacitor 30a of fig. 2 is illustrated as being connected between node 254 and the output of the combiner. Current I_EAFlows in the combiner capacitor 30 a.

For purposes of example, the arrangement of fig. 3 shows a feedback path 40, which represents feedback from the output of the linear amplifier (prior to the combiner) to the input of the linear amplifier. The feedback is not described in more detail herein, as it does not form part of the present invention. The current in the feedback path is assumed to be low enough to be ignored.

No DC current may flow through the combiner capacitor 30 a. Thus, in the prior art arrangement of fig. 3, the average source current I from the current source 250_SRCMust be equal to the average sink current I from the current sink 252_SNKThe value of (c).

In general, the desired modulator output voltage provided by linear amplifier 24 may typically exhibit significant asymmetry, and this in turn causes the output current I of linear amplifier 24_EAIs asymmetric.

This is illustrated by the waveforms of FIG. 4(a), which shows the output current I_EAPlot against time. A current above zero level 302 represents an output positive current flowing in source transistor 250 and a current below zero level 302 represents an output negative current flowing in sink transistor 252. The combined source and sink current represents the output current I_EA。

Average source current I_SRCAnd sink current I_SNKThe values of each of which are equal, as shown in fig. 4(b) and 4(c), which show plots of source and sink currents versus time.

Line 304 in fig. 4(b) shows the average current in the source device 250, and line 306 in fig. 4(c) shows the average current in the sink device 252. The average current in the source device 250 is equal to the average current in the sink device 252.

However, in the example as shown, the power dissipated in the upper device (current source 250) is much greater than the power dissipated in the lower device (current sink 252). This power dissipation inequality is due to the waveform asymmetry and the much higher voltage induced across the upper (source) device.

It can therefore be seen that the necessity of having the average sink current equal to the average source current for the output topology of figure 3 is disadvantageous.

According to a preferred embodiment of the present invention, an additional voltage supply is used to add a DC (or low frequency) compensation current to the output node of the linear amplifier 24 via an inductor. The average source current is no longer required to be equal to the average sink current.

Fig. 5 shows such a modified topology. The arrangement of fig. 3 is modified such that the inductor 256 is comprised in the second supply voltage V_SUPPLY2And node 254. Inductor 256 provides a compensation current I_OSCompensating current I_OSFrom voltage source V_SUPPLY2Flows in the inductor 256.

The instantaneous current in the current source 250 is modified to I_SRC'And the instantaneous current in the current sink 252 is I_SNK'. Output current I_EAFlows in output capacitor 30a and forms an output voltage V at node 254_EA。

The waveform of FIG. 6(a) shows the output current I of the linear amplifier 24_EAWhich is the same as the output current shown in fig. 4 (a). Thus, according to the invention, the output current I of the linear amplifier_EAAnd is not changed. As illustrated in fig. 6, the portion of the output current above line 602 is provided by source transistor 250. Part of the output current under line 602And is provided by sink transistor 252.

FIGS. 6(b) and 6(c) show modified source currents I from current source element 250 and current sink element 252, respectively_SRC'And sink current I_SNK'. In the example shown, the modified source current I_SRC'Is reduced by compensating the current Ios and a modified sink current I_SNK'Is increased by the offset current Ios. Lines 604 and 606 in respective fig. 6(b) and 6(c) represent modified average currents flowing in respective source and sink transistors.

As illustrated, by comparing fig. 4(b) and 4(c) with fig. 6(b) and 6(c), the additional compensation current I supplied through the inductor 256_OSIs effected by_OSAverage source current from I_SRCIs reduced to I_SRC'And through I_OSAverage sink current from I_SNKIncrease to I_SNK'. This reduces the power dissipated in the current source 250 and increases the power dissipated in the current sink 252.

FIG. 7(a) shows dissipations 702, 704 in the output stage source current device 250 and sink current device 252, respectively, and total dissipation 706 with compensation current I for the arrangement of FIG. 5_OSAnd so on.

It can be seen that for the particular waveforms illustrated, the minimum dissipation in fig. 7(a) is about 20% less than the dissipation without the compensation current. This dissipation difference strongly depends on the waveform asymmetry and is larger for more asymmetric waveforms.

Instantaneous power dissipation in the source and sink output devices 250 and 252 cannot be easily measured directly, but both the average current and average output voltage through the source and sink devices 250 and 252 can be easily measured. Thus, it is possible to calculate 'sensed' power as a proxy for dissipated power using these average parameters.

Fig. 7(b) shows the sensed power for the source device 710, sink device 708, and total sensed power 712.

Inductor 250 ideally has zero DC resistance, and therefore the DC voltage at both terminals of inductor 250 is the same.

Referring to FIG. 5, the sensed source power may be calculated as:

avg(V_SUPPLY-V_EA)×avg(I_SRC')

the sensed absorbed power may be calculated as:

avg(V_EA)×avg(I_SNK')

wherein:

V_SUPPLYa supply voltage applied to a feed inductor;

V_EAan output voltage of the stage;

avg(I_SRC') -average source current; and

avg(I_SNK') Average absorption current.

The minimum of the total sensed power (as shown in fig. 7) occurs at the same compensation current value as the minimum dissipated power, thus minimizing the sensed power maximizes the efficiency of the supply modulator.

Additional voltage supply V in FIG. 5_SUPPLY2Is assumed to be generated using a high efficiency power converter and the power loss in the feed inductor 256 is assumed to be minimal.

FIG. 8 shows a direct technique for generating the compensation current, where a negative feedback loop may be used to integrate the error to the second supply V_SUPPLY2Is made small adjustments to minimize the difference between the two sensed powers while minimizing the total sensed power.

Fig. 8 shows the output stage of an error amplifier, which includes a current source 250, a current sink 252, a combination capacitor 30a and a DC current compensation feed inductor 256. Supply voltage V_SUPPLY2Is provided by a switched mode converter 810, the switched mode converter 810 being connected to a supply voltage V indicated by reference numeral 814_SUPPLY。

The input to the switch mode converter 810 is provided by an integrator 816. The input to the integrator 816 is provided by a signal processing block 818, the signal processing block 818 based on the second supply voltage Vsupply2, the average output voltage Vea and the source current I_SRC'And sink current I_SNK'Is generated as a mean value ofThe sensed power difference signal on line 818 at the input of integrator 816.

An indirect method of controlling the compensation current exploits the fact that the required compensation current depends on the asymmetry of the waveform. If the waveform is symmetrical, the average voltage is in the middle of the minimum and maximum values of the waveform. If the average voltage is less than a value intermediate the minimum and maximum values of the waveform, then a positive compensation current is required to minimize the output stage power dissipation. Similarly, if the average voltage is greater than the value between the minimum and maximum values of the waveform, a negative compensation current is required to minimize the output stage dissipation.

FIG. 9 shows a control loop for implementing this indirect concept.

The control loop includes a current source 250 and a current sink 252 of the output stage, a combining capacitor 30a, and a DC current compensation feed inductor 256. Inductor 256 is connected to node 254 via a current sense resistor 800.

The supply voltage Vsupply2 is provided by a switched mode converter 802, the switched mode converter 802 being connected to a supply voltage V indicated by reference numeral 804_SUPPLY。

The input to the switch mode converter is provided by integrator 806. The first input to the integrator is provided by subtractor 808, subtractor 808 providing the difference on line 812 between the voltage (a value intermediate the maximum and minimum values of the input waveform (equal to (Vmax + Vmin)/2)) and the input waveform Vin 810 to give a voltage at the first input to integrator 806 representing the compensation current target on line 814. A second input to the integrator 806 is provided by a voltage source 816, the voltage source 816 measuring the current in the resistor 800 and providing a voltage representative of the compensation current.

The current compensation target on line 814 is set to the difference between the average waveform voltage and the median waveform voltage, as described above. The error between the target compensation current and the measured compensation current is integrated by an integrator 806 and used to control a switch mode converter 802, the switch mode converter 802 generating a second supply voltage V_SUPPLY2Second supply voltage V_SUPPLY2The compensation current is supplied to the linear amplifier output stage via inductor 256.

The generation of the compensation current and the second power supply may be accomplished in a variety of ways (both indirectly and directly), and the invention is not limited to any particular technique.

As discussed above, the present invention may be applied to the output of a linear amplifier (e.g., the linear amplifier of fig. 1 or fig. 2) in the correction path of a modulated power supply.

Such a modulated power supply may be used to provide modulated power to an RF amplifier, which may include the load of fig. 1 or fig. 2.

RF amplifiers are used in mobile communication systems, wireless devices, and wireless infrastructures.

The present invention and its embodiments relate to the application of Envelope Tracking (ET) to Radio Frequency (RF) power amplifiers, and are applicable to a wide range of implementations including cellular handsets, wireless infrastructures at high to microwave frequencies, and military power amplifier applications.

The invention has been described herein by way of example with reference to the embodiments. The invention is not limited to the described embodiments nor to a specific combination of features in the embodiments. Modifications may be made to the embodiments within the scope of the invention. The scope of the invention is defined by the appended claims.