阅读说明：本技术 开关电源装置 (Switching power supply device ) 是由 高桥直也 久保谦二 小野琢磨 于 2018-04-12 设计创作，主要内容包括：本发明提供一种电压控制的分辨率高的开关电源装置。本发明的开关电源装置具备：电力转换电路,其包含开关元件而构成；以及控制电路部,其根据电力转换电路的输出电压指令值和输出电压值,向开关元件输出驱动脉冲,控制电路部根据基于输出电压指令值和输出电压值的操作量的运算值与操作量的设定值的差分,使开关周期变化,输出驱动脉冲。(The invention provides a switching power supply device with high voltage control resolution. The switching power supply device of the present invention includes: a power conversion circuit including a switching element; and a control circuit unit that outputs a drive pulse to the switching element based on the output voltage command value and the output voltage value of the power conversion circuit, wherein the control circuit unit changes a switching cycle based on a difference between an operation value of an operation amount based on the output voltage command value and the output voltage value and a set value of the operation amount, and outputs the drive pulse.)

1. A switching power supply device is characterized by comprising:

a power conversion circuit including a switching element; and

a control circuit unit that outputs a drive pulse to the switching element based on an output voltage command value of the power conversion circuit and an output voltage value of the power conversion circuit,

the control circuit unit outputs a drive pulse by changing a switching cycle in accordance with a difference between a calculated value of an operation amount based on the output voltage command value and the output voltage value and a set value of the operation amount.

2. Switching power supply unit according to claim 1,

the on-time of the drive pulse is taken as the operation amount.

3. Switching power supply unit according to claim 1,

the power conversion circuit includes:

a 1 st switching branch connecting the 1 st switching element and the 2 nd switching element in series; and

a 2 nd switching branch connecting the 3 rd switching element and the 4 th switching element in series,

a full-bridge circuit in which a direct current terminal is formed between both ends of the 1 st switching leg and the 2 nd switching leg, and an alternating current terminal is formed between a series connection point of the 1 st switching element and the 2 nd switching element and a series connection point of the 3 rd switching element and the 4 th switching element,

setting a phase difference between the drive pulse for the 1 st switching leg and the drive pulse for the 2 nd switching leg as the operation amount.

4. Switching power supply unit according to claim 2 or 3,

the control circuit unit includes a unit for generating a clock signal,

the set value is a natural number multiple of the clock period.

5. Switching power supply unit according to claim 2 or 3,

the control circuit unit causes the amount of change in the switching period to be proportional to the difference.

6. Switching power supply unit according to claim 5,

the control circuit unit includes a means for detecting an input voltage and/or an output voltage of the power conversion circuit,

and changing a proportionality coefficient for determining the variation of the switching period according to the input voltage and/or the output voltage.

7. Switching power supply unit according to claim 2 or 3,

in the power conversion circuit, the control circuit unit includes a unit that generates a clock signal,

the amount of change in the switching period is calculated so that the switching period becomes an even number of clocks.

Technical Field

The present invention relates to a switching power supply device.

Background

In recent years, digital control systems have been widely used in place of conventional analog control systems in order to meet the demands for smaller size, higher performance, and higher functionality of switching power supply devices. In the digital control system, the digital control system includes an a/D converter sampling a voltage value to be controlled, a digital controller calculating an operation amount from the voltage value, a counter that increments and decrements in synchronization with a clock signal, a comparator that compares a count value of the counter with the operation amount to determine a drive pulse width and a phase of a switching element, and the like. For example, in PWM (pulse width modulation) control, the voltage of the control target is controlled by changing the on duty (time ratio of logic "H") of the drive pulse output from the comparator by changing the operation amount. In addition, the respective functions are also included in the digital controller in many cases.

In such a digital control method, since the minimum variation width of the operation amount is limited to a time of 1 cycle of the clock signal, there is a problem that the control resolution of the voltage is lowered compared to the analog control method. On the other hand, although a method of increasing the control resolution by increasing the frequency of the clock signal can be considered, the increase in the clock frequency has problems of an increase in power consumption and an increase in the cost of the controller.

Therefore, patent document 1 discloses the following method: the drive circuit includes a pulse smoothing circuit for smoothing a digital pulse output from a digital controller and converting the digital pulse into a smoothing voltage having a continuous value, a sawtooth wave generating circuit, and a comparison circuit, and generates a drive pulse by inputting the sawtooth wave voltage and the smoothing voltage to the comparison circuit. Thus, since the operation amount is a continuous value that is not limited by the clock frequency, the control resolution can be improved even if a low-cost digital controller with a low clock frequency is used.

Disclosure of Invention

Problems to be solved by the invention

However, the method described in patent document 1 has a problem in that the number of components, cost, and mounting area increase because a pulse smoothing circuit, a sawtooth wave generating circuit, and a comparator circuit need to be newly provided. In addition, since the smoothing is performed by the pulse smoothing circuit, there is a problem that the response of the control is slightly delayed.

As described above, the conventional digital control switching power supply device has a disadvantage that it is necessary to increase the cost and reduce the performance in order to improve the resolution of the control.

The present invention has been made in view of such circumstances, and an object thereof is to provide a switching power supply device with high resolution of voltage control without providing an additional circuit and without delaying the response of control.

Means for solving the problems

The switching power supply device of the present invention includes: a power conversion circuit including a switching element; and a control circuit unit that outputs a drive pulse to the switching element based on the output voltage command value and the output voltage value of the power conversion circuit, wherein the control circuit unit changes a switching cycle based on a difference between an operation value of an operation amount based on the output voltage command value and the output voltage value and a set value of the operation amount, and outputs the drive pulse.

ADVANTAGEOUS EFFECTS OF INVENTION

According to the present invention, in a digital control type switching power supply device, it is possible to improve the resolution of PWM control and phase shift control without using a high-performance digital processor or providing an additional circuit and without delaying the response. Thus, a switching power supply device with high resolution of voltage control, low cost, high response speed, and small voltage fluctuation can be provided.

Drawings

Fig. 1 is a configuration diagram of a power supply system in a vehicle 1 represented by a hybrid vehicle.

Fig. 2 is a circuit block diagram of the DC-DC converter 5 according to embodiment 1.

Fig. 3 is an explanatory diagram showing a difference in the on duty ratio variation width between the comparative example and embodiment 1.

Fig. 4 is a detailed block circuit diagram of the switching cycle calculation unit 26 of the present embodiment.

Fig. 5 is a circuit block diagram of the DC-DC converter 5 according to embodiment 2.

Fig. 6 shows waveforms of on/off states of the drive pulses Ps1 to Ps4, an output voltage (N1 voltage) of the switching circuit 31, a primary winding N1 current, and a smoothing inductor Lo current of the switching elements S1 to S4 according to embodiment 2.

Fig. 7(a) is a flow chart of the current path of the power conversion circuit 30 corresponding to the period of fig. 6 (a).

Fig. 7(b) is a flow chart of the current path of the power conversion circuit 30 corresponding to the period of fig. 6 (b).

Fig. 7(c) is a flow chart of the current path of the power conversion circuit 30 corresponding to the period of fig. 6 (c).

Fig. 7(d) is a flow chart of the current path of the power conversion circuit 30 corresponding to the period of fig. 6 (d).

Detailed Description

Hereinafter, embodiments of the present invention will be described with reference to the drawings.

Fig. 1 is a configuration example of a power supply system in a vehicle 1 represented by a hybrid vehicle. The vehicle 1 includes a main unit battery 2, an inverter 3, a motor 4, a DC-DC converter 5, an auxiliary unit battery 6, and auxiliary units 7. The inverter 3 converts dc power stored in the host battery 2 into ac power to drive the motor 4.

The DC-DC converter 5 converts the voltage of the DC power stored in the main engine battery 2 by switching control of a semiconductor device or the like, and supplies the converted DC power to the auxiliary engine battery 6 and the auxiliary machines 7. The present embodiment is preferably applied to a switching power supply device such as the DC-DC converter 5. As an application example of the DC-DC converter 5, embodiment 1 and a second embodiment will be described.

(embodiment 1)

Embodiment 1 of the present invention will be described with reference to fig. 2 to 4. In the present embodiment, when a non-isolated step-down converter controlled by PWM (pulse width modulation) is used as the DC-DC converter 5, the resolution of voltage control is improved.

Fig. 2 is a circuit block diagram of the DC-DC converter 5 of embodiment 1. The DC-DC converter 5 shown in fig. 2 includes a power conversion circuit 10 and a control circuit 20.

The power conversion circuit 10 includes a switching element 11, a diode 12, an inductor 13, an input smoothing capacitor 14, and an output smoothing capacitor 15, and the control circuit 20 performs power conversion by driving the switching element 11 with a drive pulse P1 output by PWM control.

The power conversion circuit 10 is a so-called step-down chopper that steps down an input voltage Vi between input terminals T1 to T2 and outputs an output voltage Vo between output terminals T3 to T4. The output voltage of the step-down chopper is denoted by Vo ═ Vi × Don, and Vo can be adjusted by changing Don. Here, Don is an on duty ratio, that is, a ratio of an on period Ton of the switching element to a switching period Tsw, and has a relationship of Don ═ Ton ÷ Tsw. In the normal PWM control, Vo is adjusted by controlling Ton while keeping the switching period Tsw constant.

The control circuit 20 includes a clock signal generation unit 21 and a DSP (digital signal processor) 22 that performs calculation in synchronization with the clock signal, and calculates and outputs an appropriate drive pulse P1 so that the output voltage Vo approaches a predetermined voltage command value Vref.

The DSP22 includes an a/D conversion unit 23, an on time operation unit 24, a less than 1-clock discard processing unit 25, a switching period operation unit 26, an on time setting register 27, a switching period setting register 28, and a drive pulse generation unit 29.

The a/D conversion unit 23 samples the analog voltage of the output voltage Vo, performs a/D conversion of the sampled analog voltage into a digital value, and outputs the converted value to the on-time calculation unit 24.

The on-time calculation unit 24 calculates the on-time Ton of the drive pulse P1 as the operation amount from the deviation between the a/D conversion value of the output voltage Vo and the output voltage command value Vref. The on-time calculating means 24 is constituted by PI (proportional integral) control, for example.

The discard processing unit 25 of less than 1 clock converts the on time calculation value into a clock signal cycle unit, and outputs the on time setting value set in the on time setting register 27.

The switching period calculation unit 26 calculates a switching period from a difference between the on time calculation value and the on time setting value, and outputs the switching period to the switching period setting register 28. Specifically, the switching cycle calculating means 26 calculates the set value of the switching cycle by adding the change amount Δ T of the switching cycle to the basic switching cycle T and multiplying the difference by a scaling factor.

The on-time setting register 27 and the switching cycle setting register 28 hold the set values of the on-time and the switching cycle, and output the set values to the drive pulse generating unit 29 for each predetermined control cycle Tc.

Although not shown, the drive pulse generating unit 29 includes a counter that is incremented/decremented in synchronization with a clock signal and resets a count value every predetermined reset period, and a comparator that outputs a PWM (pulse width modulation) pulse by inputting the count value and an operation amount of the counter and comparing them. In the present embodiment, the set value of the switching period setting register 28 is input as the reset period, and the set value of the on-time setting register is input as the operation amount. The above is the operation and configuration of the DC-DC converter 5.

Next, the effects of the present embodiment will be described with reference to fig. 3. Fig. 3 is an explanatory diagram showing a difference in the on duty ratio variation width between the comparative example and embodiment 1.

Column (a) of fig. 3 shows changes in the on-time Ton, the switching period Tsw, and the on-duty Don when the on-time Ton, which is the operation amount, is changed by 1 clock amount in the PWM control of the switching power supply device of the comparative example.

Column (b) of fig. 3 shows changes in the on-time Ton, the switching period Tsw, and the on-duty Don when the on-time Ton is changed by 1 clock in the PWM control to which the switching power supply device of the present embodiment is applied.

In the column (a) of fig. 3, since the switching period Tsw is made constant, the minimum variation width of the on duty Don is determined depending on 1 clock width of the on time Ton. In contrast, in the column (b) of fig. 3, the switching period Tsw is also changed according to the magnitude of the difference between the calculated value of the on time Ton and the set value.

Therefore, the on duty Don is determined by both the on time Ton and the switching period Tsw, so that a plurality of values of the on duty Don can be taken for a certain on time Ton. As can be seen by comparing the on duty Don in fig. 3(a) and 3(b), the on duty Don can be finely controlled, and the resolution can be improved.

The specific reason why the resolution of the on duty Don is improved when the switching period Tsw is changed is as follows.

First, if Δ D1 is set as the Don variation amount when Ton is varied by 1 clock, this is expressed by the following equation:

ΔD1＝(Ton+1)÷Tsw-Ton÷Tsw＝1÷Tsw。

on the other hand, if Δ D2 is set as the Don variation amount when Tsw is varied by 1 clock, this is expressed by the following equation:

ΔD2＝Ton÷Tsw-Ton÷(Tsw+1)＝(Ton÷Tsw)×1÷(Tsw+1)。

here, in general, the switching period Tsw is sufficiently larger than 1 clock cycle, Tsw > >1, and thus can be approximated as:

ΔD2≒(Ton÷Tsw)×1÷Tsw＝Don×1÷Tsw＝ΔD2＝Don×ΔD1。

finally, the amount of change in the on duty Don when the switching period Tsw is changed by 1 clock is Don times larger than the amount of change in the on duty Don when the on time Ton is changed by 1 clock, and the resolution is improved by 1 ÷ Don times. Here, since the on duty Don is determined by the input voltage Vi and the output voltage Vo, the optimum amount of change in the switching period varies depending on the input/output voltage condition in order to improve the resolution.

Therefore, the switching cycle calculation means 26 may have the configuration shown in fig. 4, for example. Fig. 4 is a detailed block circuit diagram of the switching cycle calculation unit 26 of the present embodiment. In fig. 4, the switching cycle computing means 26 computes a scaling factor from the input voltage Vi and the output voltage Vo, and determines the switching cycle set value as the amount of change in the switching cycle by multiplying the difference by the scaling factor. Thus, the resolution can be improved regardless of the input/output voltage conditions.

As described above, according to embodiment 1, the on duty Don can be finely controlled by changing the switching period Tsw in accordance with the difference between the operation amount of the on time Ton and the set value.

(embodiment 2)

Embodiment 2 of the present invention will be described with reference to fig. 5 to 7. In the present embodiment, when an insulated step-down converter of a phase shift control method is used as the DC-DC converter 5, the resolution of voltage control is improved. The configuration and operation of the DC-DC converter 5 to which the present embodiment is applied will be described with reference to fig. 5.

The DC-DC converter 5 shown in fig. 5 includes a power conversion circuit 30 and a control circuit 40. The control circuit 30 changes the on time of the operation amount in the control circuit 20 of fig. 2 to the phase shift amount, and detailed description thereof is omitted.

First, the configuration of the power conversion circuit 30 will be described. The power conversion circuit 30 includes a transformer T having a primary winding N1, a secondary winding N2, and a secondary winding N3(N2 turns to N3 turns), a switching circuit 31 having switching elements S1 to S4, a rectifier circuit 32 including diodes D1 and D2, a switching circuit 31, a rectifier circuit 32, a smoothing inductor Lo, an input smoothing capacitor 35, and an output smoothing capacitor 36.

The switch circuit 31 is constituted by a 1 st switching leg to which the switching elements S1 and S2 are connected in series and a 2 nd switching leg to which the switching elements S3 and S4 are connected in series.

The switching circuit 31 has a full-bridge configuration in which the connection between the two ends of the 1 st switching leg and the 2 nd switching leg is connected between the two ends of the input smoothing capacitor 35, and the connection between the series connection point of the switching elements S1 and S2 and the series connection point of the switching elements S3 and S4 is connected between the two ends of the primary winding of the transformer T1. In fig. 5, MOSFETs are used as the switching elements S1 to S4, but other semiconductor elements such as IGBTs may be used.

Next, the operation of the power conversion circuit 30 will be described. First, in the switching circuit 31, the operations of simultaneously turning on S1 and S4 and simultaneously turning on the groups of S2 and S3 in the switching elements S1 to S4 are alternately repeated, whereby a rectangular wave ac voltage is generated from the dc voltage input to the smoothing capacitor 35.

The drive pulses Ps1 to Ps4 of the switching elements S1 to S4 complementarily switch the switching elements S1 and S2 and the switching elements S3 and S4 on the basis of the on duty of 50%. In addition, the lengths of the simultaneous on periods of the switching elements S1 and S4 and the simultaneous on periods of S2 and S3 are changed by changing the phase difference between the driving pulse of the 1 st switching leg and the driving pulse of the 2 nd switching leg. Thereby, the voltage application time ratio of the positive side and the negative side of the rectangular alternating-current voltage is adjusted.

The rectangular wave ac voltage generated by the switching circuit 31 is input between both ends of the primary winding N1 of the transformer, and an ac current flows through the primary winding N1. The ac current flowing through the primary winding flows through the ac induced current in the secondary windings N2 and N3. The ac induced current is rectified by the rectifier circuit 32, and a dc voltage smoothed by the smoothing inductor Lo and the output smoothing capacitor 36 is output.

The above operation is shown in fig. 6 and 7. Fig. 6 shows waveforms of on/off states of the drive pulses Ps1 to Ps4 of the switching elements S1 to S4, an output voltage (equal to N1 voltage) of the switching circuit 31, a primary winding N1 current, and a smoothing inductor Lo current. Further, it is preferable to set the dead time between the S1 drive pulse Ps1 and the S2 drive pulse Ps2 and between the S3 drive pulse Ps3 and the S4 drive pulse Ps4, but this is omitted here. Fig. 7(a) to 7(d) show current paths of the power conversion circuit 30, and the current paths (a) to (d) in fig. 7 correspond to the periods (a) to (d) in fig. 6. The operations in the periods (a) to (d) are as follows.

In the period (a), S1 and S4 are turned on, and a positive voltage is applied to N1. The primary side flows a current in paths of S1, N1, S4, and the secondary side flows a current in paths of D1, N21, Lo, and increases.

In the period (b), S4 is turned off and S3 is turned on from the state (a). On the primary side, since the path of the current flowing through S4 is cut off, no voltage is applied to N1. The energy accumulated in the leakage inductance Lr of the transformer T (not shown) is returned and discharged through S1 and S3. The primary-side current flows through paths S1, N1, and S3, and the current decreases. The secondary side also releases the energy stored in Lo, and thus the current is continuously reduced.

In the period (c), S1 is turned off and S2 is turned on from the state (b). S2 and S3 are turned on, and N1 is applied with a negative voltage. The primary side flows a current in paths of S3, N1, S2, and the secondary side flows a current in paths of D2, N3, Lo, and increases.

In the period (d), S3 is turned off and S4 is turned on from the state (c). On the primary side, since the path of the current flowing through S2 is cut off, no voltage is applied to N1. The energy accumulated in the leakage inductance Lr of the transformer T is returned and released through S2 and S4. The primary-side current flows through paths S4, N1, and S2, and the current decreases. The secondary side also releases the energy stored in L, and thus the current is continuously reduced.

After the period (d), S3 is turned off, S1 is turned on, and the operation returns to the period (a) again, and the operations from the period (a) to the period (d) are repeated.

As described above, the power conversion circuit 30 of the DC-DC converter 5 in the present embodiment controls the output voltage Vo by changing the phase shift amount Tps corresponding to the phase difference of the driving pulses of the 1 st switching leg and the 2 nd switching leg. Here, if the ratio of the phase shift amount to the half period of the switching period Tsw is defined as the phase shift duty ratio Dps, Dps ═ Tps ÷ (Tsw ÷ 2), the relationship of the output voltage Vo to the phase shift amount Tps is expressed as Vo ═ N × Vi × Dps ═ N × Vi × Tps ÷ (Tsw ÷ 2). Therefore, as in embodiment 1, by changing the switching period Tsw in addition to the phase shift amount Tps which is the operation amount, the resolution of the phase shift duty Dps can be improved, and the resolution of the output voltage control can be improved.

In the case where the transformer T is provided as in the present embodiment, it is preferable to equalize the periods of the positive side and the negative side of the rectangular wave ac voltage input to the transformer T in order to suppress the magnetic bias phenomenon of the transformer T. However, if the switching period is set to an odd number, a difference may occur between the periods of the positive side and the negative side of the rectangular wave ac voltage. For example, if the switching period Tsw is 601 clocks, when complementary driving is performed with the on duty ratio of the driving pulse set to 50% as in the switching elements S1 and S2, one on time becomes 300 clocks and the other on time becomes 301 clocks. As a result, a difference occurs between the periods of the positive side and the negative side of the rectangular wave ac voltage input to the transformer T. Therefore, in the case where the transformer T is provided, the basic switching cycle is set to the even-numbered clock, and in the switching cycle calculation means in fig. 5, the amount of change in the switching cycle Tsw is preferably limited to the even-numbered clock so that the switching cycle becomes the even-numbered clock.

(conclusion)

As described above, according to embodiments 1 and 2 of the present invention, in the digital control type switching power supply device, the on duty Don can be finely controlled by changing the switching period Tsw in accordance with the difference between the calculated value of the operation amount and the set value. As a result, a switching power supply device with high voltage control resolution and small voltage fluctuation can be provided. Further, in the present invention, since a new circuit is not required to be added in order to improve the resolution of control, it is possible to contribute to a small-sized and low-cost switching power supply device. In embodiments 1 and 2, the switching power supply device mounted on the vehicle is taken as an example, but the present invention is not limited to the vehicle, and any switching power supply device may be applied. The operation amount is not limited to the on time and the phase shift amount shown in the present embodiment, and the present invention can be applied to any operation amount that can improve the resolution of the control target in combination with the change in the switching period Tsw. Further, the power conversion circuit is not limited to the step-down chopper and the step-down converter of the phase shift system shown in the present embodiment, and can be applied to any circuit system such as a step-up chopper, a bidirectional step-up/down chopper, a step-up converter of the phase shift system, another flyback converter, and a forward converter. The control unit is not limited to a DSP and may be replaced by a microcomputer or other digital control unit.

Description of the symbols

1 … vehicle, 2 … host battery, 3 … inverter, 4 … motor, 5 … DC-DC converter, 6 … auxiliary battery, 7 … auxiliary, 10 … power conversion circuit, 11 … switching element, 12 … diode, 13 … inductor, 14 … input smoothing capacitor, 15 … output smoothing capacitor, 20 … control circuit, 21 … clock signal generation unit, 22 … DSP (digital signal processor), 23 … a/D conversion unit, 24 … on time operation unit, 25 … discard processing unit less than 1 clock, 26 … switching period operation unit, 27 … on time setting register, 28 … switching period setting register, 29 … driving pulse generation unit, 30 … power conversion circuit, 31 … switching circuit, 35 … input smoothing capacitor, 36 … output smoothing capacitor, 40 … control circuit, D1 … diode, A D2 … diode, an S1 … switching element, an S2 … switching element, an S3 … switching element, an S4 … switching element, a T … transformer, a Lo … smoothing inductor, an N1 … primary winding, an N2 … secondary winding, and an N3 … secondary winding.