阅读说明：本技术 对开关功率变换器充电的自举电容 (Bootstrap capacitor for charging switching power converter ) 是由 刘文铎 郑聪 于 2019-02-01 设计创作，主要内容包括：本公开涉及对开关功率变换器充电的自举电容,提供了用于对自举电容器充电的充电路径,该自举电容器存储用于驱动反激式变换器中的有源钳位开关晶体管的驱动器电源电压。该充电路径将来自有源钳位电容器的电荷耦合以对自举电容器充电。(The present disclosure relates to a bootstrap capacitor for charging a switching power converter, providing a charging path for charging a bootstrap capacitor that stores a driver supply voltage for driving an active clamp switching transistor in a flyback converter. The charging path couples charge from the active clamp capacitor to charge the bootstrap capacitor.)

1. A flyback converter comprising:

an active clamp capacitor;

a bootstrap capacitor to store a driver supply voltage to drive an active clamp switching transistor coupled to the active clamp capacitor;

a power supply capacitor for storing a power supply voltage of a controller for controlling switching of the power switching transistor;

a first charging path to couple charge from the supply capacitor to charge the bootstrap capacitor; and

a second charging path to couple charge from the active clamping capacitor to charge the bootstrap capacitor.

2. The flyback converter of claim 1, wherein a terminal of the active clamp switch transistor is coupled to a drain of the power switch transistor.

3. The flyback converter of claim 2, wherein the second charging path includes a resistor coupled between the positive plate of the active clamp capacitor and the positive plate of the bootstrap capacitor.

4. The flyback converter of claim 3, wherein the resistor is coupled in series with a diode between the positive plate of the active clamp capacitor and the positive plate of the bootstrap capacitor.

5. The flyback converter of claim 2, wherein the second charging path comprises:

a charge transistor coupled between a positive plate of the active clamp capacitor and a positive plate of the bootstrap capacitor; and

a control circuit configured to turn on the charging transistor in response to a supply voltage of the controller being less than a reference voltage.

6. The flyback converter of claim 5, wherein the reference voltage is a breakdown voltage of a zener diode.

7. The flyback converter of claim 5, wherein the control circuit comprises a comparator.

8. The flyback converter of claim 5, wherein the charging transistor and the comparator are integrated into an integrated circuit.

9. The flyback converter of claim 2, wherein the active clamp switch transistor is an NMOS transistor.

10. The flyback converter of claim 2, wherein the active clamp switching transistor is a PMOS transistor.

11. The flyback converter of claim 1, wherein a terminal of the active clamp switch transistor is coupled to a source of the power switch transistor.

12. The flyback converter of claim 11, wherein a terminal of the active clamp switching transistor is coupled to a source of the power switching transistor through the active clamp capacitor.

13. The flyback converter of claim 12, wherein the second charging path includes a resistor coupled between the positive plate of the active clamp capacitor and the positive plate of the bootstrap capacitor.

14. The flyback converter of claim 11, wherein the terminal of the active-clamp switching transistor is a source connected to ground, and wherein a drain of the active-clamp switching transistor is coupled to a primary winding of the flyback converter through the active-clamp capacitor.

15. The flyback converter of claim 11, wherein the active clamp switch transistor is an NMOS transistor.

16. The flyback converter of claim 11, wherein the active clamp switching transistor is a PMOS transistor.

17. A method of operating a flyback converter, comprising:

charging a power supply capacitor to provide a power supply voltage to a controller of the flyback converter;

conducting charge from the supply capacitor through a first charging path to charge a bootstrap capacitor to provide a driver supply voltage for driving an active clamp switching transistor; and

charge is conducted from the active clamp capacitor through a second charging path to charge the bootstrap capacitor.

18. The method of claim 17, wherein conducting charge from the active clamp capacitor through the second charging path comprises conducting charge from the active clamp capacitor through a resistor to charge the bootstrap capacitor.

19. The method of claim 17, wherein conducting charge from the active clamp capacitor comprises conducting charge from the active clamp capacitor through a charging transistor to charge the bootstrap capacitor.

20. The method of claim 19, further comprising turning on the charging transistor in response to the supply voltage being less than a threshold voltage, and turning off the charging transistor in response to the supply voltage being greater than the threshold voltage.

Technical Field

The present application relates to switching power converters, and more particularly to charging of bootstrap capacitors in switching power converters.

Background

High efficiency switching power converters, such as flyback converters, have become virtual universal adapters for battery chargers for mobile devices. In a flyback converter, a primary side controller controls the cycling of a power switching transistor connected between the primary winding of a transformer and ground. When the power switch is cycled on, the rectified ac supply voltage drives the primary winding current. The rectified ac supply voltage may be several hundred volts, so that it may stress the power switching transistor. In order to minimize the switching stress of the power switching transistors, quasi-resonant (valley mode switching) and zero voltage switching techniques are known. For example, it is known to employ valley switching techniques for resonant oscillation of the drain voltage when the cycling of the power switching transistor is off. The peak voltage of the resonant oscillation may be relatively robust (up to 200V or higher), while the minimum voltage (the trough in the resonant oscillation) is much lower. Thus, valley mode switching involves detecting or predicting a valley in the resonant oscillation so that the power switching transistor can be turned on at that valley.

Although valley mode switching thus reduces the voltage stress on the power switching transistors, note that the valley voltage is not zero, but can be in the range of 20V or even higher (e.g., up to 250V). Then, when the power switch transistor is turned on, the relatively high drain voltage is discharged to ground, which reduces efficiency. An alternative to higher power efficiency for valley mode switching is Zero Voltage Switching (ZVS), which may also be denoted as active clamp operation. In active clamp operation, leakage energy in the transformer is stored and recovered in an active clamp capacitor, which is coupled to a terminal of the power switch transistor through the active clamp switch transistor. The active clamp switching transistor cycles at the peak of the resonant oscillation, with the leakage voltage of the power switching transistor discharging to ground as the leakage energy is recovered. Thus, the active clamp architecture does not stress switch the on-time of the power switch transistor, since the on-time occurs when the drain voltage is discharged.

While active clamp operation is therefore advantageous, the switching of an active clamp switch requires a suitable driver. The driver of the active clamp switching transistor is typically powered using charge from a bootstrap capacitor, which is in turn charged by a supply voltage VCC from a supply capacitor. An exemplary flyback converter 100 is shown in fig. 1. The controller U1 controls the switching of the power switch transistor S1 to regulate the output voltage stored on the output capacitor Cout. The drain of the power switch transistor S1 is connected to the primary winding of the transformer T so that the input voltage Vin forces a magnetizing current to flow in the primary winding. During the on-time of the power switch transistor S1, the output diode D3 prevents current from flowing into the secondary winding of the transformer. In an alternative embodiment, the rectification may also be performed by synchronous rectifier switching transistors. When the controller U1 cycles off the power switch transistor S1, the output diode D3 becomes forward biased so that a secondary current flows to charge the output capacitor Cout with the output voltage. When the power switch transistor S1 is off, the drain of the power switch transistor S1 is charged high. Similarly, when the power switch transistor S1 is off, the auxiliary winding (not shown) is also charged high. To obtain this energy to support the supply voltage VCC, the auxiliary winding (Aux) is coupled through a current limiting resistor R1 and a supply diode D2 to charge the supply capacitor VCC with the supply voltage VCC.

The supply voltage VCC also powers a driver Dr for the active clamp switch transistor S2, which is coupled between the drain of the power switch transistor S1 and an active clamp capacitor Ca, which is in turn coupled to the input supply rail supplying the input voltage Vin. In particular, diode D1 couples the supply voltage VCC to the bootstrap capacitor CB. The resulting voltage from the bootstrap capacitor CB powers the driver Dr so that active clamping operation can be achieved. However, as the switching frequency of the power switching transistor S1 decreases during low load operation, the charging of the supply capacitor VCC decreases, causing the supply voltage to drop accordingly. Then, the voltage across the bootstrap capacitor used to power the driver Dr may be too low, so that the active clamping operation is lost, resulting in additional switching losses, voltage spikes and electromagnetic interference (EMI) problems. In particular, leakage inductance energy that would normally be released may accumulate on the active clamp capacitor Ca. The voltage rise generated on the active clamp capacitor Ca may damage the active clamp capacitor and cause a safety problem. Additional circuitry is therefore required to avoid breakdown of the active clamp capacitor Ca during low frequency operation, which increases the number of components and increases the cost.

Accordingly, there is a need in the art for improved active clamp operation of flyback converters.

Disclosure of Invention

To meet the need in the art for improved active clamp operation of flyback converters, a low frequency charging path for a bootstrap capacitor that supplies power to an active clamp switching transistor is provided. The low frequency charging path is complementary to a conventional high frequency charging path that conducts charge from the supply capacitor to charge the bootstrap capacitor. This conventional charging path is denoted as a "high frequency" charging path because it provides most of the bootstrap capacitor charging during relatively high loads, while the power switch transistor cycles at a correspondingly relatively high rate. However, as previously mentioned, the efficiency of the conventional charging path becomes an issue during light loads when the power switching transistors are cycled at a correspondingly relatively low rate. Conversely, the low frequency charging path conducts charge from the active clamp capacitor to charge the bootstrap capacitor. The low frequency charging path conducts less charge than the high frequency charging path. But during light load operation this relatively small amount of charge is sufficient to keep the bootstrap capacitor charged. As a result, active clamp operation is maintained in both high and low frequency modes of operation, which improves efficiency and reliability.

These advantageous features may be better understood by considering the following detailed description.

Drawings

Fig. 1 shows a flyback converter with a conventional bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for a high-side active clamp switch.

Fig. 2 shows a flyback converter with an improved bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for an NMOS high-side active clamp switching transistor that is coupled to an input voltage rail through an active clamp capacitor.

Fig. 3 shows the flyback converter of fig. 2, wherein the improved bootstrap capacitor charging path includes a resistor.

Fig. 4 shows the flyback converter of fig. 2, wherein the improved bootstrap capacitor charging path is actively regulated.

Fig. 5 shows a flyback converter with an improved bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for an NMOS high-side active clamp switching transistor that is directly connected to an input voltage rail.

Fig. 6A shows a flyback converter with an improved bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for a PMOS high-side active clamp switching transistor that is coupled to an input voltage rail through an active clamp capacitor.

Fig. 6B shows a flyback converter with an improved bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for a PMOS high-side active clamp switching transistor that is directly connected to the input voltage rail.

Fig. 7A shows a flyback converter with an improved bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for an NMOS low side active clamp switching transistor that is coupled to ground through an active clamp capacitor.

Fig. 7B shows a flyback converter with an improved bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for an NMOS low side active clamp switching transistor that is coupled to ground through an active clamp capacitor.

Fig. 8A shows a flyback converter with an improved bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for a PMOS low-side active-clamp switching transistor that is coupled to ground through an active clamp capacitor.

Fig. 8B shows a flyback converter with an improved bootstrap capacitor charging path for charging a bootstrap capacitor that powers a driver for a PMOS low-side active clamp switching transistor that is directly coupled to ground.

Embodiments of the present invention and their advantages are best understood by referring to the detailed description that follows. It should be appreciated that like reference numerals are used to identify like elements illustrated in one or more of the figures.

Detailed Description

To support active clamping operation of a flyback converter with active clamping (zero voltage switching) operation during high and low load conditions, two charging paths are provided for a bootstrap capacitor that stores a supply voltage for powering a driver of the active clamping switching transistor. The first of the charging paths is conventional, such as discussed with respect to fig. 1. Specifically, the controller U1 of the flyback converter 200 shown in fig. 2 controls the switching of the power switching transistor S1 to regulate the output voltage stored on an output capacitor (not shown) on the secondary side of the transformer T. The drain of the power switch transistor S1 is connected to the primary winding of the transformer T so that the input voltage Vin forces a magnetizing current to flow in the primary winding. During the on-time of the power switch transistor S1, the output diode D3 prevents current from flowing into the secondary winding of the transformer. In an alternative embodiment, the rectification may also be performed by synchronous rectifier switching transistors. When an output capacitor (not shown) on the secondary side of the transformer T is charged, the active clamp capacitor Ca is also charged by the conduction of the active clamp switching transistor S2, so that the active clamp capacitor Ca is charged to a voltage greater than the input voltage Vin. When the controller U1 cycles the power switch transistor S1 off, the output diode D3 becomes forward biased so that a secondary current flows to charge the output capacitor with the output voltage. As discussed with respect to fig. 1, when power switch transistor S1 is off, the auxiliary winding (Aux) is charged high. To obtain this energy to support the supply voltage VCC, the auxiliary winding is coupled through a current limiting resistor R1 and a supply diode D2 to charge the supply capacitor VCC with the supply voltage VCC.

The supply voltage VCC supplies the driver Dr of the active clamp switching transistor S2. The active clamp transistor S2 is coupled between the drain of the power switch transistor S1 and the positive plate of the active clamp capacitor Ca. The active clamp capacitor Ca is in turn coupled to an input power rail supplying the input voltage Vin. As part of the first charging path, the supply voltage VCC forward biases diode D1 to apply the driver supply voltage across bootstrap capacitor CB. The resulting driver supply voltage from bootstrap capacitor CB supplies driver Dr so that active clamping operation can be achieved by turning on and off active clamping switch transistor S2, as is known in the art of active clamping operation. But as previously mentioned, charging the driver supply voltage through this conventional high switching frequency VCC path is dependent on the switching frequency. During low load conditions (where the switching frequency of the power switch transistor S1 is reduced to prevent the output voltage from exceeding regulation), the current consumption of the driver Dr may cause the driver supply voltage to drop too low to support active clamping operation. Without active clamping operation, the efficiency during these low load conditions is reduced due to switching losses. Furthermore, voltage spikes across the power switch transistor may cause damage and EMI problems. Similarly, the voltage across the active clamp capacitor Ca may rise to dangerous levels due to the loss of active clamp operation. To maintain active clamp operation, flyback converter 200 includes a low switching frequency charge path circuit 205 that passively or actively controls the charge flow from active clamp capacitor Ca to bootstrap capacitor CB to maintain the driver supply voltage even as the switching frequency of power switch transistor S1 decreases during low load conditions of flyback converter 200. The positive plate for bootstrap capacitor CB is coupled to the charge path circuit 205 and to the driver supply voltage input for driver Dr. The negative plate for the bootstrap capacitor CB is connected to the source of the active clamp switch transistor S2 and to the drain of the power switch transistor S1.

The charge path circuit 205 may include an active circuit or include a passive resistor. An example flyback converter 300 is shown in fig. 3, where the charge path circuit 205 includes a resistor R. To assist in the desired charging of bootstrap capacitor CB, a resistor R may be connected in series with diode D4. The series combination of resistor R and diode D4 is coupled between active clamp capacitor Ca and bootstrap capacitor CB. In particular, the series combination of resistor R and diode D4 is coupled between the positively charged plate of active clamp capacitor Ca and the positively charged plate of bootstrap capacitor CB. It should be understood that diode D4 may be omitted in alternative embodiments, such that resistor R would be coupled directly between the positively charged plates for bootstrap capacitor CB and active clamp capacitor Ca. For clarity, the secondary side of the transformer T is not shown in fig. 3.

A zener diode Z1 may be coupled across bootstrap capacitor CB to prevent overcharging of the driver supply voltage. With respect to this charging of the driver supply voltage, it is noted that resistor R provides a bootstrap capacitor charging path regardless of whether the power switch transistor S1 is on or off. If the power switch transistor S1 is turned on, the charging path extends from the active clamp capacitor Ca through the charging path circuit 205, the bootstrap capacitor CB and the power switch transistor S1 to ground. When the power switch transistor S1 is turned on, the voltage across the charge path circuit 205 is Vin plus the voltage across the active clamp capacitor Ca. Thus, when power switch transistor S1 is turned on, the voltage across charge path circuit 205 is approximately Vin + nVout, where Vin is the input voltage, Vout is the output voltage, and n is the turns ratio of transformer T. If power switch transistor S1 is turned off, a charging path extends from active clamp capacitor Ca through charging path circuit 205, bootstrap capacitor CB, and the primary winding of transformer T1. When the power switch transistor S1 is off, the voltage across the charge path circuit 205 is approximately vout. The resistor R limits the charging current to a relatively small value, such as a few tens of microamps. Driver Dr may consume tens of milliamps during high frequency switching such that a conventional charge path from the positive plate of supply capacitor VCC to the positive plate of bootstrap capacitor CB through diode D1 will dominate the charging of the driver supply voltage during high frequency operation.

Although resistor R discharges energy through resistive losses while driver Dr switches active clamp switch transistor S2 and thus consumes the driver current supplied through resistor R to maintain the driver supply voltage, it is noted that this driver current is relatively small if driver Dr is implemented using silicon technology. In particular, it is noted that the current consumption of the driver Dr depends on whether it is in a quiescent state (active clamp switch transistor S2 is not switching) or whether it is active such that the driver Dr drives the active clamp switch transistor S2 on/off in each switching cycle. For CMOS embodiments, the quiescent current consumption is typically small. In this case, the losses introduced by the resistor R are relatively small. However, if the driver Dr of the active clamp switching transistor S2 is implemented using, for example, GaN technology, the quiescent current consumption is higher, so that the driver Dr consumes significantly more current, making the losses from the resistor R more significant. An active circuit embodiment for the charge path circuit 205 (where the charge path circuit 205 includes an NMOS transistor S3 for the flyback converter 400 such as shown in fig. 4) avoids such resistive losses. The secondary side of transformer T1 is not shown in fig. 4 for clarity. A PMOS transistor S4 having a drain connected to the drain of the power switch transistor S1 controls whether the transistor S3 is on or off. The source of transistor S3 is coupled to the positive plate of bootstrap capacitor CB through resistor R6. The drain of transistor S3 is coupled to the positive plate of active clamp capacitor Ca through resistor R7. Therefore, when the transistor S3 is cycled on, the charging path is activated from the active clamp capacitor Ca through the resistor R7, the transistor S3, and the resistor R7 to charge the bootstrap capacitor CB. Specifically, in response to transistor S4 cycling on, transistor S3 cycles on. The source of transistor S4 is coupled to the positive plate of active clamp capacitor Ca through a voltage divider formed by a pair of resistors R3 and R4 in series. The node between resistors R4 and R4 drives the gate of transistor S3. If the transistor S4 is turned on, the divided voltage across the resistors R3 and R4 causes the gate-source voltage of the transistor S3 to drop sufficiently to turn on the transistor S3.

A zener diode Z2 arranged in parallel with the resistor R2 is used to clamp the gate voltage of the transistor S3 to protect it from excessive voltages. To control whether transistor S4 is turned on or off, comparator C1 compares the reference voltage Vref with a divided version of the driver supply voltage stored across bootstrap capacitor CB. A pair of resistors R4 and R5 are arranged in series between the positive plate of the bootstrap capacitor CB and the drain of the power switching transistor S1 to supply the divided voltage to the comparator C1. A resistor R8 in series with the zener diode Z3 is coupled between the positive plate of the bootstrap capacitor CB and the drain of the power switch transistor S1 to form a reference voltage that is the clamping voltage across the zener diode Z3. If the driver supply voltage drops, the divided voltage from resistors R4 and R5 will also drop so that it will drop below the reference voltage provided by zener diode Z3, causing the output of comparator C1 to go high to turn on transistor S4. Conversely, if the driver supply voltage is high enough, the output of comparator C1 will be low, causing transistor S4 to turn off, since in this case the divided voltage from resistors R4 and R5 is greater than the reference voltage. Thus, comparator C1 is used to regulate the driver supply voltage by indirectly controlling whether transistor S3 is on or off. It should be appreciated that when the voltage across bootstrap capacitor CB reaches an acceptable level, the charge path circuit 205 in flyback converter 400 may be modified to turn off transistor S3 during high frequency switching operations. In an alternative embodiment, a controller (not shown) controlling the switching of the active clamp transistor S2 may be configured to switch the transistor S3 in response to whether the bootstrap capacitor CB voltage is above or below a threshold level.

The positions of the active clamp switching transistor S2 and the active clamp capacitor Ca may be switched as shown in fig. 5 for the flyback converter 500. The drain of the active clamp switching transistor is connected to the input voltage rail. An active clamp capacitor Ca is connected between the source of the active clamp switch transistor S2 and the drain of the power switch transistor S1. Driver Dr, transformer T, bootstrap capacitor CB, controller U1, resistor R1, and diodes D1 and D2 are arranged as discussed with respect to flyback converter 200. However, the charge path circuit 205 is now connected between the positive plate of the bootstrap capacitor CB and the drain of the power switch transistor S1.

Flyback converters 500, 400, 300, and 200 are "high-side" active-clamp embodiments, with active-clamp switching transistor S2 coupled in series with power switching transistor S1. In flyback converters 200, 300, and 400, the source of active clamp switch transistor S2 is connected to the drain of power switch transistor S1 to complete the series connection. In contrast, the source of the active clamp transistor S2 is coupled to the drain of the power switch transistor S1 through an active clamp capacitor Ca to complete the series connection in the flyback converter 500. In all of these embodiments, the active clamp switch transistors are NMOS transistors. A similar "high side" active clamp embodiment may be constructed in which the active clamp switch transistors are PMOS transistors. An example flyback converter 600 is shown in fig. 6A, where the negative plate for the active clamp capacitor Ca is connected to the input voltage rail. The PMOS active clamp switch transistor S5 has its source connected to the positive plate of the active clamp capacitor Ca, and its drain connected to the drain of the power switch transistor S1. Transformer T1, driver Dr, and bootstrap capacitor CB are arranged as discussed with respect to flyback converter 200. The charge path circuit 205 is connected between the input voltage rail and the negative plate of the bootstrap capacitor CB. In another PMOS high-side embodiment, the positions of the active clamp switching transistor S5 and the active clamp capacitor Ca may be reversed, as shown in the flyback converter 650 of fig. 6B. The positive plate of bootstrap capacitor CB is connected to the input voltage rail such that the input voltage on the input voltage rail serves as the driver supply voltage for driver Dr. The charge path circuit 205 is connected between the negative plate of the bootstrap capacitor CB and the negative plate of the active clamp capacitor Ca. For clarity, the conventional charging path formed by the supply capacitor VCC, the diodes D1 and D2, and the resistor R1 is not shown in fig. 6A and 6B. Similarly, the controller U1 is not shown in fig. 6A and 6B.

The previous discussion has been directed to a "high side" active clamp embodiment, however, an alternative charging path for charging the bootstrap capacitor may also be implemented in a "low side" active clamp architecture, where the active clamp capacitor is coupled to the source of the power switch transistor S1. Since the source is grounded, this coupling is a low voltage compared to the higher voltage coupling of the "high side" active clamp architecture. NMOS embodiments of active clamp switching transistors in low side embodiments will be discussed first, followed by a discussion of PMOS low side embodiments. An example d low side NMOS flyback converter 700 is shown in fig. 7A. The negative plate of the active clamp capacitor Ca is connected to the source of the ground/power switch transistor S1, while the positive plate of the active clamp capacitor Ca is connected to the source of the NMOS active clamp switch transistor S6, whose drain is connected to the drain of the power switch transistor S1. The positive plate of bootstrap capacitor CB is connected to the positive plate of active clamp capacitor Ca through charge path circuit 205. In low-side embodiments such as flyback converter 700, the voltage across charge path circuit 205 is nearly constant equal to Vin + n Vout.

For flyback converter 750, the positions of active clamp capacitor Ca and active clamp switching transistor S6 may be reversed, as shown in fig. 7B. The positive plate of the active clamp capacitor Ca in the flyback converter 750 is connected to the drain of the power switch transistor S1, while the negative plate of the active clamp capacitor Ca is connected to the drain of the active clamp switch transistor S6, the source of which is grounded. The charge path circuit 205 is connected between the positive plate of the active clamp capacitor Ca and the positive plate of the bootstrap capacitor CB.

A low side PMOS active-clamped flyback converter 800 is shown in fig. 8A. As with the other PMOS embodiments, the charge path circuit 205 is connected between the negative plate of the active clamp capacitor Ca and the negative plate of the bootstrap capacitor CB. The negative plate of the active clamp capacitor Ca is grounded, and the positive plate thereof is connected to the positive plate of the active clamp capacitor CB and the source of the PMOS active clamp switching transistor S7. The drain of the PMOS active clamp switch transistor S7 is connected to the drain of the power switch transistor S1 and also to a terminal of the transformer primary winding. The positions of the active clamp capacitor Ca and the active clamp switching transistor S7 are reversed in the flyback converter 850 of fig. 8B compared to the flyback converter 800. Thus, the drain of the active clamp switching transistor S7 is connected to ground, while its source is connected to the negative plate of the active clamp capacitor Ca, whose positive plate is connected to the drain of the power switching transistor S2. Since it is a PMOS embodiment, the charge path circuit 205 is connected between the negative plate of the bootstrap capacitor CB and the active clamp capacitor Ca. The positive plate of the bootstrap capacitor CB is grounded. For clarity, the conventional charging path formed by VCC capacitor VCC, diodes D1 and D2, and resistor R1 is not shown in fig. 7A, 7B, 8A, and 8B. Similarly, the controller U1 is not shown in these figures. Similarly, as discussed with respect to converters 300 and 400, the charge path circuit 205 may be passive or active.

As some skilled in the art will now recognize, and in light of the specific applications for which it is useful, many modifications, substitutions, and variations can be made in the materials, devices, configurations, and methods of use of the devices of the present invention without departing from the scope of the devices of the present invention. In view of this, the scope of the present invention should not be limited to the particular embodiments shown and described herein, as they are intended merely as examples of the present invention, but rather should be fully commensurate with the following appended claims and their functional equivalents.