阅读说明：本技术 一种宽增益范围llc谐振变换器及其控制方法 (Wide-gain-range LLC resonant converter and control method thereof ) 是由 余逸群 李思远 李永昌 于 2019-09-06 设计创作，主要内容包括：本发明涉及一种宽增益范围LLC谐振变换器及其控制方法,变换器包括逆变电路、LLC谐振腔、变压器和整流网络；LLC谐振腔包括谐振电感Lr、Lm和谐振电容Cr；LLC谐振腔内还增设有双向开关；谐振电容Cr、谐振电感Lr依次串联在逆变电路的1号输出端与变压器原边线圈的1端之间,逆变电路的2号输出端与变压器原边线圈的2端连接,谐振电感Lm与变压器原边线圈并联,双向开关的1端通过连在谐振电容Cr、谐振电感Lr之间与变压器原边线圈的1端连接,双向开关的2端与变压器原边线圈的2端连接；逆变电路为全桥/半桥相结合的变拓扑电路。本发明通过全桥/半桥这种拓扑结构的变换,大大拓宽本发明的增益范围,而且本发明电路结构降低了变换器的设计难度。(The invention relates to a wide gain range LLC resonant converter and a control method thereof, wherein the converter comprises an inverter circuit, an LLC resonant cavity, a transformer and a rectifier network; the LLC resonant cavity comprises resonant inductors Lr and Lm and a resonant capacitor Cr; a bidirectional switch is additionally arranged in the LLC resonant cavity; the resonant capacitor Cr and the resonant inductor Lr are sequentially connected in series between the output end 1 of the inverter circuit and the end 1 of the primary coil of the transformer, the output end 2 of the inverter circuit is connected with the end 2 of the primary coil of the transformer, the resonant inductor Lm is connected in parallel with the primary coil of the transformer, the end 1 of the bidirectional switch is connected with the end 1 of the primary coil of the transformer by being connected between the resonant capacitor Cr and the resonant inductor Lr, and the end 2 of the bidirectional switch is connected with the end 2 of the primary coil of the transformer; the inverter circuit is a full-bridge/half-bridge combined variable topology circuit. The invention greatly widens the gain range of the invention through the conversion of the topological structure of full bridge/half bridge, and the circuit structure of the invention reduces the design difficulty of the converter.)

1. A wide gain range LLC resonant converter comprises an inverter circuit, an LLC resonant cavity, a transformer and a rectifier network which are sequentially connected from input to output;

the LLC resonant cavity comprises resonant inductors Lr and Lm and a resonant capacitor Cr;

the LLC resonant cavity is internally provided with a bidirectional switch; the resonant capacitor Cr and the resonant inductor Lr are sequentially connected in series between the output end 1 of the inverter circuit and the end 1 of the primary coil of the transformer, the output end 2 of the inverter circuit is connected with the end 2 of the primary coil of the transformer, the resonant inductor Lm is connected in parallel with the primary coil of the transformer, the end 1 of the bidirectional switch is connected with the end 1 of the primary coil of the transformer by being connected between the resonant capacitor Cr and the resonant inductor Lr, and the end 2 of the bidirectional switch is connected with the end 2 of the primary coil of the transformer;

the inverter circuit is a full-bridge/half-bridge combined variable topology circuit.

2. The LLC resonant converter according to claim 1, wherein said inverter circuit is comprised of four switching tubes S1, S2, S3, S4, wherein the switching tubes S1, S2 are respectively used for controlling whether the positive pole of the input power Vin is connected to the output terminals No. 1, No. 2 of said inverter circuit, and the switching tubes S3, S4 are respectively used for controlling whether the output terminals No. 1, No. 2 of said inverter circuit are connected to the negative pole of the input power Vin;

the input voltage range of the input power Vin is divided into two sections: the inverter circuit works in an FBLLC mode when the low voltage is input, and works in an HBLLC mode when the high voltage is input, namely when the high voltage is input, the switch tube S2 is continuously turned off, and the switch tube S4 is continuously turned on.

3. The LLC resonant converter according to claim 2, wherein said bidirectional switch is formed by two switching tubes S5, S6 connected in series in opposite directions, wherein a parasitic diode pointing to a connection of said bidirectional switch with said resonant inductor Lr and resonant capacitor Cr is a switching tube S5.

4. The LLC resonant converter of claim 3, wherein said rectification network employs a full wave rectification architecture.

5. The LLC resonant converter of claim 4, wherein the rectifying network employs a full bridge rectifying structure, the full bridge rectifying structure being constituted by rectifying diodes or switching tubes.

6. A method for controlling an LLC resonant converter as claimed in any one of claims 1 to 5, wherein said LLC resonant converter is controlled using fixed frequency PWM.

7. A method of controlling an LLC resonant converter as claimed in any one of claims 3 to 5, wherein said LLC resonant converter is controlled using fixed frequency PWM.

8. The control method of claim 7, wherein when a low voltage is input and the inverter circuit operates in the FBLLC mode, switching frequencies of the switching tubes S1-S6 are equal and fixed, the switching tube S1 and the switching tube S5 are complementarily turned on, the switching tube S2 and the switching tube S6 are complementarily turned on, the switching tube S1 and the switching tube S4 are simultaneously turned on and off, the switching tube S2 and the switching tube S3 are simultaneously turned on and off, a duty ratio of the switching tube S1 is equal to a duty ratio of the switching tube S2 and is not greater than 0.5 but 180 degrees out of phase, a duty ratio of the switching tube S5 is equal to a duty ratio of the switching tube S6 and is not less than 0.5 but 180 degrees out of phase, and an output voltage V is achieved by adjusting the duty ratio of the switching tube S1₀The larger the duty ratio of the switching tube S1 is, the larger the output voltage gain is;

when a high voltage is inputtedWhen the inverter circuit works in an HBLLC mode, the switching frequencies of the switching tubes S1-S6 are equal and fixed, the switching tube S1 and the switching tube S5 are in complementary conduction, the switching tube S3 and the switching tube S6 are in complementary conduction, the switching tube S4 is in continuous conduction, the switching tube S2 is in continuous turn-off, the duty ratio of the switching tube S1 is equal to that of the switching tube S3, not more than 0.5 and 180 degrees of phase difference between the switching tube S3 and the switching tube S6, not less than 0.5 and 180 degrees of phase difference between the switching tube S5 and the switching tube S6, and the output voltage V is realized by adjusting the duty ratio of the switching tube S1₀The larger the duty ratio of the switching tube S1 is, the larger the output voltage gain is.

Technical Field

The invention relates to the technical field of switching converters, in particular to an LLC resonant converter and a control method thereof.

Background

The switching converter is applied more and more widely due to rapid development in the field of power electronics. More requirements are put on switching converters: high power density, high reliability and small volume. An LLC resonant converter, as a resonant converter, has many advantages, such as low noise, low stress, low switching losses, etc. The LLC resonant converter generally adopts two working modes of variable frequency control and fixed frequency control, however, when the variation range of input voltage and load is very wide, the variation range of the working frequency of the LLC resonant converter which independently adopts variable frequency control is wide, so that the design of magnetic elements in the LLC resonant converter is difficult, and when the voltage gain is wide, the efficiency of the traditional variable frequency control LLC resonant converter is obviously reduced; the LLC controlled by the fixed frequency and the phase shift is independently adopted, the working frequency is fixed, so that the magnetic element is convenient to design, but in order to ensure that the input voltage and the load are unchanged in a wide range, the circuit is required to work under a larger phase shift angle, and because the phase shift circuit has the characteristic that a hysteresis bridge arm is difficult to realize soft switching, the circuit possibly loses the characteristic of the soft switching, so that under the control mode, the requirement that the hysteresis bridge arm can still realize the soft switching under the maximum phase shift angle needs to be considered, the larger the phase shift angle is, the wider the gain range is, and due to the consideration, the phase shift angle is limited, so that the LLC resonant converter controlled by the traditional fixed frequency and the phase shift can achieve higher efficiency but the limited gain range. In a word, when the LLC resonant converter is applied to an ultra-wide input occasion, the circuit cannot give consideration to the characteristics of high efficiency and high gain.

In "variable topology LLC circuit applied to ultra wide input range" published in university of zhejiang university proceedings (proceedings of industry) in 2013 by political, snow, ewing et al, a half-bridge LLC (HBLLC) structure is found in full-bridge LLC (FBLLC) topology, FBLLC topology is adopted when input voltage is low, HBLLC topology is adopted when input voltage is high, the equivalent circuit is shown in fig. 1, fig. 2 is a gain comparison under both topologies, it can be seen from the accompanying drawings that by switching between full-bridge and half-bridge structures, circuit gain can be doubled, and circuit efficiency is also improved favorably. However, it still belongs to frequency conversion control, and the design of the magnetic element is still more complicated.

Disclosure of Invention

The invention aims to solve the technical problem of providing an LLC resonant converter with a wide gain range and a control method thereof.

The technical scheme adopted by the converter is as follows: a wide gain range LLC resonant converter comprises an inverter circuit, an LLC resonant cavity, a transformer and a rectifier network which are sequentially connected from input to output;

the LLC resonant cavity comprises resonant inductors Lr and Lm and a resonant capacitor Cr;

the LLC resonant cavity is internally provided with a bidirectional switch; the resonant capacitor Cr and the resonant inductor Lr are sequentially connected in series between the output end 1 of the inverter circuit and the end 1 of the primary coil of the transformer, the output end 2 of the inverter circuit is connected with the end 2 of the primary coil of the transformer, the resonant inductor Lm is connected in parallel with the primary coil of the transformer, the end 1 of the bidirectional switch is connected with the end 1 of the primary coil of the transformer by being connected between the resonant capacitor Cr and the resonant inductor Lr, and the end 2 of the bidirectional switch is connected with the end 2 of the primary coil of the transformer;

the inverter circuit is a full-bridge/half-bridge combined variable topology circuit.

The gain range of the half bridge is half of the full bridge, if the gain range of the full bridge is 2-4, the gain range of the whole circuit can be 1-4, and the circuit structure can greatly widen the gain range of the converter. In addition, the bidirectional switch is additionally arranged in the resonant cavity, the output voltage stabilization can be realized by controlling the conduction time of the bidirectional switch, so that the fixed frequency control is realized, the requirements on magnetic elements such as a transformer are reduced, and the switching device and the inverter circuit are positioned on the primary side of the transformer without considering the problem of isolation driving.

It should be noted that the terminals 1 and 2 stated above are only given reference numerals for convenience of description, and correspond to the upper input/output terminal and the lower input/output terminal of the inverter circuit, the bidirectional switch and the primary winding of the transformer, respectively, if corresponding to fig. 3.

The inverter circuit is composed of four switching tubes S1, S2, S3 and S4, wherein the switching tubes S1 and S2 are respectively used for controlling whether the anode of an input power Vin is communicated with the No. 1 and No. 2 output ends of the inverter circuit, and the switching tubes S3 and S4 are respectively used for controlling whether the No. 1 and No. 2 output ends of the inverter circuit are communicated with the cathode of the input power Vin;

the input voltage range of the input power Vin is divided into two sections: the inverter circuit works in an FBLLC mode when the low voltage is input, and works in an HBLLC mode when the high voltage is input, namely when the high voltage is input, the switch tube S2 is continuously turned off, and the switch tube S4 is continuously turned on.

The circuit is switched between HBLLC (full bridge LLC) and FBLLC (half bridge LLC) along with the change of the input voltage, and the variable topology circuit is completely realized by only depending on software control without adding extra devices.

The bidirectional switch is composed of two switching tubes S5 and S6 which are connected in series in an opposite direction, wherein a parasitic diode points to the connection end of the bidirectional switch, the resonant inductor Lr and the resonant capacitor Cr and is a switching tube S5.

The rectification network adopts a full-wave rectification structure, such as a full-bridge rectification structure, and the full-bridge rectification structure is composed of rectifier diodes or switching tubes.

The control method adopts the following technical scheme: the control method of the LLC resonant converter is characterized in that the LLC resonant converter is controlled by using fixed-frequency PWM (Pulse Width Modulation for short).

Specifically, the method comprises the following steps: when low voltage is input and the inverter circuit works in an FBLLC mode, the switching frequencies of the switching tubes S1-S6 are equal and fixed, the switching tube S1 and the switching tube S5 are complementarily conducted, the switching tube S2 and the switching tube S6 are complementarily conducted, the switching tube S1 and the switching tube S4 are simultaneously conducted and simultaneously turned off, and the switching tube S2 and the switching tube S3 are simultaneously conducted and simultaneously turned onThe time is turned off, the duty ratio of the switch tube S1 is equal to the duty ratio of the switch tube S2, the duty ratios are not more than 0.5 and 180 degrees of phase difference between the two, the duty ratio of the switch tube S5 is equal to the duty ratio of the switch tube S6, the duty ratios are not less than 0.5 and 180 degrees of phase difference between the two, and the output voltage V is realized by adjusting the duty ratio of the switch tube S1₀The duty ratio of the switching tube S1 is changed, and the conduction time of the bidirectional switch is synchronously changed, the larger the duty ratio of the switching tube S1 is, the larger the output voltage gain is;

when high voltage is input and the inverter circuit works in an HBLLC mode, the switching frequencies of the switching tubes S1-S6 are equal and fixed, the switching tube S1 and the switching tube S5 are complementarily conducted, the switching tube S3 and the switching tube S6 are complementarily conducted, the switching tube S4 is continuously conducted, the switching tube S2 is continuously turned off, the duty ratio of the switching tube S1 is equal to that of the switching tube S3 and not more than 0.5, the phase difference between the switching tube S3 and the switching tube S6 is 180 degrees, the duty ratio of the switching tube S5 is equal to that of the switching tube S6 and not less than 0.5, the phase difference between the switching tube S6 is 180 degrees, and the output voltage V1 is realized by adjusting the₀The larger the duty ratio of the switching tube S1 is, the larger the output voltage gain is.

Has the advantages that:

1) the invention adopts a full-bridge/half-bridge combined variable topology circuit as an inverter circuit, the gain range of the half-bridge is half of that of the full-bridge, and the gain range of the invention can be greatly widened and the working efficiency of the invention can be improved through the conversion of the topology structure;

2) the bidirectional switch is additionally arranged in the resonant cavity, the output voltage stabilization can be realized by controlling the conduction time of the bidirectional switch, so that the fixed frequency control is realized, the requirements on magnetic elements such as a transformer are reduced, and the switching device and the inverter circuit are positioned on the same side of the transformer, so that the circuit control and driving difficulty is reduced;

3) when the bidirectional switch is conducted, the energy of the resonant current is stored in a loop formed by the excitation inductor Lm, the resonant inductor Lr and the bidirectional switch in the circulation stage, and does not flow through the resonant capacitor Cr, and the parasitic resistance of the resonant capacitor Cr is beneficial to reducing the loss of the energy on the parasitic resistance of the resonant capacitor Cr, so that the working efficiency of the circuit is improved;

4) the invention adopts fixed frequency PWM control and can be respectively combined with half-bridge and half-bridge topologies, thereby realizing wider voltage gain range and higher efficiency by matching with a circuit and enabling the converter to be suitable for occasions with wider gain range requirements. The control method has the advantages of small frequency conversion range, low requirements on magnetic elements such as transformers and inductors, no leading bridge arm and lagging bridge arm, wide voltage gain range and high efficiency compared with frequency conversion control even if the control method does not combine a variable topology.

Drawings

Fig. 1 is an equivalent circuit diagram of a resonant converter when an HBLLC topology is employed;

FIG. 2 is a gain comparison of HBLLC and FBLLC;

FIG. 3 is a schematic circuit diagram of a wide gain range LLC resonant converter in accordance with a preferred embodiment of the invention;

FIG. 4 is a diagram showing the main operating waveforms of the wide gain range LLC resonant converter of the preferred embodiment of the invention when operating in FBLLC mode with fixed frequency control;

fig. 5 to 10 are equivalent circuit diagrams of switching modes of the wide gain range LLC resonant converter of the preferred embodiment of the present invention when operating in FBLLC mode and adopting fixed frequency control;

FIG. 11 is a diagram showing the main operating waveforms of the wide gain range LLC resonant converter of the preferred embodiment of the invention when operating in the HBLLC mode and using fixed frequency control;

fig. 12 to 17 are equivalent circuit diagrams of the switching modes of the wide gain range LLC resonant converter according to the preferred embodiment of the present invention when the wide gain range LLC resonant converter operates in the HBLLC mode and adopts fixed frequency control.

Detailed Description

As shown in FIG. 3, the wide gain range LLC resonant converter of the present embodiment includes an inverter connected in series from input to outputCircuit 10, LLC resonant cavity 20, transformer T and rectifier network 30. In the figure, Vin is the input power supply of the converter, and Ro is the output load R of the converter₀。

The inverter circuit 10 is a full-bridge/half-bridge combined variable topology circuit, and is composed of a switching tube S1, a switching tube S2, a switching tube S3, and a switching tube S4. The LLC resonant cavity 20 includes resonant inductor Lr, excitation inductor Lm, and resonant capacitor Cr, and is further provided with a bidirectional switch formed by a switch tube S5 and a switch tube S6. The rectifier network 30 is a full bridge rectifier circuit consisting of 4 diodes D1-D4 and connected with an output filter capacitor C in parallel₀And (4) forming.

The drain of the switch tube S1 is connected to the drain of the switch tube S2 and the positive terminal of the input power Vin, the source of the switch tube S1 is connected to the drain of the switch tube S3 and one end of the resonance capacitor Cr, the other end of the resonance capacitor Cr is connected to one end of the resonance inductor Lr and the drain of the switch tube S5, the other end of the resonance inductor Lr is connected to one end of the excitation inductor Lm and the 1 end of the primary winding Np of the transformer T, the 2 end of the transformer T is connected to the other end of the excitation inductor Lm, the source of the switch tube S2, the drain of the switch tube S4 and the drain of the switch tube S6, the source of the switch tube S4 is connected to the source of the switch tube S3 and the negative electrode of the input power Vin, and the source of the switch tube S483; the 1 end of the secondary winding Ns of the transformer T is connected to the anode of the secondary rectifying diode D1 and the cathode of the secondary rectifying diode D3, the cathode of the secondary rectifying diode D1 is connected to the cathode of the secondary rectifying diode D2, one end of the secondary output filter capacitor Co and one end of the output load Ro, the other end of the output load Ro is connected to the other end of the secondary output filter capacitor Co, the anode of the secondary rectifying diode D3 and the anode of the secondary rectifying diode D4, and the cathode of the secondary rectifying diode D4 is connected to the anode of the secondary rectifying diode D2 and the 2 end of the secondary winding Ns of the transformer T.

The ends 1 of the primary winding and the secondary winding of the transformer are homonymous ends, and the ends 2 of the primary winding and the secondary winding of the transformer are homonymous ends.

The wide gain range LLC resonant converter can adopt the following fixed frequency PWM control method:

when low voltage is input, the inverter circuit works in an FBLLC mode, and the switching frequency of the switching tubes S1-S6The output voltage V is equal and fixed, the switch tube S1 and the switch tube S5 are complementarily conducted, the switch tube S2 and the switch tube S6 are complementarily conducted, the switch tube S1 and the switch tube S4 are simultaneously conducted and simultaneously turned off, the switch tube S2 and the switch tube S3 are simultaneously conducted and simultaneously turned off, the duty ratio of the switch tube S1 is equal to the duty ratio of the switch tube S2, the duty ratio of the switch tube S5 is equal to the duty ratio of the switch tube S6, the phase difference of the switch tube S5 and the phase difference of the switch tube S6 are both equal to each other and 180 degrees₀The larger the duty ratio of the switching tube S1 is, the larger the output voltage gain is;

when high voltage is input, the inverter circuit works in an HBLLC mode, the switching frequencies of the switching tubes S1-S6 are equal and fixed, the switching tube S1 and the switching tube S5 are in complementary conduction, the switching tube S3 and the switching tube S6 are in complementary conduction, the switching tube S4 is in continuous conduction, the switching tube S2 is in continuous off, the duty ratio of the switching tube S1 is equal to that of the switching tube S3 and not more than 0.5, the phase difference between the switching tube S3 and the switching tube S6 is 180 degrees, the duty ratio of the switching tube S5 is equal to that of the switching tube S6 and not less than 0.5, the phase difference between the switching tube S6 and the switching tube S1 is 180 degrees, and the₀The larger the duty ratio of the switching tube S1 is, the larger the output voltage gain is.

In practical implementation, a reasonable dead time must be set between the switching signals of the switching tube S1 and the switching tube S5 to realize soft switching of the switching tube S1, the switching tube S4 and the switching tube S5; a reasonable dead time must be set between the switching signals of the switching tube S2 and the switching tube S6 to realize soft switching of the switching tube S2, the switching tube S3 and the switching tube S6. Coss 1-Coss 6 represent output capacitances to the sixth switching tubes S1-S6, respectively.

The following describes the working process of the wide gain range LLC resonant converter using fixed frequency PWM control in detail with reference to fig. 3.

In this embodiment, the parameters are selected as follows: the transformer has the advantages that Lr is 78nH, Lm is 300nH, Cr is 200nF, the input voltage range is 18-75 VDC, and the primary-secondary turn ratio of the transformer is 1.2: 1.

When the input voltage of the converter is 18V-36V, the converter works in a full-bridge mode, and fig. 4 shows the resonant conversionThe device adopts a main working waveform diagram when constant frequency PWM control is adopted, Vgs1/4 is a driving signal of a switching tube S1 and S4, Vgs2/3 is a driving signal of a switching tube S2 and S3, Vgs5 is a driving signal of a switching tube S5, Vgs6 is a driving signal of a switching tube S6, Vc, iLr, iLm, i are₀Respectively representing the voltage across Cr, the current through Lr, the current through Lm and the current through resistor R₀The current of (2). As can be seen from FIG. 4, the present invention outputs a current I₀The change is gentle and the stress of the device is small. The converter has six switching modes in a half cycle, which are respectively shown in fig. 5-10 (the working modes of the second half cycle and the first half cycle of the LLC resonant converter are symmetrical from the waveform diagram, and generally, the description of the LLC resonant converter only describes a half cycle).

Switched mode 1[ t ]₀，t₁]: as shown in fig. 5, at t₀Before the moment, the switch tube S6 is conducted, the switch tube S5 is turned off, and the body diode bears reverse voltage and is cut off in the reverse direction; at the time t0, the switch tube S1 and the switch tube S4 are switched on at zero voltage; the rectifier diode D1 and the rectifier diode D4 are conducted, and the current flowing through the diodes is in direct proportion to the difference value of the resonance current and the excitation current; the voltage at two ends of the excitation inductor Lm is output and clamped to nV_O(n is the transformer turn ratio); the primary side resonance inductor Lr and the resonance capacitor Cr participate in resonance, the resonance current iLr is a standard sine wave and is a negative value, and the excitation inductor current iLm is linearly increased but is smaller than the resonance current iLr;

switched mode 2[ t ]₁，t₂]: as shown in fig. 6, at t₁At the moment, the resonant current iLr crosses zero; the rectifier diode D1 and the rectifier diode D4 continue to be conducted; the voltage across the magnetizing inductor Lm is output and clamped to nVO; the primary side resonance inductor Lr and the resonance capacitor Cr participate in resonance, the resonance current iLr is a standard sine wave and is a positive value, and the excitation inductor current iLm is linearly increased but is smaller than the resonance current iLr;

switching mode 3[ t ]₂，t₃]: as shown in fig. 7, at t₂The switching tube S1 and the switching tube S4 are turned off at the moment, the resonant current iLr is still larger than the excitation inductor current iLm, and the rectifier diode D1 and the rectifier diode D4 are continuously turned on; the resonant current iLr is supplied to the switch tube S1 and the switch tube S4Output capacitor C_oss1、C_oss4Charging, and outputting a capacitor C to the switch tube S2 and the switch tube S3_oss2、C_oss3Discharged to the output capacitor C of the switch tube S5_oss5Discharging; when the capacitance C_oss5When the voltage at the two ends drops to zero, the body diode tube of the switch tube S5 is conducted, so that a condition is provided for realizing zero voltage switching-on of the switch tube S5;

switch mode 4[ t ]₃，t₄]: as shown in fig. 8, at t₃When the switch tube S5 is switched on at zero voltage, the switch tube S6 is continuously conducted with the rectifier diode D1 and the rectifier diode D4; the excitation inductor Lm is still clamped by the output voltage, the excitation current iLm continues to increase linearly, and the resonant current iLr decreases linearly;

switching mode 5[ t ]₄，t₅]: as shown in fig. 9, at t₄At the moment, the resonant current iLr is equal to the exciting current iLm, the current flowing through the rectifier diode D1 naturally passes through 0, and the secondary rectifier diode D1 and the rectifier diode D4 are switched off at zero current, so that the problem of reverse recovery of the diodes is solved; the switch tube S5 and the switch tube S6 are continuously conducted, and the exciting current and the resonant current iLr are equal and are kept unchanged;

switched mode 6[ t ]₅，t₆]: as shown in FIG. 10, t₅At the moment, the switch tube S6 is turned off, and the switch tube S5 continues to be turned on; the resonant current iLr is equal to the exciting current iLm, and the secondary rectifier diode is still in a reverse cut-off state; the resonant current iLr is supplied to the switch tube S1 and the output capacitor C of the switch tube S4_oss1、C_oss4Charging the output capacitor C of the switch tube S2 and the switch tube S3_oss2、C_oss3Discharged to the output capacitor C of the switch tube S6_oss6Charging; when the capacitance C_oss2、C_oss3When the voltage of the two ends is reduced to 0, the body diode tubes of the switch tube S2 and the switch tube S3 are conducted, so that a condition is provided for realizing zero-voltage switching-on of the switch tube S2 and the switch tube S3; at time t6, ZVS is realized by the switching tube S2 and the switching tube S3, and the circuit enters the second half cycle.

When the input voltage of the converter is 36V-75V, the converter operates in a half-bridge mode, i.e. the switch tube S2 is constantly turned off, and the switch tube S4 is constantly turned on, fig. 11 shows that the resonant converter adopts fixed-frequency PWM controlIn the main operating waveform diagram during the manufacturing process, Vgs1 is a driving signal of a switching tube S1, Vgs2 is a driving signal of a switching tube S2, Vgs3 is a driving signal of a switching tube S3, Vgs4 is a driving signal of a switching tube S4, Vgs5 is a driving signal of a switching tube S5, Vgs6 is a driving signal of a switching tube S6, Vc, iLr, iLm, i₀Respectively representing the voltage across Cr, the current through Lr, the current through Lm and the current through resistor R₀The current of (2). As can be seen from FIG. 11, the present invention outputs a current I₀The change is gentle and the stress of the device is small. The variator also has six switching modes in this half-cycle, as shown in figures 12-17 respectively.

Switched mode 1[ t ]₀，t₁]: as shown in fig. 12, at t₀Before the moment, the switch tube S6 is conducted, the switch tube S5 is turned off, and the body diode bears reverse voltage and is cut off in the reverse direction; t is t₀At the moment, the switching tube S1 is switched on at zero voltage; the rectifier diode D1 and the rectifier diode D4 are conducted, and the current flowing through the diodes is in direct proportion to the difference value of the resonance current and the excitation current; the voltage at two ends of the excitation inductor Lm is output and clamped to nV_O(ii) a The primary side resonance inductor Lr and the resonance capacitor Cr participate in resonance, the resonance current iLr is a standard sine wave and is a negative value, and the excitation inductor current iLm is linearly increased but is smaller than the resonance current iLr;

switched mode 2[ t ]₁，t₂]: as shown in fig. 13, at t₁At the moment, the resonant current iLr crosses zero; the rectifier diode D1 continues to conduct; the voltage at two ends of the excitation inductor Lm is output and clamped to nV_O(ii) a The primary side resonance inductor Lr and the resonance capacitor Cr participate in resonance, the resonance current iLr is a standard sine wave and is a positive value, and the excitation inductor current iLm is linearly increased but is smaller than the resonance current iLr;

switching mode 3[ t ]₂，t₃]: as shown in fig. 14, at t₂The switching tube S1 and the switching tube S4 are turned off at the moment, the resonant current iLr is still larger than the excitation inductor current iLm, and the rectifier diode D1 and the rectifier diode D4 are continuously turned on; the resonant current iLr is supplied to an output capacitor C of a switching tube S1_oss1Charging, and outputting a capacitor C to a switch tube S3_oss3Discharged to the output capacitor C of the switch tube S5_oss5Discharging; when the capacitance C_oss5When the voltage at the two ends drops to zero, the body diode of the switch tube S5 is conducted, so as to provide a condition for the switch tube S5 to realize zero voltage switching-on.

Switch mode 4[ t ]₃，t₄]: as shown in fig. 15, at t₃When the switch tube S5 is switched on at zero voltage, the switch tube S6 is continuously conducted with the rectifier diode D1 and the rectifier diode D4; the excitation inductor Lm is still clamped by the output voltage, the excitation current iLm continues to increase linearly, and the resonant current iLr decreases linearly;

switching mode 5[ t ]₄，t₅]: as shown in fig. 16, at t₄At the moment, the resonant current iLr is equal to the exciting current iLm, the current flowing through the rectifier diode D1 naturally passes through 0, and the secondary rectifier diode D1 and the rectifier diode D4 are switched off at zero current, so that the problem of reverse recovery of the diodes is solved; the switch tube S5 and the switch tube S6 are continuously conducted, and the exciting current and the resonant current iLr are equal and are kept unchanged;

switched mode 6[ t ]₅，t₆]: as shown in FIG. 17, t₅At the moment, the switch tube S6 is turned off, and the switch tube S5 continues to be turned on; the resonant current iLr is equal to the exciting current iLm, and the secondary rectifier diode is still in a reverse cut-off state; the resonant current iLr is supplied to an output capacitor C of a switching tube S1_oss1Charging the output capacitor C of the switch tube S3_oss3Discharged to the output capacitor C of the switch tube S6_oss6Charging; when the capacitance C_oss3When the voltage of the two ends is reduced to 0, the body diode of the switch tube S3 is conducted, and a condition is provided for the switch tube S3 to realize zero voltage switching-on; at time t6, ZVS is realized by switching tube S3 and the circuit enters the second half cycle.

The high voltage and low voltage ranges are divided by the gain range as a standard. For example, if the gain range of the full-bridge mode is 0.5 to 1 and the gain at 18V (the lowest input voltage) is 1, the range of the voltage segment (low voltage segment) operating in the full-bridge mode is 18 to 36V (36: 18 × 1/0.5), and 36 to 75V belongs to the high voltage segment, in the input voltage range of 18 to 75 VDC. The wider the gain range, the lower the circuit efficiency, and the span of the gain range needs to be considered when dividing the voltage range, and the continuity of the gain ranges corresponding to the high voltage and the low voltage is ensured after superposition.

According to the description of the working process of the converter, all the switching devices of the converter can realize zero-voltage switching-on, the rectifying devices on the secondary side can realize zero-current switching-off, the problem of reverse recovery of diodes does not exist, and all the switching devices are in a soft switching working state.

The invention adopts a full-bridge/half-bridge combined variable topology circuit as an inverter circuit, the gain range of the half-bridge is half of that of the full-bridge, and the gain range of the invention can be greatly widened and the working efficiency of the invention can be improved through the conversion of the topology structure.

The bidirectional switch is additionally arranged in the resonant cavity, the output voltage stabilization can be realized by controlling the conduction time of the bidirectional switch, so that the fixed frequency control is realized, the requirements on magnetic elements such as a transformer are reduced, the switch device and the inverter circuit are positioned on the same side of the transformer, the circuit control and driving difficulty is reduced, moreover, the bidirectional switch, the resonant inductor Lr and the resonant capacitor Cr form a voltage reduction circuit, the stress of the device is small, in short, the LLC resonant converter with the structure reduces the design difficulty of the LLC resonant converter, and can adopt fixed frequency PWM control.

When the bidirectional switch is conducted, the energy of the resonant current is stored in a loop formed by the excitation inductor Lm, the resonant inductor Lr and the bidirectional switch in the circulation stage, and does not flow through the resonant capacitor Cr, and the parasitic resistance of the resonant capacitor Cr is beneficial to reducing the loss of the energy on the parasitic resistance of the resonant capacitor Cr, so that the working efficiency of the circuit is improved.

The invention adopts fixed frequency PWM control and can be respectively combined with half-bridge and half-bridge topologies, thereby realizing wider voltage gain range and higher efficiency by matching with a circuit and enabling the converter to be suitable for occasions with wider gain range requirements. The control method has the advantages of small frequency conversion range, low requirements on magnetic elements such as transformers and inductors, no leading bridge arm and lagging bridge arm, wide voltage gain range and high efficiency compared with frequency conversion control even if the control method does not combine a variable topology.

The above embodiments are only for the understanding of the inventive concept of the present application and are not intended to limit the present invention, and any modification, equivalent replacement, improvement, etc. made by those skilled in the art without departing from the principle of the present invention should be included in the protection scope of the present invention.