阅读说明：本技术 用于窄带物联网干扰抑制的滤波器构建方法 (Filter construction method for narrowband Internet of things interference suppression ) 是由 匡红刚 易鹏飞 张召涛 钟慧 张雪 殷家敏 王郑兴 姚远 于 2021-08-02 设计创作，主要内容包括：本发明涉及无线通信技术领域,具体公开了一种用于窄带物联网干扰抑制的滤波器构建方法,该方法首先基于NB-IoT信号与误差信号在高阶统计域上的差异性,构建从混合信号中滤除干扰信号的代价函数进一步基于该代价函数构建滤波器,实现三阶累积量意义下的最佳滤波,实现对Chirp/DME/GSM信号在高阶域上进行准确估计,而不用受到NB-IoT(高斯有色噪声)信号影响,能够从混合的NB-IoT信号中准确减去Chirp/DME/GSM估计值,最终实现对Chirp/DME/GSM干扰信号的自适应抑制。(The invention relates to the technical field of wireless communication, and particularly discloses a filter construction method for narrowband Internet of things interference suppression Further based on the cost function A filter is constructed, optimal filtering under the meaning of third-order cumulant is realized, accurate estimation of a Chirp/DME/GSM signal on a high-order domain is realized without being influenced by an NB-IoT (Gaussian colored noise) signal, a Chirp/DME/GSM estimation value can be accurately subtracted from a mixed NB-IoT signal, and finally interference on the Chirp/DME/GSM signal is realizedAdaptive suppression of numbers.)

1. The filter construction method for narrowband Internet of things interference suppression is characterized by comprising the following steps:

s1: based on the difference of the interference signal and the narrowband Internet of things signal in a high-order statistical domain, constructing a cost function for filtering the interference signal from the mixed signal:

wherein, cum⁽³⁾(-) represents the pairs e (n), e (n-tau)₁)、e(n-τ₂) Performing a third order cumulant calculation, e (-) representing the estimated error between the actual output signal and the expected signal, τ₁And τ₂Representing two time intervals before time n, p (n-m) representing the mixed signal p (n) after m taps, y (-) representing the desired output,representing the actual output; m represents the dimension of p (n), and also represents that the filter has M taps, and M tap weight coefficients are correspondingly arranged; w is a_m(n) a tap weight vector matrix w (n) indicating time n ═ w₀(n)w₁(n)…w_M-1(n)]The weight vector of the m-th tap, E [ ·]Expressing the expectation;

s2: and constructing a corresponding filter according to the cost function.

2. The filter construction method for narrowband internet of things interference suppression according to claim 1, wherein the interference signal is a Chirp signal, a DME signal, or a GSM signal;

the difference between the interference signal and the narrowband internet of things signal in a high-order statistical domain is specifically as follows:

the cumulant of the narrow-band Internet of things signal above three orders is always 0;

the accumulation amount of the interference signal above the third order is not 0.

3. The method for constructing a filter for narrowband internet of things interference suppression according to claim 1, wherein the purpose of the filter is to suppress interference of the narrowband internet of thingsThe value is reduced as much as possible, and the optimal weight value is searched towards the direction of negative gradient when the value is reduced, so that the adjustment amount of the tap weight vector of the filter is equal to the adjustment amount of the tap weight vector of the filterWith reference to the negative gradient of (a), the tap weight vector gradually converges to a certain value, i.e. to a certain value, during the iterationAnd then, the tap weight vector obtains an optimal solution.

4. The filter construction method for narrowband internet of things interference suppression according to claim 3, wherein the cost function isThe gradient of (d) is expressed as:

wherein, C₁、C₂、C₃、C₄、C₅、C₆、C₇Denotes a number from p (n), p (n-tau)₁)、p(n-τ₂)、y(n)、y(n-τ₁)、y(n-τ₂) 7 kinds of cubic polynomial matrixes of which any 3 items are arranged and combined, wherein M multiplied by M matrixes are formed byConsistently, it is:

c_ppp(n,i,j),i、j＝[0,M-1]representing the third-order accumulated quantity value of any cubic polynomial in the square matrix at the time of n;

in which a 1 xM row matrix is formed byConsistently, it is:

c_ppy(n,0,j),j＝[0,M-1]representing the third order accumulated magnitude of any cubic polynomial in the row matrix at time n.

5. The filter construction method for narrowband internet of things interference suppression according to claim 4, characterized by iteratively calculating the cumulative magnitude at n time by using the cumulative magnitude estimated value of each cubic polynomial at n-1 time; iteratively calculating cumulative magnitudesFor representation, the corresponding calculation is:

6. the filter construction method for narrowband internet of things interference suppression according to claim 5, wherein a parameter step size μ (n) determining a response speed of the filter is dynamically adjusted according to:

μ(n)＝ψ·(exp(β·e(n))-θ)

here, ψ, β, and θ are constants that are adjusted at any time according to the tracking convergence performance of the adaptive filter.

7. The filter construction method for narrowband internet of things interference suppression according to any one of claims 1 to 6, characterized by comprising: the filter adopts a single-input transverse structure, is provided with M taps and correspondingly has M tap weight coefficients; the input of the signals from the same signal source at the taps can be expressed as u (n), u (n-1), …, u (n-M +1), which form an input vector of dimension M x 1, matched with the tap weight vector of the filter; wherein u (n) and u (n-1), …, u (n-M +1) correspond to the input value of the filter at the current time and the past time respectively, and the filter has M-1 delay units in total;

the adaptive operation of the filter is to calculate the estimated error e (n) between the actual output signal and the expected signal, and to track the change of the error signal e (n) at each moment in time during the operation of the filtering algorithm, and to make corresponding adjustments to the weight vectors of the taps of the filter based on the cost function.

Technical Field

The invention relates to the technical field of wireless communication, in particular to a filter construction method for narrowband internet of things interference suppression.

Background

The narrowband internet of things (NB-IoT) is a new 3GPP radio technology standard, and can solve the pursuit of the Internet of things on low power consumption and wide area services. The technology improves an indoor coverage mechanism, supports a large number of low-throughput devices, is extremely low in cost and power consumption, and is optimized in a network architecture. The NB-IoT is capable of supporting a large number of connections, ultra-low power consumption, wide area coverage and triggering between bi-directional signaling planes. It is also supported by excellent cellular communication networks. Therefore, NB-IoT is a very promising technology.

Even though NB-IoT is a thriving technology, it still has the following problems: network coverage distance, deployment and long-term support, network congestion, insufficient power consumption, information security risks, business profitability, interference coexistence, and the like. This also results in NB-IoT internet of things connections accounting for less than 10% of the total cellular internet of things connections worldwide. In particular, NB-IoT systems are deployed in coexistence with deployed systems that mainly include radar systems and communication signals with similar operating frequency bands, typically Chirp, DME, and GSM signals. The Chirp/DME/GSM signal is additive noise relative to the NB-IoT signal and is co-channel interference, the mixed signal shows that the two signals overlap each other in the time-frequency domain, the NB-IoT signal is seriously distorted, the signal enters a nonlinear area of a radio frequency front-end amplifying circuit of a receiver to generate radio frequency nonlinear distortion due to the distortion, and the synchronous receiving performance of the NB-IoT is also seriously deteriorated along with frequency and timing deviation caused by the distortion. If the Chirp/DME/GSM Interference is processed at the baseband, Inter-Carrier Interference (ICI) and Inter-Symbol Interference (ISI) may be caused, and the signal-to-noise ratio of the receiving end may be reduced.

When a Chirp/DME/GSM and NB-IoT system coexist, the mutual influence and interference suppression problems of the Chirp/DME/GSM and the NB-IoT system are not systematically researched by domestic and foreign research institutions at present.

Disclosure of Invention

The invention provides a filter construction method for narrowband Internet of things interference suppression, and solves the technical problems that: how to effectively suppress interference signals (Chirp/DME/GSM signals) mixed in NB-IoT signals through a filter.

In order to solve the technical problems, the invention provides a filter construction method for narrowband internet of things interference suppression, which comprises the following steps:

s1: based on the difference of the interference signal and the narrowband Internet of things signal in a high-order statistical domain, constructing a cost function for filtering the interference signal from the mixed signal:

wherein, cum⁽³⁾(-) represents the pairs e (n), e (n-tau)₁)、e(n-τ₂) Performing a third order cumulant calculation, e (v) representing the estimated error between the actual output signal and the desired signal, τ₁And τ₂Representing two time intervals before time n, p (n-m) representing the mixed signal p (n) after m taps, y (-) representing the desired output,representing the actual output; m represents the dimension of p (n), and also represents that the filter has M taps, and M tap weight coefficients are correspondingly arranged; w is a_m(n) a tap weight vector matrix w (n) indicating time n ═ w₀(n) w₁(n)…w_M-1(n)]The weight vector of the m-th tap, E [ ·]Expressing the expectation;

s2: and constructing a corresponding filter according to the cost function.

Specifically, the interference signal is a Chirp signal, a DME signal or a GSM signal;

the difference between the interference signal and the narrowband internet of things signal in a high-order statistical domain is specifically as follows:

the cumulant of the narrow-band Internet of things signal above three orders is always 0;

the accumulation amount of the interference signal above the third order is not 0.

Further, the purpose of the filter is to beThe value is reduced as much as possible, and the optimal weight value is searched towards the direction of negative gradient when the value is reduced, so that the adjustment amount of the tap weight vector of the filter is equal to the adjustment amount of the tap weight vector of the filterWith reference to the negative gradient of (a), the tap weight vector gradually converges to a certain value, i.e. to a certain value, during the iterationAnd then, the tap weight vector obtains an optimal solution.

Further, the cost functionThe gradient of (d) is expressed as:

wherein, C₁、C₂、C₃、C₄、C₅、C₆、C₇Denotes a number from p (n), p (n-tau)₁)、p(n-τ₂)、y(n)、y(n-τ₁)、y(n-τ₂) 7 kinds of cubic polynomial matrixes of which any 3 items are arranged and combined, wherein M multiplied by M matrixes are formed byThe consistency is kept between the first and the second,comprises the following steps:

c_ppp(n,i,j),i、j＝[0,M-1]representing the third-order accumulated quantity value of any cubic polynomial in the square matrix at the time of n;

in which a 1 xM row matrix is formed byConsistently, it is:

c_ppy(n,0,j),j＝[0,M-1]representing the third order accumulated magnitude of any cubic polynomial in the row matrix at time n.

Further, the accumulated quantity value of the n moment is calculated iteratively through the accumulated quantity estimated value of each cubic polynomial at the n-1 moment; iteratively calculating cumulative magnitudesFor representation, the corresponding calculation is:

further, the step size μ (n) of the parameter determining the response speed of the filter is dynamically adjusted according to the following formula:

μ(n)＝ψ·(exp(β·e(n))-θ)

here, ψ, β, and θ are constants that are adjusted at any time according to the tracking convergence performance of the adaptive filter.

In a preferred embodiment, the filter adopts a single-input horizontal structure, and has M taps, corresponding to M tap weight coefficients; the input of the signals from the same signal source at the taps can be expressed as u (n), u (n-1), …, u (n-M +1), which form an input vector of dimension M x 1, matched with the tap weight vector of the filter; wherein u (n) and u (n-1), …, u (n-M +1) correspond to the input value of the filter at the current time and the past time respectively, and the filter has M-1 delay units in total;

the adaptive operation of the filter is to calculate the estimated error e (n) between the actual output signal and the expected signal, and to track the change of the error signal e (n) at each moment in time during the operation of the filtering algorithm, and to make corresponding adjustments to the weight vectors of the taps of the filter based on the cost function.

The invention provides a filter construction method for narrowband Internet of things interference suppression, which comprises the steps of firstly constructing a cost function for filtering interference signals from mixed signals based on the obvious difference of NB-IoT signals and error signals in a high-order statistical domainFurther based on the cost functionA filter is constructed, optimal filtering in the sense of third-order cumulant is achieved, accurate estimation of a Chirp/DME/GSM signal in a high-order domain is achieved, the Chirp/DME/GSM estimated value can be accurately subtracted from a mixed NB-IoT signal without being influenced by an NB-IoT (Gaussian colored noise) signal, and finally adaptive suppression of the Chirp/DME/GSM interference signal is achieved.

Drawings

Fig. 1 is a flowchart of a filter construction method for narrowband internet of things interference suppression provided in this embodiment;

fig. 2-1 is a constellation diagram of NB-IoT signals added only with white gaussian noise provided by the present embodiment;

fig. 2-2 is a bit error rate graph of NB-IoT signals added only with white gaussian noise provided by the present embodiment;

fig. 3-1 and 3-2 are a time domain waveform diagram and an amplitude spectrogram of the Chirp signal provided in this embodiment, respectively;

FIGS. 4-1 and 4-2 are a time domain waveform diagram and a power spectrum diagram of the DME signal provided by the present embodiment, respectively;

fig. 5-1, 5-2, and 5-3 are a block diagram, a time domain waveform diagram, and a phase waveform diagram of the GSM signal provided in this embodiment, respectively;

fig. 6-1 is a third order cumulative quantity estimation diagram of NB-IoT signals of GSM signals provided by the present embodiment;

6-2, 6-3, and 6-4 are graphs of third-order accumulated quantities of the Chirp signal, the DME signal, and the GSM signal provided by the present embodiment, respectively;

fig. 7 is an overall flow diagram of interference suppression for NB-IoT signals provided by the present embodiment;

fig. 8 is a schematic diagram of a filter provided in the present embodiment;

fig. 9 is a schematic diagram of an LMS algorithm using a lateral structure according to the present embodiment;

fig. 10-1 is an NB-IoT signal constellation diagram with gaussian white noise only added in the experiment provided by the present embodiment;

fig. 10-2 is a constellation diagram of the NB-IoT signal provided in the present embodiment in the presence of Chirp signal interference;

fig. 10-3 is a constellation diagram of an NB-IoT signal with Chirp signal interference processed by the present method according to this embodiment;

fig. 10-4 is a bit error rate diagram of the NB-IoT signal with Chirp signal interference processed by the present method according to this embodiment;

fig. 11-1 is a constellation diagram of NB-IoT signals provided by the present embodiment in the presence of DME signal interference;

fig. 11-2 is a constellation diagram of an NB-IoT signal with DME signal interference processed by the present method according to this embodiment;

fig. 11-3 is a bit error rate diagram of NB-IoT signals with DME signal interference processed by the present method according to the present embodiment;

fig. 12-1 is a constellation diagram of NB-IoT signals provided by the present embodiment in the presence of GSM signal interference;

fig. 12-2 is a constellation diagram of NB-IoT signals with GSM signal interference processed by the present method according to this embodiment;

fig. 12-3 is a bit error rate diagram of NB-IoT signals with GSM signal interference processed by the present method.

Detailed Description

The embodiments of the present invention will be described in detail below with reference to the accompanying drawings, which are given solely for the purpose of illustration and are not to be construed as limitations of the invention, including the drawings which are incorporated herein by reference and for illustration only and are not to be construed as limitations of the invention, since many variations thereof are possible without departing from the spirit and scope of the invention.

In order to effectively filter interference signals (mainly referred to as Chirp signals, DME signals, or GSM signals) mixed in NB-IoT signals at a receiving end, an embodiment of the present invention provides a filter construction method for narrowband internet of things interference suppression, as shown in fig. 1, including the steps of:

s1: based on the difference of the interference signal and the narrowband Internet of things signal in a high-order statistical domain, constructing a cost function for filtering the interference signal from the mixed signal:

wherein, cum⁽³⁾(-) represents the pairs e (n), e (n-tau)₁)、e(n-τ₂) Performing a third order cumulant calculation, e (-) representing the estimated error between the actual output signal and the expected signal, τ₁And τ₂Representing two time intervals before time n, p (n-m) representing the mixed signal p (n) after m taps, y (-) representing the desired output,representing the actual output; m represents the dimension of p (n), and also represents that the filter has M taps, and M tap weight coefficients are correspondingly arranged; w is a_m(n) a tap weight vector matrix w (n) indicating time n ═ w₀(n) w₁(n)…w_M-1(n)]The weight vector of the m-th tap, E [ ·]Expressing the expectation;

s2: and constructing a corresponding filter according to the cost function.

Before the steps are formally specified, the NB-IoT, Chirp signal, DME signal, and GSM signal need to be introduced as necessary.

The physical layer block diagram of NB-IoT is shown in fig. 2, and includes three parts, a transmitting end, a receiving end, and a channel environment. In this embodiment, a QPSK modulation method is adopted to modulate and map a baseband signal at a transmitting end, and a data stream obtained by modulation also needs to be subjected to serial-to-parallel conversion (S/P) to obtain multiple parallel sub-data streams, and then IFFT (inverse fast fourier transform) operation is performed on the parallel data streams to transform the parallel data streams from a frequency domain to a time domain. Since NB-IoT has 48 subcarriers and the IFFT algorithm is based on 64 points, there is also data rearrangement, i.e. mapping 48 subcarriers to corresponding subcarriers, and filling zero at other points, in actual simulation, to reduce inter-subcarrier interference, 48 subcarriers are mapped to "two ends" of 64 subcarriers, leaving out the middle carrier. And then adding a cyclic prefix to obtain a time domain signal, and finally transmitting the time domain signal to a receiving end through a channel after parallel-serial conversion (P/S). Since an NB-IoT terminal does not always have a direct-view path when deployed in a complex power electromagnetic environment, the channel environment may be considered as a rayleigh channel, but the physical layer simulation is performed in the present embodiment, and this process is simplified, so that only Additive White Gaussian Noise (AWGN) is added. And then the receiving end carries out serial-to-parallel conversion (S/P), cyclic prefix removal, FFT (fast Fourier transform) operation and parallel-to-serial conversion (P/S) on the received signal in sequence, and finally demodulates the data stream to recover the data of the transmitting end baseband.

Simulation is performed according to the above block diagram and parameter settings, and the result is shown in fig. 2-1 and fig. 2-2, where fig. 2-1 is a constellation diagram corresponding to an NB-IoT signal received by a receiving end when a channel is an ideal channel (only gaussian white noise is added) and a signal-to-noise ratio is 20 dB. Fig. 2-2 is a simulation diagram of bit error rate performance of the NB-IoT system, and it can be seen that under the gaussian channel, the simulated NB-IoT bit error rate and the theoretically calculated bit error rate curve are substantially consistent, which indicates that the simulation procedure is correct. In fact, under gaussian channel, the bit error rate of OFDM and general digital communication should be identical. The strongest advantage of OFDM is multipath resistance and fading resistance, and the performance of OFDM can be embodied only in the channel environment with multipath and fading.

The Chirp signal is named Chirp signal (zh ō u ji g) and has linear relation between its frequency and time to obtain linear frequency modulated signal, so that it has high resistance to interference and Doppler shift and can meet the requirements of high delay resolution and fine timing.

The complex Chirp signal with rectangular envelope can be represented as:

wherein A (t) is a gate function, f₀For the radar radio frequency center frequency, k is the chirp rate, i.e., the chirp coefficient or the tone frequency. A (t) can be represented as:

t is the duration.

The instantaneous frequency of the Chirp signal is:

as can be seen from equation (1.3), the rate of change of the instantaneous frequency of the Chirp signal can be represented by a frequency modulation rate k, where the frequency modulation rate represents the Chirp of the Chirp signal, and k is a constant that can determine the direction of frequency sweep of the Chirp signal and is linearly related to the direction: the Up-Chirp signal and Down-Chirp signal correspond to the cases when k >0 and k <0, respectively. From the equation (1.3), it can be derived that the bandwidth of the Chirp signal satisfies B ═ k × T, and thus, the bandwidth of the Chirp signal also satisfies a proportional relationship with the modulation frequency k, as well as the frequency. Therefore, when the Chirp signal interferes with other signals, not only the interference of the duration to the time period but also the interference to the frequency band need to be considered.

As shown in fig. 3-1, a time domain diagram of a Chirp signal generated by MATLAB shows that the waveform of the signal shows characteristics such as a shorter period and an increased frequency of the waveform as time increases. The fresnel integration characteristic causes the amplitude spectrum of the signal to fluctuate at the top, and as can be seen from the amplitude spectrum (amplitude-frequency response curve) of the Chirp signal shown in fig. 3-2, the amplitude spectrum of the Chirp signal is not completely limited within the Chirp signal sweep bandwidth B.

The DME short-range navigation system is widely applied to navigation positioning of the airplane and control and monitoring of the flight path of the airspace by the ground station. The specified workflow is: the airborne equipment sends DME signals actively, and response pulses of the ground station are sent at an interval of 50us (the pulse transmission frequency of the DME signals is 2700 times/second) after the signals are received, so that the response time is uniform. Therefore, the angle distance information of the airplane and the ground station can be calculated according to the interval time between two pulses and the propagation speed of the electromagnetic wave. The peak power of the DEM system can reach between 50W and 1kW, the sensitivity of-10 dBm to-63 dBm can support the ranging under the condition that the DEM system responds 60-75%, the precision can reach 0.185km, and the distance query request of hundreds of airplanes can be simultaneously supported within the coverage radius range of 370 km.

The DME system operating frequency band is 962-1213MHz, wherein the ground responder can work in the whole DME frequency band, but the airborne equipment can only work in 1025-1150MHz frequency band. The frequency band of the DME system can be divided into 256 channels spaced 1MHz apart, where the number of X and Y channels is 126 each, and there is a 63MHz separation between the transmit and receive frequencies of each channel.

The DME signal is represented in a time domain by a pair of Gaussian pulse signals with fixed intervals, and the mathematical expression of the DME signal is shown as a formula (2.4).

Wherein the constant alpha is 4.5 x 10¹¹And Δ t is the pulse interval. The fixed pulse pair intervals Δ t corresponding to the X and Y channels are 12us and 36us, respectively.

The time domain waveform of the DME signal is shown in FIG. 4-1 with a pulse duration (width) T of 3.5 us. After fourier transform, the signal with gaussian characteristic in time domain still has gaussian characteristic in frequency domain, as shown in fig. 4-2, the DME signal frequency spectrum shows a gaussian pulse group with channel gap of 1MHz, and its mathematical expression is:

global System for Mobile Communication (GSM) is the most widely accepted standard in the telecommunications industry and has been implemented worldwide. It is widely used in digital mobile communication because of its compact spectrum and good error characteristics. Gaussian Minimum Shift Keying (GMSK) is a key technology of the GSM system, the GMSK modulation scheme is developed based on Minimum Shift Keying (MSK), and the GMSK modulation scheme is superior in that the requirement of carrier switching on hopping energy can be reduced as much as possible, so that the channel spacing can be made tighter while maintaining the same data transmission rate. Compared with the MSK, the GMSK has the advantages of narrower frequency band, smoother frequency spectrum, simpler realization and stronger anti-jamming capability.

The GSM signal is generated as shown in fig. 5-1, and data is passed through a gaussian filter and then MSK modulated to obtain a GSM signal.

The main parameters of the Gaussian low pass filter design in FIG. 5-1 are its 3dB bandwidth B and input symbol width T_bProduct time bandwidth constant BT_b。

The mathematical expression of the GSM signal generated after GMSK modulation is as follows:

wherein: carrier angular frequency of w_c(ii) a And T_bThen represents the symbol width; a is_nIs the non-return-to-zero data of the input. g (t) is the rectangular impulse response of the gaussian filter. Since g (t) is ideally infinite in length, this is not the caseEngineering implementation, therefore, a truncated approximation of g (t) is needed, and the method is a window function. The truncation length is (2N +1) T_bThe requirements for relative intersymbol interference are met. In specific implementation, the response of the Gaussian filter needs to pass through the filter with the length of 3T_bOr 5T_bTo make a truncation approximation. Out-of-band energy passing through 3T_bAnd 5T_bThe rectangular window of length is truncated to a ratio of 0.7% and 1.5124 × 10^-8. In specific calculation, in order to obtain sufficient precision, the truncation length of g (T) is usually taken to be 5T_b. Therefore, the truncation length of g (T) taken in this embodiment is 5T_b。

The parameter configuration is the symbol widthTime bandwidth constant BT_b0.3, the number of samples per carrier is 64.

The time domain waveform and the phase waveform of the GSM signal with data length 20 generated by MATLAB are shown in fig. 5-2 and fig. 5-3, it can be seen from fig. 5-2 that the GSM signal is a sinusoidal signal whose phase is modulated, and consistent with the expression, it can be seen from fig. 5-3 that the phase of the GSM signal is continuously changed, and at the 4 th and 18 th data, the phase appears to jump because the phase takes the modulus of 2 pi, and is actually continuous, but the reason is that a gaussian premodulation filtering process is added before the digital signal is modulated.

The k order statistic has a low order and a high order, and is called a low order statistic when k is 1,2, and representative first order and second order statistics respectively have a mean value, a variance and the like, and is called a high order statistic when k is more than or equal to 3. Its main forms are high-order moments, high-order cumulants and their spectra, and it has several forms, whether it is a random variable or a random process.

When a random signal with a Gaussian distribution characteristic is processed, the statistical characteristics of the signal can be sufficiently represented by first-order and second-order statistics if viewed from a statistical point of view. For example, a joint probability density function for a random variable can be derived from its first order statistics (mathematical expectation) and second order statistics (covariance matrix), although only variables that obey a gaussian distribution have this property; as for the stochastic process, as long as it obeys a gaussian distribution, its probability density structure can be derived from the mean (first order), autocorrelation function, or autocovariance function (second order), and its overall statistical properties can be obtained. However, many signals in practical engineering do not follow gaussian distribution, such as Chirp/DME/GSM signals studied in this embodiment, if only the first-order and second-order statistics are analyzed, sufficient information cannot be obtained, and therefore, it is necessary to study their high-order statistics.

The correlation definition can be derived by adopting a characteristic function method, and for a certain stable random process, the high-order statistical characteristics of the stable random process can be described by using moments and cumulants of multidimensional random variables. Suppose that some deterministic zero-mean stationary random process is represented by x (n), and x₁＝x(n)，x₂＝x(n+τ₁)，…，x_k＝x(n+τ_k-1) Then the k-dimensional stationary random variable X ═ X₁,x₂,…,x_k]The joint probability density function that can be used to characterize x (n) can be represented by f (x)₁,x₂,…,x_k) Is expressed by f (x)₁,x₂,…,x_k) The first characteristic function phi (omega) obtained after the fourier inversion is also a first characteristic function phi (omega) of the k-dimensional random variable X, and a second characteristic function psi (omega), also called an cumulant mother function, is generated by taking a natural logarithm of the first characteristic function, and the two characteristic functions are shown as formulas (2.1) and (2.2). Wherein ω is ═ ω₁,ω₂,…,ω_k]E {. cndot } represents the mathematical expectation, ln is logarithmic based on the natural constant E.

In this embodiment, the matrix with "T" in the upper right corner represents the transposed matrix of the original matrix.

The k-th derivative of the first characteristic function phi (omega) of the stationary random variable X at the origin is equal to the k-th moment m of the stationary random process X (t)_kx(τ₁,…,τ_k-1) As shown in formula (2.3). As to the k-order cumulant c corresponding to its random process_kx(τ₁,…,τ_k-1) The calculation method is consistent with the k-order moment, that is, the value of the k-order derivative is obtained at the origin of the second characteristic function Ψ (ω), as shown in equation (2.4). The values of both being only related to the time interval tau₁,…,τ_k-1And is independent of time t (stationary random process behavior).

Higher-order moments and higher-order cumulants of stationary stochastic process x (t) if they satisfy the absolutely-neutralizable condition, their higher-order moment spectrum M_kx(ω₁,…,ω_k-1) And higher order cumulant spectra S_kx(ω₁,…,ω_k-1) (k-1) -dimensional discrete fourier transforms (so-called high order spectra are high order cumulant spectra) that are both their corresponding k-moments and k-cumulants, the transform formula of which is shown below:

the Gaussian random process in the communication system comprises Gaussian noise and Gaussian random signals, and as a special case of the stable random process, the research on the high-order statistical characteristics (high-order moments and cumulant) of the Gaussian random process has important theoretical significance for signal processing. For a gaussian random process x (t), if it satisfies the condition of zero mean and smoothness, the corresponding k-dimensional joint probability density function is:

wherein, C_xRepresenting a covariance matrix, mean vector m_x＝[m₁,m₂,...,m_k]. The mathematical expression of the covariance matrix is shown in formula (2.8), and if it satisfies the positive definite condition, the covariance R_x(i,j)＝E{(x_i-m_i)(x_j-m_j)}。

The stationary Gaussian process x (t) can be used to obtain a moment generating function according to equations (2.1) and (2.2)Sum cumulative generation function Ψ (ω) ═ jm_x ^Tω-ω^TC_xOmega/2. Phi (omega) and psi (omega) are substituted into formulas (2.3) and (2.4) to obtain a k-order moment m of a Gaussian random process x (t)_kx(τ₁,…,τ_k-1) And accumulated amount c_kx(τ₁,…,τ_k-1) Equal to:

from the above results, it is easy to obtain: firstly, the k-order moment of the Gaussian random process is always 0 in the odd-order, and the even-order is not equal to 0; and 2, any accumulated quantity of the Gaussian random process is 0, although the second-order accumulated quantity is an exception, so that the second-order accumulated quantity is equal to covariance, which is the theoretical basis that the Gaussian random signal can be inhibited in a high-order statistical domain.

OFDM technology is employed at the physical layer of NB-IoT systems to modulate multiple carriers. The idea is to divide the bandwidth of the NB-IoT system into 64 orthogonal subcarriers derived from parallel data streams that are modulated and mapped and converted from serial to parallel. The time domain signal of the OFDM is the superposition of the pulse shaped multi-channel subcarrier signals. Referring to the central limit theorem, the NB-IoT signal can be regarded as a stationary random process due to the progressive gaussian property exhibited by its time-domain envelope, and according to the above discussion of the high-order statistic, that is, the stationary gaussian process is theoretically 0 in third order and above, so the third order cumulant of the NB-IoT signal is also 0.

The method for estimating the third-order cumulant in the embodiment is a non-parametric method, and the correctness of the assumption is verified according to the method. The verification steps are as follows: let N observation samples of NB-IoT signals x (0), x (1),.... times.x (N-1) be zero-averaged. Due to the progressive gaussian nature of NB-IoT signals, it can be considered a stationary random process and its third order cumulant (lumped meaning) can be expressed as:

C_3,OFDM＝E{x(t)x(t+τ1)x(t+τ2)} (3.1)

e {. cndot } in this example both represent the expectation for the signal set space, τ 1 and τ 2 represent the two time intervals after time t.

For stationary random processes of each state, the third-order cumulant of the stationary random process in the lumped sense is equal to that in the time sense, so the third-order cumulant of the NB-IoT signal can be expressed as the following mathematical expression:

as can be seen from equation (3.2), the third-order cumulant estimate of the NB-IoT signal is equal to the average of the current time within the sliding window of length 2T multiplied by the sample data of time intervals τ 1 and τ 2. According to the above-mentioned idea, the non-parametric verification method firstly creates a sliding window with a sample length of M and N sample data, where the sliding range of the sliding window is l 1.

Fig. 6-1 shows a simulation diagram of calculating the third-order cumulant estimation values of NB-IoT signals with respect to τ 1 and τ 2 according to equation (3.3), where the ordinate is a mathematical value and represents the multiplication of 3 signals, and since the values are expressed, no normalization is performed here. To satisfy the NB-IoT signal zero-mean condition, the modulation data sequences in the simulation parameters are set to be independent and uniformly distributed. The NB-IoT system can support 64 subcarriers with the simulation condition assuming a subcarrier spacing of 3.75kHz, so the effective number of subcarriers is 48. The symbol length is 266.7 mus. As can be seen from the figure, NB-IoT signals are suppressed in the third order statistical domain, and the normalized third order cumulant estimation values are only 10^-6In the following, the estimated value can be regarded as 0 approximately, and the simulation result verifies the correctness of theoretical derivation, thereby providing a theoretical basis for the scheme of the embodiment.

The Chirp signal is known as a Chirp signal because of its linear relationship between frequency and time. The complex Chirp signal with rectangular envelope is shown in the formula (1.1) as follows:

wherein A (t) is a gate function, the Chirp signal is also limited in duration, and correlation coefficients such as frequency modulation rate are also known conditions, and the third-order cumulant calculation formula is as follows:

C_3,Chirp(τ1,τ1)＝∫S(t)S(t+τ1)S(t+τ1)dt (3.5)

the third-order cumulant of the Chirp signal can be obtained by bringing the formula (1.1) into the formula (3.5). The third order accumulation of the Chirp signal is estimated and the result is shown in fig. 6-2:

the DME signal is a short-range navigation radar signal and is mainly applied to positioning of aerial aircrafts. It appears in the time domain as a gaussian pulse pair with a fixed interval (12us or 36us), and its mathematical expression is as the above equation (2.4):

wherein α is a constant determining the pulse width, and α is 4.5 × 10¹¹The X and Y channels correspond to pulse intervals of 12us and 36us, respectively. The DME signal is finite in duration, known in both pulse width constant and pulse interval, and can be considered a deterministic signal whose third order cumulant is calculated as:

C_3,DME(τ1,τ2)＝∫p(t)p(t+τ1)p(t+τ2)dt (3.6)

in the formula, the time intervals τ 1 and τ 2 are independent of each other. By bringing formula (2.4) into formula (3.6), a third order accumulation of the DME signal can be obtained.

The third order accumulation of the DME signal was estimated and the results are shown in FIGS. 6-3. The circularly symmetric three-dimensional surface about time intervals τ 1 and τ 2 is the normalized third-order cumulant of the DME signal, and 7 2D gaussian functions with central coordinates of (0,0), (Δ t ), (0, Δ t), (- Δ t,0), (- Δ t ), (0, - Δ t) constitute the three-dimensional surface, and are represented in space as 1/2 with (0,0) as the center, 6 two-dimensional gaussian functions uniformly distributed and the peak being the central gaussian function. And correlation coefficient ρ is 0.5, variance σ²2/α is a common characteristic of each two-dimensional gaussian function. As α gradually increases toward infinity, the DME signal will appear in the time domain to degrade from a Gaussian signal to a pulsed signal, so that as α of the two-dimensional Gaussian function gradually increases, its variance σ²Will be close to 0, these functions will degenerate to 7 two-dimensional pulse functions, but still have central coordinates (0,0), (Δ t ), (0, Δ t), (- Δ t,0), (- Δ t ), and (0, - Δ t). The analysis shows that when the pulse constant and time interval of the DME signal are known, the normalized third order cumulative amount of the DME signal can be plotted by using the twoDescription is given; and the correlation exists between the two-dimensional Gaussian function and the pulse width of the signal, and the narrower the pulse width is, the steeper the two-dimensional Gaussian function is on the third-order cumulant simulation diagram.

The GSM signal is modulated by the GSM system through GMSK, and its mathematical expression is shown in the foregoing equation (2.6):

the GSM signal and the DME/Chirp signal are also limited in duration, the conditions of the symbol width, the carrier frequency and the like are known, and the third-order cumulant calculation method is also the same:

C_3,GSM(τ1,τ2)＝∫S_GSM(t)S_GSM(t+τ1)S_GSM(t+τ2)dt (3.7)

the third order cumulant of the GSM signal can be obtained by bringing formula (2.6) into formula (3.7). The third order cumulant of the GSM signal is estimated and the results are shown in fig. 6-4.

Analysis of the third-order cumulant simulation graphs of the four signals shows that the model established for the NB-IoT and Chirp/DME/GSM interference scenarios is feasible. The three interference signals are deterministic signals: the NB-IoT signal, due to its gaussian nature, has a third order cumulant of approximately 0; the third-order cumulant of the DME signal is expressed as 7 two-dimensional Gaussian functions, and parameters such as time interval, pulse constant and the like are determined; the third-order cumulant simulation plots for GSM and Chirp, although irregular and without normalization, are also clearly not 0. The reason is that NB-IoT and Chirp/DME/GSM signals have such large difference in the high-order statistical domain, and the NB-IoT signals and the Chirp/DME/GSM interference signals are independent. The design criteria for the Chirp/DME/GSM interference suppression technique is to find the largest difference between NB-IoT signal and the suppressed interference signal. Following the above principles and design concepts, the present embodiment suppresses influence of Chirp/DME/GSM on NB-IoT signals by adaptive filtering based on high-order statistics, as described in the foregoing steps S1 to S3, and described in more detail below.

From the foregoing analysis, one can conclude that:

the difference between the interference signal and the narrowband internet of things signal in the high-order statistical domain is specifically as follows:

the accumulation of the narrow-band Internet of things signals above three orders is always 0;

the cumulative amount of interference signals above the third order is not 0.

Based on the difference of the high-order statistical domain, the HOC-LMS algorithm (high-order statistics least mean square algorithm) is used for accurately estimating the Chirp/DME/GSM signal in the high-order domain without being influenced by the NB-IoT (Gaussian colored noise) signal, the Chirp/DME/GSM estimated value can be accurately subtracted from the mixed NB-IoT signal, and the self-adaptive suppression of the Chirp/DME/GSM interference signal can be realized.

Since the NB-IoT signal is interfered by the Chirp/DME/GSM signal in the transmission process, the NB-IoT system uses the HOC-LMS algorithm to perform adaptive filtering on the Chirp/DME/GSM interference signal at the receiving end. For two mutually independent random processes of x and y, the gaussian accumulations of them can be linearly summed, i.e. the higher order cumulant of the sum is equal to the sum of the respective higher order cumulants. The NB-IoT signal shows progressive Gaussian characteristics in the envelope of the time domain due to the OFDM system and can be equivalent to a stable random process, so that the third-order cumulant of the NB-IoT signal tends to 0, which is verified by simulation, and the NB-IoT and the Chirp/DME/GSM are independent from each other and show obvious difference characteristics in a high-order statistical domain. Therefore, the interference scenario of NB-IoT (gaussian color noise) superimposed Chirp/DME/GSM signal (determination signal) can be established as a quantization model, and the interference suppression flow diagram is shown in fig. 7. In order to suppress the influence of the above three interference signals on the NB-IoT signal, before filtering, the receiving end needs to perform correlation detection of Chirp/DME/GSM signals on the received signal, and for the DME signal, since the interval between two gaussian pulses is fixed to 12us or 36us, the correlation is very strong, and therefore R can be detected_y(t) y (t + Δ t) if the correlation R is positive_y(t) when exceeding a certain set threshold, it can be considered that DME interference exists at the current moment; for GSM signals, the detection method has scoresFourier transform and Radon-Wigner transform; for Chirp signals, detection can be made by the frequency characteristics of the signal. Since the power of the Chirp/DME/GSM signal is much higher than that of the NB-IoT signal, the NB-IoT signal has negligible effect on interference detection. If the existence of interference is detected, filtering the interference by using an adaptive algorithm based on high-order statistics, because only NB-IoT signals exist simultaneously and are suppressed in a high-order domain under three scenes, in the practical engineering of the HOC-LMS algorithm, the NB-IoT signals are used as interference signals, a useful signal p (t) (Chirp/GSM/DME) is accurately estimated in the high-order statistics domain according to the obvious difference between the NB-IoT signals and the interference signals, and then the estimated Chirp/GSM/DME signals p (t) are subtracted from a mixed signal y (t) of the Chirp/GSM/DME signals and the NB-IoT signals, so that an NB-IoT estimated signal x (t) after the interference is suppressed is finally obtained.

The adaptive filter is a filter which can automatically change own characteristics so as to obtain correct response at each moment, and the operation model of the adaptive filter is expressed mathematically, namely an adaptive algorithm. The adaptive algorithm works with the goal of an optimization criterion, one of which is LMS, to minimize the mean square error between the input signal and the output signal.

As shown in fig. 8, x (n) and d (n) are the observed signal and the expected output signal respectively, which are input to the filter at time n, and the error signal e (n) can be represented as the actual output y (n) and the difference value with the expected signal, what is done in the adaptive filtering is to run the filtering algorithm and track the change of the error signal e (n) at each time, and the variable parameters of the filter (usually, the tap coefficients of the filter) are adjusted accordingly to achieve perfect fitting/filtering effect.

The filter of the present embodiment adopts an adaptive filter, and the adaptivity of the adaptive filter is embodied in that the tap coefficient of the adaptive filter is adjustable, and the coefficient is continuously adjusted to minimize the objective function or realize the optimal criterion. How the tap coefficients are adjusted is controlled by a cost function, and the setting of the cost function and the way of adjusting the coefficients affect the convergence speed and stability of the filter. In general, the cost function is nonlinear, and is related to the system error or even directly is a mathematical variant of the system error function, so that some limitation problems in the linear case can be avoided. The cost function criterion can be mainly classified into two types according to the setting of the filter: one class follows the criterion of maximizing the output SNR of the filter, and the other class is the RMSE (minimizing mean square error) and its variants mentioned above, while the two cost functions correspond to typical filters of matched filter and wiener filter, respectively.

The structure of the adaptive filter can be divided into a horizontal structure and a vertical structure, and the data input mode corresponds to a single-input structure and a multi-input structure respectively. The input of signals from the same source at the taps (delay elements) can be represented as u (n), u (n-1), …, u (n-M +1), which form an input vector u (n) of dimension M x 1, thus matching exactly the tap weight vector of the filter, tap weight vector w₀(n),w₁(n),…,w_M-1(n) is also a vector of dimension M × 1. Wherein u (n) and u (n-1), …, u (n-M +1) correspond to the input values of the filter at the current and past time respectively, and the filter has M-1 delay units in total. In order to fulfill the criteria, the two mutually independent processes of filtering and adaptation are lacking. The filtering process operates by subjecting the input signal u (n) to a series of delay elements and weighted sums of tap coefficients to obtain an estimate of the desired responseThereby achieving the effect of filtering interference signals. The self-adaptation is to calculate the actual output signalAnd the estimated error e (n) between the estimated error and the expected signal d (n), and the change of the error signal e (n) at each moment is tracked in the operation process of the filtering algorithm, and the adjustable parameters of the filter are adjusted correspondingly, so that the output signal is enabled to be perfect. The adaptive process can be realized by a steepest descent method and an adaptive gauss-newton method.

The cost function and the related function of the self-adaptive filter can guide the adjustment of the weight coefficient vector, and the output of the filter can meet the specific optimal design standard or make the target function converge to the minimum. In fact, the operation mode of the filter is determined by the cost function, and the step size of each adaptive process (updating the coefficients) affects the convergence rate and the steady-state performance. The relationship between the cost function and the error function is typically set to be non-linear in order to avoid the problem of limitations faced when the control variables are linearly related to the system error. The cost function will mainly follow two types of criteria: one class follows the criterion of maximizing the SNR of the output of the filter, and the other class is the RMSE (minimizing the mean square error) and its variants employed in this embodiment, and the two cost functions correspond to typical filters of matched filter and wiener filter, respectively. In this embodiment, the mean square error criterion is improved, and the cost function of the designed adaptive filtering algorithm (i.e., the HOC-LMS algorithm) is as follows:

how to locate inThe optimization filtering in the criterion is an optimization problem without constraints and is multidimensional, and the optimization problem is suitable for solving by a steepest descent method. The steepest descent method in the HOC-LMS algorithm is as followsThe negative gradient of (a) is used as a guide to search and continuously update the weight vector W (n) of the tap coefficient of the filter in the process, and the cost functionThe steepest descent direction is consistent with the gradient direction, and the weight vector is converged to the optimal wiener solution to satisfy the cost functionThe minimum criterion. Given that the third-order cumulant of the error signal e (n) is equal to the third-order moment (the 3 rd-order central moment is 0), and because the error signal (stationary random process) is consistent in terms of time and total third-order moment, the third-order moment can be expressed as the error signal at the current moment and the time interval tau₁And τ₂Is multiplied by the average value of the error signal.

The complex function is derived according to the chain rule to obtain the formula (4.2),the gradient expression of (a) is a polynomial composed of error signals e (n) at the current and past times, tap coefficient weight vectors w (n), and various kinds of self-accumulation amounts and mutual accumulation amounts.

Wherein, C₁、C₂、C₃、C₄、C₅、C₆、C₇Denotes a number from p (n), p (n-tau)₁)、p(n-τ₂)、y(n)、y(n-τ₁)、y(n-τ₂) And 3 random ones of the 7 kinds of cubic polynomial matrixes are arranged and combined. According to the analysis of the formula (4.2),the gradient expansion includes 7 cubic polynomial matrices (e.g. ). The result of the statistical averaging of each cubic polynomial in the time sense is its cubic cumulant, and can therefore be represented in a simplified formI.e. the gradient can representIs composed of three-order self-accumulation quantity and mutual-accumulation quantity parameters and can be obtained by calculation of formulas (4.3) and (4.4)The matrices are equal, and the composition parameters are third-order cumulants, namely M × M square matrix and 1 × M row matrix, so that the dimensions of the matrices are M and just fit the dimensions of the weight vector W (n).

Thus, it is possible to provideAn estimate of the gradient may be calculated from each of the cumulant parameters.

c_ppp(n,i,j),i、j＝[0,M-1]Representing the third order accumulated magnitude of any cubic polynomial in the matrix at time n.

c_ppy(n,0,j),j＝[0,M-1]Representing the third order accumulated magnitude of any cubic polynomial in the row matrix at time n.

Furthermore, the complexity of implementing the HOC-LMS algorithm by directly calculating the cumulant parameters is rather high. To ensure the engineering feasibility of the HOC-LMS algorithm, the third-order cumulants can be calculated by iteration To reduce computational complexity. The accumulated amount estimate through time n-1 is given by equation (4.5)Iteratively calculating cumulative magnitudes at n timesThe method of (3) can be realized only by calculating the multiplication operation of multiple signals at the current momentThe calculation complexity is simplified by the estimation of (2), and the calculation methods of the rest self-accumulation amount and the mutual accumulation amount can also be obtained by analogy and deduction in the mode.

When the HOC-LMS algorithm updates the tap weight coefficient each time, the step length plays a role of an adjustment quantity, so that the response speed of the adaptive filtering system is determined by the step length, and the response speed of the system also affects the convergence and stability of the algorithm, and the importance of the step value setting is visible. However, under the condition of a fixed step value, the convergence speed, the tracking speed and the final convergence precision cannot be obtained simultaneously, and the three have different requirements on the step length. This is mainly manifested as: when the step length is larger, the tap weight coefficient is updated each time, which is embodied in the concrete effect that the algorithm can reach convergence faster, but when the system is in a steady state, even if the gradient change is very small, the tap coefficient is greatly changed due to the overlarge step length, which is also called steady state maladjustment; the step length is small to avoid maladjustment in a steady state, but this also means that the parameter adjustment amplitude is small, and the step of dividing into two steps directly results in slow tracking and convergence speed, and in conclusion, the step value can only select the median of the two. In order to solve the contradiction that the mismatch, the fast tracking and the convergence are incompatible, the embodiment uses a nonlinear function to express the relationship between the step length mu (n) and the error e (n), the function can dynamically adjust the step length according to the change of the error, the output step length should be smaller when the error is smaller (steady state stage), and the output step length should be larger when the error is larger (convergence stage), so as to provide the optimal step length for each stage and even each output, thereby improving the convergence speed and the precision. The following formula happens to be a nonlinear function meeting the requirement, and the mathematical expression is:

μ(n)＝ψ·(exp(β·e(n))-θ) (4.6)

wherein ψ, β and θ are all variable constants, and can be adjusted at any time in practical simulation according to the tracking convergence performance of the HOC-LMS algorithm. The error signal e (n) and the step size μ (n) are the polar angle and the polar diameter, respectively, of the logarithmic spiral function. From the mathematical point of view, along with the increase of the polar angle, the more exponential increase is reflected on the polar diameter, and the error which exactly corresponds to the self-adaptive filtering system is larger at the moment, which shows that the step length is still in the convergence stage, and the step length is enlarged at the moment, so that the convergence speed is exactly accelerated. Conversely, the pole diameter is exponentially reduced along with the reduction of the pole angle, and the step length is reduced just corresponding to the situation that the self-adaptive filtering system tends to be stable, so that the maladjustment is avoided. The equation (4.6) ensures that the HOC-LMS algorithm can have faster convergence and tracking speed in the starting stage, and can keep the system stable in the steady-state stage due to smaller step length, thereby obviously improving the error convergence performance of the HOC-LMS algorithm.

The constellation diagram of NB-IoT signals added with gaussian white noise only is shown in fig. 10-1 (a scatter diagram drawn by using the scatterplot function directly), and the abscissa and ordinate tables in the diagram represent the in-phase component and the quadrature component of the data points.

Compared with fig. 10-1, after the Chirp interference signal is added, the constellation diagram becomes very disordered, and the constellation points originally concentrated around four points also become discrete. This indicates that the signal is strongly interfered and the signal distortion is severe.

After the data superimposed with the additive white gaussian noise and the Chirp interference signal is filtered by adopting the HOC-LMS algorithm, the constellation diagram is shown as fig. 10-3, and it is obvious that compared with fig. 10-2 and fig. 10-1, the filtered signal constellation diagram has no "convergence" of the constellation diagram of the NB-IoT signal only added with the white gaussian noise, but has obvious effect compared with the unprocessed signal. There is no algorithm that can achieve perfect filtering, and only approximation is performed until all information of the interfering signal is available, although this scenario does not exist in actual engineering. The intuitive embodiment in the constellation diagram is that the constellation points are not so dispersed, and the above is only based on the derivation of the constellation diagram, and the specific effect needs to be based on the bit error rate. The ratio of the number of error bits in the received data to the total number of bits is the system bit error rate, which is used as a parameter index for measuring the communication accuracy. The better the filtering effect of the HOC-LMS algorithm on the Chirp interference signal is, the lower the NB-IoT system error rate is.

The bit error rate curves of the NB-IoT system under four scenarios, namely, adding gaussian white noise only, superimposing Chirp interference with unprocessed gaussian white noise, and signal filtered by LMS and HOC-LMS (parameter configuration: filter order is 9, iteration times is 600) are shown in fig. 10-4. Wherein the abscissa and ordinate represent the signal-to-noise ratio and the bit error rate information of the NB-IoT system, respectively. From the analysis of FIGS. 10-4, the following conclusions can be drawn: firstly, the bit error rates of the signals of four different processing modes are the closest to each other only when the bit error rates are 0dB, and the bit error rates of the four processing modes are all as high as 10^-1The order of magnitude, starting from 0dB, the bit error rate performance is also improved along with the increase of the signal to noise ratio, and when the signal to noise ratio reaches 11dB, the bit error rate of the NB-IoT system after filtering processing by using the LMS algorithm is 10^-2To 10^-3The order of magnitude, the bit error rate of the system after filtering by using the HOC-LMS algorithm is 10^-3To 10^-4Order of magnitude, whereas the bit error rate of a system using only gaussian white noise as a reference is 10^-6To 10^-7Compared with the LMS algorithm, the HOC-LMS algorithm has an error gain of 0.5-3dB under the condition that the signal-to-noise ratio is 0-11dB (the bit error rate is the same, the signal-to-noise ratio difference of two scenes is the same), and the higher the signal-to-noise ratio is, the stronger the gain effect is; thirdly, the bit error rate of the system filtered by the LMS algorithm and the HOC-LMS algorithm has 0.5-3dB gain under the condition of 2-11dB, but the gain effect is almost not (approaches to 0) in the range of 0-2 dB. Analysis shows that when the HOC-LMS algorithm filters the Chirp signal, the performance of the HOC-LMS algorithm is influenced by the low signal-to-noise ratio, but from 2dB, the gain effect of the HOC-LMS algorithm is obvious along with the increase of the signal-to-noise ratioIt is also shown that the HOC-LMS algorithm has a good interference suppression effect, but requires a certain communication environment requirement to exert its performance. And fourthly, with the signal-to-noise ratio being larger, the interference suppression performance of the HOC-LMS algorithm is gradually increased and then tends to be flat, and compared with the NB-IoT system bit error rate curve only added with white Gaussian noise, the HOC-LMS algorithm performance curve cannot well approach the ideal bit error rate performance, but is still better than the LMS algorithm performance curve.

Compared with the graph shown in fig. 10-1, the constellation diagram is very disordered after the DME interference signal is added, and constellation points originally concentrated around four points become discrete. This indicates that the signal is strongly interfered and the signal distortion is severe. And compared with NB-IoT signals added with Chirp and white Gaussian noise, the interference effect of the DME signal is more serious. In contrast to fig. 10-2, the constellation shown in fig. 11-1 becomes more sparse with points and has a larger range of "spread", but there are no particularly discrete points, so the impact of DME signal on the bit error rate performance of NB-IoT systems is either greater than Chirp signal or needs a specific simulation diagram to demonstrate.

After the data superimposed with the additive white gaussian noise and the DME interference signal is filtered by using the HOC-LMS algorithm, the constellation diagram is shown in fig. 11-2, and it can be obviously seen by comparing with fig. 11-1 that the convergence effect of the constellation diagram is greatly improved, and the points in the constellation diagram shown in fig. 11-1 are almost distributed in the whole space and no longer show the concentration trend, just like uniform distribution. But after the processing of the HOC-LMS algorithm, the constellation becomes "converged" again. Compared with the filtering effect of the HOC-LMS algorithm on the Chirp interference signal and the DME interference signal, the filtering effect of the HOC-LMS algorithm on the DME interference signal is more obvious only from the constellation diagram. This assumption will be discussed in detail in the system bit error rate performance analysis.

Only gaussian white noise was added, unprocessed gaussian white noise was superimposed with DME interference, signals filtered with LMS and HOC-LMS algorithms respectively (parameter configuration:filter order is 9, iteration number is 600) the bit error rate curves of the NB-IoT system under four scenarios are shown in fig. 11-3, where the abscissa and ordinate respectively represent the signal-to-noise ratio and the bit error rate information of the NB-IoT system. From the analysis of FIGS. 11-3, the following conclusions can be drawn: firstly, the bit error rates of the signals of four different processing modes are the closest to each other only when the bit error rates are 0dB, and the bit error rates of the four processing modes are all as high as 10^-1The order of magnitude, starting from 0dB, the bit error rate performance is also improved along with the increase of the signal to noise ratio, and when the signal to noise ratio reaches 11dB, the bit error rate of the NB-IoT system subjected to filtering processing by the LMS algorithm is 10^-3The order of magnitude, the bit error rate of the system after filtering by using the HOC-LMS algorithm can reach 10^-5Order of magnitude, whereas the bit error rate of a system using only gaussian white noise as a reference is 10^-6To 10^-7Compared with unprocessed signals, the HOC-LMS algorithm can achieve the highest gain of 3.5dB within the interval of 0-11dB, and the gain effect is more obvious along with the increase of the signal-to-noise ratio; and thirdly, the filtering effect is different from that of a Chirp signal, the filtering effects of the LMS and the HOC-LMS algorithm are improved differently under the DME interference scene, and compared with the LMS algorithm, the bit error rate of the system filtered by the HOC-LMS algorithm has a gain of 0.5-3.5dB under the condition of 0-11dB, and obviously, the filtering effect of the HOC-LMS algorithm on the DME signal is better. At the moment, the HOC-LMS algorithm can also exert part of performance under the condition of low signal-to-noise ratio, and error gain blind areas do not exist. However, when the signal-to-noise ratio is within the range of 3-11dB, the error gain of the HOC-LMS algorithm also increases exponentially with the increase of the signal-to-noise ratio, which also indicates that the HOC-LMS algorithm has excellent performance, has good error gain even in a low-signal-to-noise ratio scenario, but can work better in a communication environment with a higher signal-to-noise ratio; and fourthly, with the signal-to-noise ratio being larger, the interference suppression performance of the HOC-LMS algorithm is improved gradually and then gradually, and compared with the NB-IoT system bit error rate curve filtered by the LMS algorithm, the HOC-LMS algorithm performance curve has a bottleneck but can better approach the ideal bit error rate performance. Moreover, compared with the filtering effect of the Chirp interference signal, the bit error rate curve after DME filtering not only drops faster, but also has two orders of magnitude difference when the SNR is 11dB, and the bit error rate curve is further improvedThe conclusion, which is deduced by comparing fig. 10-3 with fig. 11-2, is that the filtering effect of the HOC-LMS algorithm on the DME signal is better than that of the Chirp.

Compared with fig. 10-1, it can be obviously seen that after the GSM interference signal is added, the constellation diagram becomes very chaotic, and the constellation points originally concentrated around four points also become discrete. This indicates that the signal is strongly interfered and the signal distortion is severe. Although the main part of the constellation diagram is "converged", the deviation degree of the discrete points is quite high, and compared with fig. 10-2 and fig. 12-1, it is found that the GSM and Chirp signals have almost the same influence on the constellation diagram of the NB-IoT system, and based on the same inference, the GSM and Chirp signals can be assumed to have approximately the same influence on the bit error rate performance of the NB-IoT system, and will be verified through a bit error rate simulation diagram.

After the data superimposed with the additive white gaussian noise and the GSM interference signal is filtered by using the HOC-LMS algorithm, the constellation diagram is shown in fig. 12-2, and it is obvious that, although the main part of the constellation point is relatively dispersed, the point far away from the main part is limited in the interval where the point is located after filtering compared with fig. 12-1. In other words, the interference to the neighbourhood is reduced. Compared with the filtering effect of the HOC-LMS algorithm on the Chirp and DME interference signals, the filtering effect of the HOC-LMS algorithm on the DME interference signals is more obvious only from the view of the constellation diagram, and the width of a guard band (i.e. a region without constellation points in adjacent regions) between four regions of the constellation diagram after the Chirp and GSM signals are filtered is approximately equivalent. It can therefore be assumed that the results of filtering suppression for Chirp and GSM interference are comparable. This assumption will also be discussed in detail in the system bit error rate performance analysis.

The bit error rate curves of the NB-IoT system under four scenarios, namely adding gaussian white noise only, superimposing the GSM interference with unprocessed gaussian white noise, and filtering the signal by the LMS and HOC-LMS algorithms (parameter configuration: filter order is 9, iteration number is 600 times), are shown in fig. 12-3. Wherein the abscissa and ordinate represent the signal-to-noise ratio and the error ratio of the NB-IoT system, respectivelyAnd (4) the bitrate information. From the analysis of FIGS. 12-3, the following conclusions can be drawn: firstly, the bit error rates of the signals of four different processing modes are the closest to each other only when the bit error rates of the signals are 0dB, and the bit error rates of the signals of the four different processing modes are all as high as 10^-1The order of magnitude, starting from 0dB, the bit error rate performance is also improved along with the increase of the signal to noise ratio, when the signal to noise ratio reaches 11dB, the bit error rate of the NB-IoT system filtered by using the LMS algorithm is 10^-2To 10^-3The order of magnitude, the bit error rate of the system after filtering by using the HOC-LMS algorithm is 10^-3To 10^-4Order of magnitude, whereas the bit error rate of a system using only gaussian white noise as a reference is 10^-6To 10^-7An order of magnitude; compared with the LMS algorithm, the HOC-LMS algorithm has an error gain of 0-3dB under the condition that the signal-to-noise ratio is 0-11dB, and the gain effect is more obvious along with the increase of the signal-to-noise ratio; compared with the system bit error rate of the LMS algorithm, the HOC-LMS algorithm has the gain of 0.5-3dB under the condition of 2-11dB, but the gain effect is almost not (approaches to 0) within the range of 0-2 dB. The characteristic has the same effect as the Chirp interference filtering and accords with the result obtained by the comparison of the constellation diagrams. Analysis shows that the HOC-LMS algorithm cannot generate high gain under the condition of low signal-to-noise ratio when filtering the GSM signal, namely, is in the error gain blind zone. When the signal-to-noise ratio is within the range of 2-11dB, the error gain of the HOC-LMS algorithm increases exponentially along with the increase of the signal-to-noise ratio, which also indicates that the HOC-LMS algorithm has excellent performance but needs certain communication environment requirements to exert the performance; with the higher signal-to-noise ratio, the interference suppression performance of the HOC-LMS algorithm is improved gradually and gradually, and although the performance bottleneck still exists, the performance curve of the HOC-LMS algorithm can better approach the ideal bit error rate curve compared with the performance curve of the LMS algorithm and is slightly better than that of the Chirp algorithm. Comparing fig. 12-3 and fig. 10-4, it is found that the influence of the Chirp signal and the GSM signal on the bit error rate performance of the NB-IoT system when no filtering is performed is approximately the same, regardless of the LMS algorithm or the HOC-LMS algorithm, and the same is true for the bit error rate performance curves in the Chirp and GSM scenarios. According to the inference derived from both constellations.

Since the Chirp curve and the GSM curve are so close, the same conclusion can be drawn when comparing the GSM curve with the DME curve. The bit error rate curve after DME filtering is not only reduced more quickly, but also the bit error rates of the DME and the DME are different by two orders of magnitude when the SNR is 11dB, and the inference obtained by comparing the constellation diagrams of the DME and the DME is further proved, namely the filtering effect of the HOC-LMS algorithm on the DME signals is better than that of Chirp and GSM.

In combination with the simulation results and the third order cumulant simulation plots for NB-IoT and Chirp/DME/GSM signals, the following conclusions can be drawn:

the three interference signals are sequentially Chirp, GSM and DME according to the sequence from large to small of the influence on the bit error rate performance of the NB-IoT system, however, the difference among the three interference signals is very small, and the difference of the bit error rates after the three interference signals are filtered is mainly the difference of the applicability of the algorithm and the high-order statistic domains of the three interference signals.

Filtering effects of the HOC-LMS algorithm on three interference signals are different, filtering effects on Chirp interference signals and GSM interference signals are approximately same, and 10 can be achieved when SNR is 10dB^-3Magnitude and less than ideal effect in low signal-to-noise ratio. After DME interference signals are filtered, the bit error rate can reach 10 when SNR is 11dB^-5Magnitude, and the DME curve is closer to the ideal bit error rate curve than the Chirp and GSM curves. Actually, comparing the high-order cumulant of the three signals, it can be found that the difference between the DME signal and the NB-IoT signal in the high-order cumulant domain is larger than that between the two other signals, and the scheme adopted in this embodiment is to consider the NB-IoT signal as an interference signal together for suppression, and since the difference between the GSM/Chirp signal and the NB-IoT signal in the third-order cumulant domain is not obvious enough, the filtering effect is slightly lost.

To sum up, the filter construction method for narrowband internet of things interference suppression provided by the embodiment of the invention adopts optimized filter construction based on the difference of the NB-IoT signal and the Chirp/DME/GSM signal in the high-order cumulantCriterion accurately estimates Chirp/DME/GSM signals on a high-order domain without being influenced by NB-IoT (Gaussian colored noise) signalsThe Chirp/DME/GSM estimated value can be accurately subtracted from the mixed NB-IoT signal, the self-adaptive suppression of the Chirp/DME/GSM interference signal can be realized, the NB-IoT high-order statistical characteristic is fully represented, meanwhile, the optimal filtering in the third-order cumulant sense is realized, and experimental verification is obtained.

The above embodiments are preferred embodiments of the present invention, but the present invention is not limited to the above embodiments, and any other changes, modifications, substitutions, combinations, and simplifications which do not depart from the spirit and principle of the present invention should be construed as equivalents thereof, and all such changes, modifications, substitutions, combinations, and simplifications are intended to be included in the scope of the present invention.