阅读说明：本技术 高频脉振电流注入无轴承磁通切换电机转子偏心观测方法 (Method for observing eccentricity of rotor of high-frequency pulse vibration current injection bearingless flux switching motor ) 是由 周扬忠 陈垚 杨公德 屈艾文 钟天云 于 2021-06-29 设计创作，主要内容包括：本发明提出一种高频脉振电流注入无轴承磁通切换电机转子偏心观测方法,其在电机绕组中注入高频脉振电流,选取两套空间对称绕组中高频电流引起的高频电压差异,通过单位正弦函数信号与低通滤波器提取出高频电压差异的直流分量,再根据坐标变换将高频电压差异的直流分量分解到两相静止坐标系中,利用高频电压差异与转子径向位移的关系,观测出转子径向位移,将观测的转子径向位移作为负反馈引入电机悬浮控制回路中,可以实现电机在零速和低速情况下的稳定悬浮。本发明不需要径向位移传感器,通过电机本身的电流和电压信号即可实现转子在零速和低速情况下的悬浮控制,能够有效降低电机制造成本,提高控制系统可靠性。(The invention provides a method for observing eccentricity of a rotor of a bearingless flux switching motor by injecting high-frequency pulse vibration current into a motor winding, selecting high-frequency voltage difference caused by the high-frequency current in two sets of spatially symmetrical windings, extracting direct-current components of the high-frequency voltage difference through a unit sine function signal and a low-pass filter, decomposing the direct-current components of the high-frequency voltage difference into two-phase static coordinate systems according to coordinate transformation, observing radial displacement of the rotor by utilizing the relation between the high-frequency voltage difference and the radial displacement of the rotor, and introducing the observed radial displacement of the rotor into a suspension control loop of the motor as negative feedback, so that stable suspension of the motor under the conditions of zero speed and low speed can be realized. The invention can realize the suspension control of the rotor under the conditions of zero speed and low speed through the current and voltage signals of the motor without a radial displacement sensor, can effectively reduce the manufacturing cost of the motor and improve the reliability of a control system.)

1. A method for observing the eccentricity of a rotor of a bearingless flux switching motor by injecting high-frequency pulse vibration current is characterized by comprising the following steps of:

injecting high-frequency pulse vibration current into a motor winding, selecting high-frequency voltage difference caused by the high-frequency current in two sets of space symmetrical windings, extracting direct-current components of the high-frequency voltage difference through a unit sine function signal and a low-pass filter, decomposing the direct-current components of the high-frequency voltage difference into two-phase static coordinate systems according to coordinate transformation, observing radial displacement of a rotor by utilizing the relation between the high-frequency voltage difference and the radial displacement of the rotor, and introducing the observed radial displacement of the rotor into a motor suspension control loop as negative feedback to realize stable suspension of the motor under the conditions of zero speed and low speed.

2. The method for observing the eccentricity of the rotor of the high-frequency pulsating current injection bearingless flux switching motor according to claim 1, comprising the following steps of:

step S1: obtaining six-phase stator current i_A～i_FTime t, setting the frequency ω of the injected high frequency current signal_hCalculating the angle theta of the high-frequency signal_h：

θ_h＝ω_h·t；

Step S2: using the high frequency signal angle θ obtained in step S1_hCalculating the high-frequency pulse vibration current:

wherein, I_mIs the amplitude of the injected high-frequency current signal;

step S3: using the alpha T axis high frequency pulse vibration current signal i obtained in step S2_αThAnd beta T axis high frequency pulse vibration current signal i_βThGiven i according to the torque plane fundamental current_αTf ^*And i_βTf ^*Calculating the torque plane current given i_αT ^*And i_βT ^*：

Step S4: given i by the torque plane current obtained in step S3_αT ^*And i_βT ^*Given i according to the levitated plane current_αS ^*And i_βS ^*Zero sequence current given i_o1 ^*And i_o2 ^*Calculating six-phase stator current given i_A ^*～i_F ^*：

Wherein i_o1 ^*＝0，i_o2 ^*＝0；

Step S5: the six-phase current obtained in step S4 is given by i_A ^*～i_F ^*And the six-phase stator current i obtained in step S1_A～i_FObtaining inverter switching signal S through current closed-loop controller_A～S_FThereby realizing the injection of a high-frequency current signal i into the motor winding when the motor runs_αThAnd i_βTh；

Step S6: detecting E, B, C, F phase winding terminal voltage u_E、u_B、u_C、u_F；

Step S7: the sum u of the high-frequency voltages of the EB phase of the space symmetrical winding is obtained by an adder_EBSum u of CF-phase high-frequency voltages_CFPassing through center frequency of omega_hThe band-pass filter obtains the angular frequency of omega_hThe sum u of high-frequency voltages of EB phases of the space-symmetrical winding_EBhSum u of CF-phase high-frequency voltages_CFh：

Wherein BPF (-) denotes a band-pass filter;

step S8: using the sum u of the high-frequency voltages of the EB phases of the space symmetrical winding obtained in the step S7_EBhSum u of CF-phase high-frequency voltages_CFhMultiplying by a unit sine signal s-cos (ω)_ht) obtaining a double frequency component u of the sum of the high-frequency voltages of EB phases of the space symmetrical winding_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFh：

Step S9: using the frequency component u twice of the sum of the high-frequency voltages of the EB phases of the space symmetrical winding obtained in the step S8_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFhRespectively pass through a cut-off frequency of 0.2 omega_hThe low-pass filter obtains a direct-current component u of the sum of high-frequency voltages of EB phases of the space symmetrical winding_LEBhDC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFh：

Wherein LPF (·) represents a low-pass filter;

step S10: utilizing the DC component u of the sum of the high-frequency voltages of the EB phases of the space symmetrical winding obtained in the step S9_LEBhA DC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFhDecomposing the alpha-axis high-frequency voltage component into a two-phase static coordinate system to obtain an alpha-axis high-frequency voltage component u_LαhAnd beta axis high frequency voltage component u_Lβh：

Step S11: using the α -axis high-frequency voltage component u obtained in step S10_LαhAnd beta axis high frequency voltage component u_LβhAnd estimating the radial displacement x and y of the rotor as follows:

wherein the content of the first and second substances,

where M is the offset self-inductance and a and b are the inductance.

3. The method for observing the eccentricity of the rotor of the high-frequency pulse-width-modulated current injection bearingless flux switching motor according to claim 2, wherein:

in step S3, the torque plane fundamental current is given by i_αTf ^*And i_βTf ^*From a given speed n^*And obtaining the error between the current and the actual rotating speed n and the phase winding current through a given calculation link of the torque plane fundamental wave current according to a torque control algorithm.

4. The method for observing the eccentricity of the rotor of the high-frequency pulse-width-modulated current injection bearingless flux switching motor according to claim 2, wherein:

in step S4, the floating plane current is given by i_αS ^*And i_βS ^*By a given rotor eccentricity x^*Error from observed rotor eccentricity x, given rotor eccentricity y^*And observing the error between the rotor eccentricity y, and obtaining the error through a suspension plane current setting calculation link according to a suspension force control algorithm.

5. The method for observing the eccentricity of the rotor of the high-frequency pulse-width-modulated current injection bearingless flux switching motor according to claim 2, wherein:

in step S1, a stator winding current i is detected using a current sensor and an AD conversion channel_A～i_F。

6. The method for observing the eccentricity of the rotor of the high-frequency pulse-width-modulated current injection bearingless flux switching motor according to claim 2, wherein:

in step S11, the deviation self-inductance M and the inductance coefficients a and b are obtained by finite element simulation calculation, which specifically includes the following steps:

step S11-1: unit forward current is introduced into the A-phase winding, and self inductance L of the A-phase winding under the condition that the rotor is not eccentric is calculated₀Then, the rotor is set to be eccentric by 0.1mm in the direction phi which is equal to 45 degrees, and the obtained deviation self-inductance M is as follows:

M＝(L_e-L₀)×10⁴

wherein L is_eThe self inductance of the A-phase winding is obtained when the rotor is eccentric 0.1mm in the phi-45-degree direction;

step S11-2: the unit forward current is led into the A-phase winding, and the flux linkage coupled by the A1 coil is l_A1The flux linkage of the coil coupling of A2 is l_A2And l_A1＝l_A2，l_A1+l_A2＝L₀(ii) a The flux linkage of the coil coupling with a 30 ° difference from the a1 coil is l_aThe flux linkage of the coil coupling with a 60 ° difference from the a1 coil is l_bThe flux linkage of the coil coupling which differs by 90 ° from the a1 coil is l_cAnd calculating inductance coefficients a, b and c as follows:

7. a computer device comprising a memory, a processor, and a computer program stored on the memory and executable on the processor, wherein: the processor when executing the computer program realizes the steps adopted by the method for observing the eccentricity of the rotor of the bearingless flux switching motor by injecting the high-frequency pulsating current as claimed in any one of claims 1 to 6.

Technical Field

The invention belongs to the technical field of control of a bearingless flux switching motor, and particularly relates to a method for observing the eccentricity of a rotor of a bearingless flux switching motor by injecting high-frequency pulse vibration current.

Background

The permanent magnet of the stator permanent magnet type flux switching motor is only placed on the side of the stator, the rotor is only formed by stacking silicon steel sheets, and the permanent magnet and a winding are omitted, so that the demagnetization risk of the permanent magnet caused by temperature rise can be effectively avoided, and the permanent magnet type flux switching motor has the advantages of high torque density, high working efficiency, strong rotor operation robustness, suitability for high-speed operation and the like. However, because the motor rotor is supported by a mechanical bearing, the improvement of the rotating speed of the rotor is limited by mechanical friction, and the problems of pollution and the like caused by bearing lubrication are solved.

In order to overcome the disadvantages of mechanical bearing support, a bearingless technology is introduced into the motor, thereby forming a bearingless flux switching motor. In order to realize the radial suspension of the rotor, the stator current generated by the stator winding is needed to modulate the air gap magnetic field between the stator and the rotor so as to break the balance magnetic field in the air gap, thereby generating the suspension force meeting the radial suspension of the rotor. In order to generate the rotor suspension force, the original three-phase winding is split into the symmetrical six-phase winding by means of the original three-phase winding coil, and the suspension force meeting the suspension requirement of the rotor is generated by utilizing the suspension current components flowing in the same direction in the space symmetrical winding, so that the six-phase single-winding bearingless magnetic flux switching motor is formed. The method is beneficial to the full play of the output torque capacity of the motor.

In the operation process of a bearingless flux switching motor, accurate detection of radial displacement is a key link for stable operation of the motor, a mechanical sensor is usually arranged at the end of the motor to obtain signals of tangential position and radial displacement of a rotor, and the two signals are used for controlling rotation and suspension of the motor. However, the installation of the mechanical sensor in the motor hinders the reliable operation and the integration development of the motor, limits the improvement of the critical rotating speed, and increases the manufacturing cost of the system.

Disclosure of Invention

Aiming at the defects and shortcomings in the prior art, the invention provides a method for observing the rotor eccentricity of a high-frequency pulse vibration current injection bearingless flux switching motor, and aims to solve the problems of estimation and control of the radial displacement of the rotor of the motor under the condition of no rotor radial displacement sensor. High-frequency pulse vibration current is injected into a motor winding, high-frequency voltage difference caused by the high-frequency current in two sets of spatially symmetrical windings is selected, direct-current components of the high-frequency voltage difference are extracted through a unit sine function signal and a low-pass filter, the direct-current components of the high-frequency voltage difference are decomposed into two-phase static coordinate systems according to coordinate transformation, radial displacement of a rotor is observed by utilizing the relation between the high-frequency voltage difference and the radial displacement of the rotor, the observed radial displacement of the rotor is introduced into a motor suspension control loop as negative feedback, and stable suspension of the motor under the conditions of zero speed and low speed can be realized. The invention can realize the suspension control of the rotor under the conditions of zero speed and low speed through the current and voltage signals of the motor without a radial displacement sensor, can effectively reduce the manufacturing cost of the motor and improve the reliability of a control system.

The invention specifically adopts the following technical scheme:

a method for observing the eccentricity of a rotor of a bearingless flux switching motor by injecting high-frequency pulse vibration current is characterized by comprising the following steps of:

injecting high-frequency pulse vibration current into a motor winding, selecting high-frequency voltage difference caused by the high-frequency current in two sets of space symmetrical windings, extracting direct-current components of the high-frequency voltage difference through a unit sine function signal and a low-pass filter, decomposing the direct-current components of the high-frequency voltage difference into two-phase static coordinate systems according to coordinate transformation, observing radial displacement of a rotor by utilizing the relation between the high-frequency voltage difference and the radial displacement of the rotor, and introducing the observed radial displacement of the rotor into a motor suspension control loop as negative feedback to realize stable suspension of the motor under the conditions of zero speed and low speed.

Further, the method comprises the following steps:

step S1: obtaining six-phase stator current i_A～i_FTime t, setting the frequency ω of the injected high frequency current signal_hCalculating the angle theta of the high-frequency signal_h：

θ_h＝ω_h·t；

Step S2: using the high frequency signal angle θ obtained in step S1_hCalculating the high-frequency pulse vibration current:

wherein, I_mIs the amplitude of the injected high-frequency current signal;

step S3: using the alpha T axis high frequency pulse vibration current signal i obtained in step S2_αThAnd beta T axis high frequency pulse vibration current signal i_βThGiven according to the torque plane fundamental currentAndcalculating torque plane current giveAnd

step S4: torque plane current setting obtained in step S3Andaccording to the current given by the suspended planeAndzero sequence current settingAndcalculating six-phase stator current give

Wherein i_o1 ^*＝0，i_o2 ^*＝0；

Step S5: the six-phase current obtained in step S4 is givenAnd the six-phase stator current i obtained in step S1_A～i_FObtaining inverter switching signal S through current closed-loop controller_A～S_FThereby realizing the injection of a high-frequency current signal i into the motor winding when the motor runs_αThAnd i_βTh；

Step S6: detecting E, B, C, F phase winding terminal voltage u_E、u_B、u_C、u_F；

Step S7: the sum u of the high-frequency voltages of the EB phase of the space symmetrical winding is obtained by an adder_EBSum u of CF-phase high-frequency voltages_CFThen pass throughCenter frequency of omega_hThe band-pass filter obtains the angular frequency of omega_hThe sum u of high-frequency voltages of EB phases of the space-symmetrical winding_EBhSum u of CF-phase high-frequency voltages_CFh：

Wherein BPF (-) denotes a band-pass filter;

step S8: using the sum u of the high-frequency voltages of the EB phases of the space symmetrical winding obtained in the step S7_EBhSum u of CF-phase high-frequency voltages_CFhMultiplying by a unit sine signal s-cos (ω)_ht) obtaining a double frequency component u of the sum of the high-frequency voltages of EB phases of the space symmetrical winding_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFh：

Step S9: using the frequency component u twice of the sum of the high-frequency voltages of the EB phases of the space symmetrical winding obtained in the step S8_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFhRespectively pass through a cut-off frequency of 0.2 omega_hThe low-pass filter obtains a direct-current component u of the sum of high-frequency voltages of EB phases of the space symmetrical winding_LEBhDC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFh：

Wherein LPF (·) represents a low-pass filter;

step S10: utilizing the DC component u of the sum of the high-frequency voltages of the EB phases of the space symmetrical winding obtained in the step S9_LEBhA DC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFhDecomposing the alpha-axis high-frequency voltage component into a two-phase static coordinate system to obtain an alpha-axis high-frequency voltage component u_LαhAnd beta axis high frequency voltage component u_Lβh：

Step S11: using the α -axis high-frequency voltage component u obtained in step S10_LαhAnd beta axis high frequency voltage component u_LβhAnd estimating the radial displacement x and y of the rotor as follows:

wherein the content of the first and second substances,

where M is the offset self-inductance and a and b are the inductance.

Further, in step S3, the torque plane fundamental current is givenAndfrom a given speed n^*And obtaining the error between the current and the actual rotating speed n and the phase winding current through a given calculation link of the torque plane fundamental wave current according to a torque control algorithm.

Further, in step S4, the floating plane current is givenAndby a given rotor eccentricity x^*Error from observed rotor eccentricity x, given rotor eccentricity y^*And observing the error between the rotor eccentricity y, and obtaining the error through a suspension plane current setting calculation link according to a suspension force control algorithm.

Further, the method can be used for preparing a novel materialIn step S1, the stator winding current i is detected using the current sensor and the AD conversion channel_A～i_F。

Further, in step S11, the deviation self-inductance M and the inductance coefficients a and b are obtained by finite element simulation calculation, which specifically includes the following steps:

step S11-1: unit forward current is introduced into the A-phase winding, and self inductance L of the A-phase winding under the condition that the rotor is not eccentric is calculated₀Then, the rotor is set to be eccentric by 0.1mm in the direction phi which is equal to 45 degrees, and the obtained deviation self-inductance M is as follows:

M＝(L_e-L₀)×10⁴

wherein L is_eThe self inductance of the A-phase winding is obtained when the rotor is eccentric 0.1mm in the phi-45-degree direction;

step S11-2: the unit forward current is led into the A-phase winding, and the flux linkage coupled by the A1 coil is l_A1The flux linkage of the coil coupling of A2 is l_A2And l_A1＝l_A2，l_A1+l_A2＝L₀(ii) a The flux linkage of the coil coupling with a 30 ° difference from the a1 coil is l_aThe flux linkage of the coil coupling with a 60 ° difference from the a1 coil is l_bThe flux linkage of the coil coupling which differs by 90 ° from the a1 coil is l_cAnd calculating inductance coefficients a, b and c as follows:

and, a computer device comprising a memory, a processor, and a computer program stored on the memory and executable on the processor, characterized in that: when the processor executes a computer program, the steps adopted by the method for observing the eccentricity of the rotor of the bearingless flux switching motor by injecting the high-frequency pulse vibration current are realized.

Compared with the prior art, the invention and the preferred scheme thereof have the following beneficial effects:

1. according to the method, the estimated rotor radial displacement is adopted to replace a sampling value obtained by a radial displacement sensor, so that the manufacturing and control operation cost of the bearingless motor is reduced, the reliability of a control system is improved, and the integration level of a motor system is improved;

2. the invention does not adopt an axial support frame and a reference ring, can shorten the axial length of the motor, reduce the weight of the motor and simplify the structure of the motor;

3. the invention adopts high-frequency current injection which is far higher than the frequency of torque and suspension control current, and can effectively and accurately estimate the radial displacement of the rotor under the working conditions of zero speed and low rotating speed of the motor;

4. according to the invention, the high-frequency pulse vibration current signal is injected into the torque plane static coordinate system, so that the adverse effect of the rotor displacement estimation process on the rotor suspension control is avoided.

Drawings

The invention is described in further detail below with reference to the following figures and detailed description:

FIG. 1 is a schematic diagram of the method of the embodiment of the present invention.

Fig. 2 is a schematic cross-sectional view of a six-phase single-winding bearingless flux switching motor according to an embodiment of the present invention.

Fig. 3 is a schematic diagram of a hardware structure of a driving system according to an embodiment of the present invention.

FIG. 4 is a schematic diagram of a torque control coordinate system definition according to an embodiment of the present invention.

Fig. 5 is a schematic diagram illustrating a suspension control coordinate system definition according to an embodiment of the invention.

Fig. 6 is a schematic diagram illustrating inductance definition according to an embodiment of the invention.

Detailed Description

In order to make the features and advantages of the present invention comprehensible, embodiments accompanied with figures are described in detail as follows:

it should be noted that the following detailed description is exemplary and is intended to provide further explanation of the disclosure. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs.

It is noted that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of example embodiments according to the present application. As used herein, the singular forms "a", "an" and "the" are intended to include the plural forms as well, and it should be understood that when the terms "comprises" and/or "comprising" are used in this specification, they specify the presence of stated features, steps, operations, devices, components, and/or combinations thereof, unless the context clearly indicates otherwise.

As shown in fig. 1, the present embodiment provides a method for observing eccentricity of a rotor of a bearingless flux switching motor by injecting high-frequency pulsating current, which specifically includes the following steps:

step S1: obtaining six-phase stator current i_A～i_FTime t, setting the frequency ω of the injected high frequency current signal_hCalculating the angle theta of the high-frequency signal_h：

θ_h＝ω_h·t

Step S2: using the high frequency signal angle θ obtained in step S1_hCalculating the high-frequency pulse vibration current:

wherein, I_mIs the amplitude of the injected high frequency current signal.

Step S3: using the alpha T axis high frequency pulse vibration current signal i obtained in step S2_αThAnd beta T axis high frequency pulse vibration current signal i_βThTorque plane fundamental current givenAndcalculating torque plane current give And

step S4: torque plane current setting obtained in step S3Andsuspended plane current settingAndzero sequence current given i_o1 ^*And i_o2 ^*Calculating six-phase stator current set

Wherein i_o1 ^*＝0，i_o2 ^*＝0。

Step S5: the six-phase current obtained in step S4 is givenWith six-phase stator current i_A～i_FObtaining inverter switching signal S through current closed-loop controller_A～S_FThereby realizing the injection of a high-frequency current signal i into the motor winding when the motor runs_αThAnd i_βTh. Wherein the current closed-loop controller can be formed by using a known hysteresis comparator or a PID controller.

Step S6: detecting E, B, C, F phase winding terminal voltage u_E、u_B、u_C、u_F。

Step S7: the sum u of the high-frequency voltages of the EB phase of the space symmetrical winding is obtained by an adder_EBSum u of CF-phase high-frequency voltages_CFPassing through center frequency of omega_hThe band-pass filter obtains the angular frequency of omega_hThe sum u of high-frequency voltages of EB phases of the space-symmetrical winding_EBhSum u of CF-phase high-frequency voltages_CFh：

Wherein BPF (-) denotes a band-pass filter.

Step S8: using the sum u of the high-frequency voltages of the EB phases of the space symmetrical winding obtained in the step S7_EBhSum u of CF-phase high-frequency voltages_CFhMultiplying by a unit sine signal s-cos (ω)_ht) obtaining a double frequency component u of the sum of the high-frequency voltages of EB phases of the space symmetrical winding_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFh：

Step S9: using the frequency component u twice of the sum of the high-frequency voltages of the EB phases of the space symmetrical winding obtained in the step S8_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFhRespectively pass through a cut-off frequency of 0.2 omega_hThe low-pass filter obtains a direct-current component u of the sum of high-frequency voltages of EB phases of the space symmetrical winding_LEBhDC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFh：

Where LPF (·) denotes a low-pass filter.

Step S10: DC using the sum of the spatially symmetrical EB phase high-frequency voltages obtained in step S9Component u_LEBhDC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFhDecomposing the alpha-axis high-frequency voltage component into a two-phase static coordinate system to obtain an alpha-axis high-frequency voltage component u_LαhAnd beta axis high frequency voltage component u_Lβh：

Step S11: alpha-axis high-frequency voltage component u obtained by S10_LαhAnd beta axis high frequency voltage component u_LβhThe rotor radial displacement x and y can be estimated as:

wherein the content of the first and second substances,

where M is the offset self-inductance and a and b are the inductance.

In step S3, the torque plane fundamental wave current is givenAndcan be given a rotation speed n^*And obtaining the error between the current and the actual rotating speed n and the phase winding current through a given calculation link of the torque plane fundamental wave current according to a torque control algorithm.

In step S4, the floating plane current is givenAndcan be eccentric by a given rotor x^*And observed rotor eccentricity xError of (2), given rotor eccentricity y^*And observing the error between the rotor eccentricity y, and obtaining the error through a suspension plane current setting calculation link according to a suspension force control algorithm.

In step S11, the self-inductance M and the inductance coefficients a and b may be calculated by finite element simulation, which specifically includes the following steps:

step S11-1: unit forward current is introduced into the A-phase winding, and self inductance L of the A-phase winding under the condition that the rotor is not eccentric is calculated₀Then, the rotor is set to be eccentric by 0.1mm in the direction phi which is equal to 45 degrees, and the obtained deviation self-inductance M is as follows:

M＝(L_e-L₀)×10⁴

wherein L is_eThe self inductance of the A-phase winding is obtained when the rotor is eccentric 0.1mm in the phi-45 degree direction.

Step S11-2: the unit forward current is led into the A-phase winding, and the flux linkage coupled by the A1 coil is l_A1The flux linkage of the coil coupling of A2 is l_A2And l_A1＝l_A2，l_A1+l_A2＝L₀. The flux linkage of the coil coupling with a 30 ° difference from the a1 coil is l_aThe flux linkage of the coil coupling with a 60 ° difference from the a1 coil is l_bThe flux linkage of the coil coupling which differs by 90 ° from the a1 coil is l_cAnd calculating inductance coefficients a, b and c as follows:

preferably, in the present embodiment, in step S1, the stator winding current i may be detected by using a current sensor and an AD conversion channel_A～i_F；

The principle of the method of the present embodiment will be described in detail with reference to fig. 2 to 6.

Fig. 2 shows the structure of the motor in this embodiment, the motor has 12U-shaped iron cores, between each of which a permanent magnet magnetized in the tangential direction is sandwiched, the magnetizing directions are alternately opposite, and the rotor has 10 teeth. Each phase of winding of the stator is wound on the stator teeth which are vertical to each other in space in series to form 6 symmetrical windingsAnd (4) grouping. The winding space of the A phase and the D phase is symmetrical, the winding space of the B phase and the E phase is symmetrical, and the winding space of the C phase and the F phase is symmetrical. Six-phase windings with axial lines spatially different from each other by 60 degrees in mechanical angle, and six-phase symmetric torque current i for controlling the tangential rotation of the motor simultaneously flows through the windings_AT～i_FTAnd six-phase symmetric suspension current i for controlling radial suspension of rotor_AS～i_FSMeanwhile, in order to ensure that the suspension force generated by the motor rotor is in direct proportion to the suspension current, the magnitude and the direction of the suspension current flowing in the space symmetrical winding are equal, namely i_A＝i_AT+i_AS，i_B＝i_BT+i_BS，i_C＝i_CT+i_CS，i_D＝i_DT+i_DS，i_E＝i_ET+i_ES，i_F＝i_FT+i_FS(ii) a Wherein i_AT＝-i_DT，i_ET＝-i_BT，i_CT＝-i_FT，i_AS＝i_DS，i_ES＝i_BS，i_CS＝i_FS. An XY coordinate system is defined in which the X axis coincides with the a1 coil axis.

Preferably, the hardware structure of the driving system of the present embodiment is shown in fig. 3. The method comprises the following steps: the device comprises a rectifying circuit, a filter capacitor, a six-phase inverter, a bearingless flux switching motor, a six-phase winding current acquisition circuit, a six-phase winding voltage sampling circuit, an isolation drive, a central controller, a human-computer interface and the like. Wherein the six-phase inverter dc bus voltage may also be provided using a suitable dc power supply. The power tube in the six-phase inverter adopts IGBT or MOSFET, and the central controller adopts DSP or singlechip. The winding current acquisition circuit is formed by combining a Hall current sensor and an operational amplifier, and can also be formed by combining a winding series power resistor and a differential operational amplifier. The Hall scheme can effectively realize the electrical isolation of the control loop and the main loop, and the winding series power resistance scheme can reduce the cost of the driving system. The six-phase winding voltage sampling circuit is formed by combining a Hall voltage sensor and an operational amplifier, and can also be formed by combining a voltage follower formed by an operational amplifier after voltage division of a parallel resistor. And weak current signals output by the current detection and terminal voltage sampling circuit are transmitted to the A/D conversion module of the central controller. According to the obtained signals and the rotor radial deviation observation method, the rotor radial deviation x and y are observed, then according to the observed rotor radial displacement and the stator current, a control signal to be sent is calculated by a rotor radial suspension and tangential rotation control strategy, and the switching action of a power switching tube in the inverter is controlled through isolation driving.

The basic principle of the method of the embodiment is as follows:

the following torque control and levitation control coordinate systems are defined. As defined for the torque control coordinate system in fig. 4, the a-phase and D-phase windings are spatially symmetric, the B-phase and E-phase windings are spatially symmetric, and the C-phase and F-phase windings are spatially symmetric. The axes of the six-phase windings are spatially separated by a mechanical angle of 60 deg.. Six-phase symmetric torque current i for controlling the tangential rotation of the motor flows in the winding at the same time_AT～i_FTWherein i_AT＝-i_DT，i_ET＝-i_BT，i_CT＝-i_FT. Changing stator current of a natural coordinate system A-F of the motor to a static rectangular coordinate system alpha T beta T by using a constant power matrix, wherein the projection of the torque current in the alpha T beta T coordinate system is i_αT、i_βTThen, the torque current is changed from a static rectangular coordinate system alpha T beta T to a dq rotating coordinate system, and the projection i of the torque current in a dTqT coordinate system_dTAnd i_qT. Wherein, T₆The constant power matrix is:

make the suspension current i flowing in the space symmetrical winding_AS～i_FSAll directions of magnitude being equal, i.e. i_AS＝i_DS，i_ES＝i_BS，i_CS＝i_FS. As can be seen from FIG. 1, the A-phase winding axis is located at a mechanical angle that the centerline of the rotor tooth leads the A1 coil by 9 degrees counterclockwise, and the rotor is subjected to a levitation force in a spatial direction of approximately 45 degrees according to xy-direction magnetic tension analysis, so that a levitation control coordinate system definition is established, as shown in FIG. 5. XY is a horizontal-vertical right angleA coordinate system, wherein the X axis is coincident with the axial direction of the A1 coil in the figure 1 and is different from the axial direction of the A phase winding by 9 degrees of mechanical angle, is defined similarly to the torque control coordinate system, a static rectangular coordinate system alpha S beta S is established, and the projection of the suspension current in the alpha S beta S coordinate system is i_αS、i_βS。

As shown in FIG. 6, the inductance is defined schematically, a unit forward current is applied to the A-phase winding, and the self-inductance L of the A-phase winding under the condition of no eccentricity of the rotor is calculated₀Then the rotor is set to be eccentric 0.1mm in the direction phi which is equal to 45 degrees,

the offset self-inductance M is obtained as:

M＝(L_e-L₀)×10⁴ (2)

wherein L is_eThe self inductance of the A-phase winding is obtained when the rotor is eccentric 0.1mm in the phi-45 degree direction.

The unit forward current is led into the A-phase winding, and the flux linkage coupled by the A1 coil is l_A1The flux linkage of the coil coupling of A2 is l_A2And l_A1＝l_A2，l_A1+l_A2＝L₀. The flux linkage of the coil coupling with a 30 ° difference from the a1 coil is l_a，

The flux linkage of the coil coupling with a 60 ° difference from the a1 coil is l_bThe flux linkage of the coil coupling which differs by 90 from the A1 coil is l_cAnd calculating inductance coefficients a, b and c as follows:

setting the frequency omega of the injected high-frequency current signal_hObtaining six-phase stator current i_A～i_FTime t, calculated high

Frequency signal angle theta_h：

θ_h＝ω_h·t (4)

By high frequency signal angle theta_hCalculating the high-frequency pulse vibration current:

wherein, I_mIs the amplitude of the injected high frequency current signal.

The six-phase high-frequency current injected into the motor winding is given by i_Ah～i_Fh：

The angular frequency generated by the E-phase winding is omega_hThe high-frequency voltage of (d) is:

wherein M is_kEK is A-D, F is A-D, mutual inductance between the F phase winding and the E phase winding, L_EThe self inductance of the E-phase winding is obtained.

The angular frequency generated by the B-phase winding is omega_hThe high-frequency voltage of (d) is:

wherein M is_kBK is A, C-F are mutual inductance between the windings of the phases A, B-F and B, L_BThe phase B winding is self-inductance.

The angular frequency generated by the C-phase winding is omega_hThe high-frequency voltage of (d) is:

wherein M is_kCK is mutual inductance between A, B, D-F phase windings and C phase windings, L_CIs the self-inductance of the C-phase winding.

Angular frequency omega generated by the F-phase winding_hThe high-frequency voltage of (d) is:

wherein M is_kFK is mutual inductance between the A-E phase winding and the F phase winding, L_FIs the self-inductance of the F-phase winding.

Wherein the content of the first and second substances,

according to the formulas (7) to (11), the sum u of high-frequency voltages of EB phases of the space-symmetrical winding_EBhSum u of CF-phase high-frequency voltages_CFh：

Wherein the content of the first and second substances,

wherein M is deviation self-inductance, e is rotor eccentric distance, and phi is rotor eccentric angle.

The sum u of EB phase high-frequency voltages of the space symmetrical winding_EBhSum u of CF-phase high-frequency voltages_CFhMultiplying the unit sine signal s to obtain a frequency component u of the sum of the high-frequency voltages of the EB phases of the space symmetrical winding_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFh：

Doubling frequency component u of the sum of space symmetrical winding EB phase high-frequency voltage_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFhRespectively obtaining the DC components u of the sum of the high-frequency voltages of EB phases of the space symmetrical winding through a low-pass filter_LEBhDC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFh：

Doubling frequency component u of the sum of space symmetrical winding EB phase high-frequency voltage_eEBhDouble frequency component u of the sum of the CF-phase high-frequency voltages_eCFhRespectively obtaining the DC components u of the sum of the high-frequency voltages of EB phases of the space symmetrical winding through a low-pass filter_LEBhDC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFh：

Direct current component u of the sum of spatially symmetrical winding EB phase high-frequency voltages_LEBhDC component u of the sum of the high-frequency voltages of the CF phases of the spatially symmetrical winding_LCFhDecomposing the alpha-axis high-frequency voltage component into a two-phase static coordinate system to obtain an alpha-axis high-frequency voltage component u_LαhAnd beta axis high frequency voltage component u_LβhThe α -axis high-frequency voltage component u can be obtained by expressing the rotor eccentricity distance e and the rotor eccentricity angle Φ as x ═ e · cos Φ and y ═ e · sin Φ from the displacement of the rotor in the x direction and the displacement of the rotor in the y direction_LαhAnd beta axis high frequency voltage component u_LβhThe relationship to x and y is:

the rotor radial displacement x and y can be estimated by the formula (17):

therefore, the method for observing the eccentricity of the rotor of the bearingless flux switching motor by injecting the high-frequency pulse vibration current into the torque plane static coordinate system and collecting the spatially symmetrical phase E, phase B, phase C and phase F voltages can be obtained.

In summary, in order to obtain the estimated rotor radial displacement to replace the sampling value obtained by the radial displacement sensor, the embodiment adopts the high-frequency current far higher than the frequencies of the torque and the suspension control current, injects the high-frequency pulse vibration current signal into the torque plane static coordinate system, and can effectively and accurately estimate the rotor radial displacement under the working conditions of zero speed and low rotating speed of the motor through the relation between the rotor radial displacement and the high-frequency voltage difference of the space symmetric winding, thereby reducing the manufacturing and control operation cost of the bearingless motor, improving the reliability of the control system, and improving the integration level of the motor system.

As will be appreciated by one skilled in the art, embodiments of the present application may be provided as a method, system, or computer program product. Accordingly, the present application may take the form of an entirely hardware embodiment, an entirely software embodiment or an embodiment combining software and hardware aspects. Furthermore, the present application may take the form of a computer program product embodied on one or more computer-usable storage media (including, but not limited to, disk storage, CD-ROM, optical storage, and the like) having computer-usable program code embodied therein.

The present application is described with reference to flowchart illustrations and/or block diagrams of methods, apparatus (systems), and computer program products according to embodiments of the application. It will be understood that each flow and/or block of the flow diagrams and/or block diagrams, and combinations of flows and/or blocks in the flow diagrams and/or block diagrams, can be implemented by computer program instructions. These computer program instructions may be provided to a processor of a general purpose computer, special purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, create means for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.

These computer program instructions may also be stored in a computer-readable memory that can direct a computer or other programmable data processing apparatus to function in a particular manner, such that the instructions stored in the computer-readable memory produce an article of manufacture including instruction means which implement the function specified in the flowchart flow or flows and/or block diagram block or blocks.

These computer program instructions may also be loaded onto a computer or other programmable data processing apparatus to cause a series of operational steps to be performed on the computer or other programmable apparatus to produce a computer implemented process such that the instructions which execute on the computer or other programmable apparatus provide steps for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.

The foregoing is directed to preferred embodiments of the present invention, other and further embodiments of the invention may be devised without departing from the basic scope thereof, and the scope thereof is determined by the claims that follow. However, any simple modification, equivalent change and modification of the above embodiments according to the technical essence of the present invention are within the protection scope of the technical solution of the present invention.

The present invention is not limited to the above-mentioned preferred embodiments, and any other various methods for observing the eccentricity of the rotor of the bearingless flux switching motor by injecting the high-frequency pulsating current into the rotor can be obtained according to the teaching of the present invention.