阅读说明：本技术 单绕组无轴承磁通切换电机缺相邻两相转子悬浮控制方法 (Suspension control method for open-phase adjacent two-phase rotor of single-winding bearingless flux switching motor ) 是由 周扬忠 刘汪彤 屈艾文 杨公德 钟天云 于 2020-12-09 设计创作，主要内容包括：本发明涉及一种单绕组无轴承磁通切换电机缺相邻两相转子悬浮控制方法。将六相绕组电流i-A～i-F经T-6变换,获得转矩平面α-Tβ-T坐标系电流i-(αT)i-(βT)及悬浮平面α-sβ-s坐标系电流i-α-Si-(βS)；根据转子位置角θ-r及i-(αT)i-(βT)计算转矩平面d-Tq-T坐标系电流i-(dT)i-(qT),并分别计算出偏置磁场悬浮力系数K-(PMT)及偏移角而后计算出悬浮电流系数矩阵M-(S-PMT)及其逆矩阵根据xy及其给定值x～*y～*经过xy偏移调节器,获得xy悬浮力给定根据i-(αT)i-(βT)、K-(PMT)、及θ-r计算悬浮平面α-sβ-s坐标系电流给定利用矢量控制方法,获得转矩平面α-Tβ-T坐标系电流给定根据分别计算出剩余健康四相电流给定及中心点电流给定根据和实际电流i-Ci-Di-Ei-Fi-O,通过电流闭环控制输出控制剩余四相健康桥臂及中心点桥臂的开关变量S-CS-DS-ES-FS-O,从而实现电机缺任意相邻两相容错运行控制。(The invention relates to a suspension control method for a single-winding bearingless flux switching motor with two adjacent rotor phases. The six-phase winding current i A ～i F Warp beam T 6 Transforming to obtain a torque plane alpha T β T Coordinate system current i αT i βT And a suspension plane alpha s β s Coordinate system current i α S i βS (ii) a According to rotor position angle theta r And i αT i βT Calculating the torque plane d T q T Coordinate system current i dT i qT And respectively calculating the suspension force coefficient K of the bias magnetic field PMT And offset angle Then is covered withCalculating suspension current coefficient matrix M S_PMT And inverse matrix thereof According to xy and given value x thereof * y * Obtaining xy suspension force given through xy offset regulator According to i αT i βT 、K PMT 、 And theta r Calculating the levitation plane alpha s β s Coordinate system current setting Obtaining a torque plane alpha by a vector control method T β T Coordinate system current setting According to Respectively calculating the remaining healthy four-phase current given value And center point current is given According to And the actual current i C i D i E i F i O Controlling the switch variables S of the rest four-phase healthy bridge arm and the central point bridge arm through the current closed-loop control output C S D S E S F S O Thereby to makeAnd realizing the fault-tolerant operation control of any two adjacent phases of the motor.)

1. A suspension control method for a single-winding bearingless flux switching motor with two adjacent rotor phases, which is characterized in that six-phase winding current i is used_A～i_FWarp beam T₆Matrix transformation to obtain a torque plane alpha_Tβ_TCoordinate system current i_αTi_βTAnd a suspension plane alpha_sβ_sCoordinate system current i_αSi_βS(ii) a According to rotor position angle theta_rAnd i_αTi_βTCalculating the torque plane d_Tq_TCoordinate system current i_dTi_qT(ii) a According to i_dTi_qTRespectively calculating the suspension force coefficient K of the bias magnetic field_PMTAnd offset angleAccording to K_PMT、And theta_rCalculating suspension current coefficient matrix M_{S_PMT}Further calculate M_{S_PMT}Inverse matrix ofAccording to xy and given value x thereof^*y^*Obtaining xy suspension force given through xy offset regulatorAccording toi_αTi_βT、K_PMT、And theta_rCalculating the levitation plane alpha_sβ_sCoordinate system current settingObtaining a torque plane alpha by a vector control method_Tβ_TCoordinate system current settingAccording toRespectively calculating the remaining healthy four-phase current given valueAnd center point current is givenAccording to And the actual current i_Ci_Di_Ei_Fi_OControlling the switch variables S of the rest four-phase healthy bridge arm and the central point bridge arm through the current closed-loop control output_CS_DS_ES_FS_OTherefore, the fault-tolerant operation control of any two adjacent phases of the motor is realized.

2. The suspension control method for the rotor of the single-winding bearingless flux switching motor with two adjacent phases missing according to claim 1, wherein S is used when the upper tube of the corresponding phase bridge arm is switched on and the lower tube of the corresponding phase bridge arm is switched off_i1(i ═ C, D, E, F, O); when the upper tube of the corresponding bridge arm is turned off and the lower tube is turned on S_i0(i ═ C, D, E, F, O); the upper pipe and the lower pipe are conducted complementarily; at S_iUnder the control action, the inverter outputs stator current meeting the requirement, and the fault-tolerant operation control of any two adjacent phases of the motor is realized.

3. The method for controlling the suspension of the two adjacent phase-lacking rotors of the single-winding bearingless flux-switching motor according to claim 1 or 2, wherein the current closed-loop control can obtain S by using a current hysteresis comparator_C～S_OOr S can be obtained by PWM control method_C～S_O。

4. The suspension control method for the two adjacent rotor phases of the single-winding bearingless flux switching motor, which is characterized by comprising the following concrete implementation steps of:

step S1, detecting stator six-phase winding current i by using current sensor and AD conversion channel_A～i_FThe rotor position angle theta is detected by using the rotor position angle sensor and the detection channel_rAnd rotor electrical angular velocity ω_r(ii) a Detecting radial displacements X and Y of the rotor in the X and Y directions by using radial displacement sensors and detection channels of the rotor in the X and Y directions;

step S2, sampling current i_Ai_Bi_Ci_Di_Ei_FWarp beam T₆Matrix transformation to obtain a torque plane alpha_Tβ_TCoordinate system current i_αTi_βTAnd a suspension plane alpha_sβ_sCoordinate system current i_αSi_βS：

Wherein the content of the first and second substances,

step S3, according to theta_rAnd i_αTi_βTCalculating the torque plane d_Tq_TCoordinate system current i_dTi_qTAnd further calculating the suspension coefficient K of the bias magnetic field_PMTAnd offset angleAccording to K_PMT、And theta_rCalculating suspension current coefficient matrix M_{S_PMT}Further calculate M_{S_PMT}Inverse matrix ofAnd according to xy and its given value x^*y^*Obtaining xy suspension force given through xy offset regulator

Step S4 according toi_αTi_βT、K_PMT、And theta_rOutputting a suspension plane alpha through a suspension current given calculation link_sβ_sCoordinate system current setting

Step S5, according to the rotor electrical angular speed settingω_rAnd theta_rCalculating the torque plane alpha_Tβ_TCoordinate system current settingThen according toCalculating the current given at the center pointAnd remaining healthy four-phase current is given

Step S6, theAnd i_Ci_Di_Ei_Fi_OSending the current to a current closed-loop control link, and outputting a switching state S for controlling a healthy four-phase bridge arm and a current bridge arm with a central point_CS_DS_ES_FS_OTherefore, the fault-tolerant operation control of any two adjacent phases of the motor is realized.

5. The levitation control method for the two phase rotor in the open phase of the single-winding bearingless flux switching motor as claimed in claim 4, wherein in step S5, the torque plane α is_Tβ_TCoordinate system current settingThe calculation is as follows:

1) given according to the electrical angular speed of the rotorAnd omega_rObtaining q by a rotational speed PI regulator_TShaft current setting

2) According to the magnetic field control link, obtaining d_TShaft current setting

3) Rotor position angle theta_rAndis fed to d_Tq_TCoordinate direction alpha_Tβ_TConversion link, output

6. The levitation control method for the rotor of the single-winding bearingless flux switching motor of the two adjacent missing phases as recited in claim 4, wherein in step S3,obtaining the available offset error Δ x ═ x of xy^*-x、Δy＝y^*Y passes through the x-offset PI regulator and the y-offset PI regulator output, respectively.

7. The levitation control method for the rotor of the single-winding bearingless flux switching motor of the two adjacent missing phases as recited in claim 4, wherein in step S3,K_PMTandthe calculation method of (2) is as follows:

1) according to i_αTi_βTAnd theta_rThrough α_Tβ_TCoordinate direction d_Tq_TCoordinate transformation link, output d_Tq_TAxial current i_dTi_qT：

2) Handle i_dTi_qTFeeding into K_PMTComputing link, output K_PMT：

3) Handle i_dTi_qTIs sent intoAngle calculation link, output

4) Handle K_PMT、θ_rFeeding into matrix M_{S_PMT}Computing link, output M_{S_PMT}：

5) Solving the inverse matrix according to the matrix inversion formula

8. The levitation control method for the two phase rotor in the open phase of the single-winding bearingless flux switching motor as claimed in claim 4, wherein the center point current i is obtained by the step S6_OIs calculated byThe following were used:

9. the levitation control method for the two phase rotor of the single-winding bearingless flux switching motor in the absence of the phase according to claim 4, wherein in step S6, the switching state quantity S of the inverter bridge arm is_C～S_OIf the calculation adopts the current hysteresis control strategy, the calculation mode is as follows:

when in useWhen S is present_i＝1(i＝C,D,E,F,O)；

When in useWhen S is present_i＝0(i＝C,D,E,F,O)；

Where ε is the allowable current control error.

10. The levitation control method for the two phase rotor of the single-winding bearingless flux switching motor in the absence of the phase according to claim 4, wherein in step S6, the switching state quantity S of the inverter bridge arm is_C～S_OIf the calculation adopts the PWM control strategy, the calculation mode is as follows:

1) calculating the current control error of each branch

2) The current control error of each branch circuit passes through a PI regulator to output the given voltage of each branch circuit

3) Each branch voltage is given and compared with the same triangular carrier wave to output S_C～S_O。

Technical Field

The invention relates to a suspension control method for a single-winding bearingless flux switching motor with two adjacent rotor phases.

Background

The permanent magnet of the bearingless flux switching motor is positioned on the stator, and heat in the stator is easily and rapidly taken away in an air cooling, water cooling or oil cooling mode, so that the demagnetization risk of the permanent magnet caused by temperature rise is effectively reduced; the motor rotor runs in the mechanical center in a magnetic suspension mode, so that the problems of mechanical friction, lubricating oil pollution and the like caused by the traditional mechanical bearing support are avoided, and the high-speed running of the rotor is facilitated. In view of the above advantages, bearingless flux switching motors are receiving great attention from the industry and academia.

The stator winding of the bearingless flux switching motor has a single winding mode and a double winding mode, and because the torque current component and the suspension current component simultaneously flow through the same winding in the single winding mode, the tangential rotation control and the radial suspension control of a rotor are simultaneously realized, and the competition of the double winding mode on the limited space of a stator slot is avoided; only one set of power converter is provided, and the driver is simple; the proportion of the torque current component and the suspension current component can be distributed at will, which is favorable for fully utilizing the sectional area of the winding. Therefore, the single-winding bearingless flux switching motor driving system has great application value.

When two-phase faults occur continuously in a bridge arm of the power converter or a motor winding, the reliability of rotor suspension operation can be greatly enhanced if a fault-tolerant control method is adopted to ensure that the rotor suspension operation can be ensured. When the stator winding adopts a star connection mode and is not led out from a central point, only 5 degrees of freedom are controllable when a six-phase single-winding bearingless flux switching motor system has no fault, and the tangential rotation and radial suspension control of a rotor occupy 4 degrees of freedom; if one phase is lacked, the remaining 4 degrees of freedom can be used for controlling the rotor to rotate in a suspended mode. However, if there are two adjacent phases, the remaining controllable degrees of freedom are only 3, and obviously the requirement of 4 degrees of freedom required for controlling the tangential rotation and radial suspension of the rotor is not enough, how to realize the fault-tolerant operation of the motor is a scientific problem to be solved urgently! The torque current component and the suspension current component in the single-winding bearingless flux switching motor simultaneously flow through the same winding, so that after two phases of the winding are lost, the stability of tangential rotation and radial suspension control of a rotor is greatly influenced.

Therefore, the invention provides a suspension operation control method for any adjacent two-phase rotor of a six-phase single-winding bearingless flux switching motor.

Disclosure of Invention

The invention aims to solve the problem of stable suspension operation of rotors under the condition that a six-phase single-winding bearingless flux switching motor lacks any two adjacent phases, and provides a suspension control method for the rotor of the single-winding bearingless flux switching motor lacking two adjacent phases.

In order to achieve the purpose, the technical scheme of the invention is as follows: a suspension control method for a single-winding bearingless flux switching motor with two adjacent rotor phases in a default phase is characterized in that six-phase winding current i is used_A～i_FWarp beam T₆Matrix transformation to obtain a torque plane alpha_Tβ_TCoordinate system current i_αTi_βTAnd a suspension plane alpha_sβ_sCoordinate system current i_αSi_βS(ii) a According to rotor position angle theta_rAnd i_αTi_βTCalculating the torque plane d_Tq_TCoordinate system current i_dTi_qT(ii) a According to i_dTi_qTRespectively calculating the suspension force coefficient K of the bias magnetic field_PMTAnd offset angleAccording to K_PMT、And theta_rCalculating suspension current coefficient matrix M_{S_PMT}Further calculate M_{S_PMT}Inverse matrix ofAccording to xy and given value x thereof^*y^*Obtaining xy suspension force given through xy offset regulatorAccording toi_αTi_βT、K_PMT、And theta_rCalculating the levitation plane alpha_sβ_sCoordinate system current settingObtaining a torque plane alpha by a vector control method_Tβ_TCoordinate system current settingAccording toRespectively calculating the remaining healthy four-phase current given valueAnd center point current is givenAccording to And the actual current i_Ci_Di_Ei_Fi_OControlling the switch variables S of the rest four-phase healthy bridge arm and the central point bridge arm through the current closed-loop control output_CS_DS_ES_FS_OTherefore, the fault-tolerant operation control of any two adjacent phases of the motor is realized.

In an embodiment of the invention, when the upper tube and the lower tube of the corresponding bridge arm are turned on and off, S_i1(i ═ C, D, E, F, O); when the upper tube of the corresponding bridge arm is turned off and the lower tube is turned on S_i0(i ═ C, D, E, F, O); the upper pipe and the lower pipe are conducted complementarily; at S_iUnder the control action, the inverter outputs stator current meeting the requirement, and the fault-tolerant operation control of any two adjacent phases of the motor is realized.

In an embodiment of the present invention, the current closed-loop control may obtain S by using a current hysteresis comparator_C～S_OOr S can be obtained by PWM control method_C～S_O。

In an embodiment of the present invention, the method specifically includes the following steps:

step S1, detecting stator six-phase winding current i by using current sensor and AD conversion channel_A～i_FThe rotor position angle theta is detected by using the rotor position angle sensor and the detection channel_rAnd rotor electrical angular velocity ω_r(ii) a Detecting radial displacements X and Y of the rotor in the X and Y directions by using radial displacement sensors and detection channels of the rotor in the X and Y directions;

step S2, sampling current i_Ai_Bi_Ci_Di_Ei_FWarp beam T₆Matrix transformation to obtain a torque plane alpha_Tβ_TCoordinate system current i_αTi_βTAnd a suspension plane alpha_sβ_sCoordinate system current i_αSi_βS：

Wherein the content of the first and second substances,

step S3, according to theta_rAnd i_αTi_βTComputingPlane of torque d_Tq_TCoordinate system current i_dTi_qTAnd further calculating the suspension coefficient K of the bias magnetic field_PMTAnd offset angleAccording to K_PMT、And theta_rCalculating suspension current coefficient matrix M_{S_PMT}Further calculate M_{S_PMT}Inverse matrix ofAnd according to xy and its given value x^*y^*Obtaining xy suspension force given through xy offset regulator

Step S4 according toi_αTi_βT、K_PMT、And theta_rOutputting a suspension plane alpha through a suspension current given calculation link_sβ_sCoordinate system current setting

Step S5, according to the rotor electrical angular speed settingω_rAnd theta_rCalculating the torque plane alpha_Tβ_TCurrent supply of coordinate systemStatorThen according toCalculating the current given at the center pointAnd remaining healthy four-phase current is given

Step S6, theAnd i_Ci_Di_Ei_Fi_OSending the current to a current closed-loop control link, and outputting a switching state S for controlling a healthy four-phase bridge arm and a current bridge arm with a central point_CS_DS_ES_FS_OTherefore, the fault-tolerant operation control of any two adjacent phases of the motor is realized.

In one embodiment of the present invention, in step S5, the torque plane α_Tβ_TCoordinate system current settingThe calculation is as follows:

1) given according to the electrical angular speed of the rotorAnd omega_rObtaining q by a rotational speed PI regulator_TShaft current setting

2) According to the magnetic field control link, obtaining d_TShaft current setting

3) Rotor position angle theta_rAndis fed to d_T q_TCoordinate direction alpha_Tβ_TConversion link, output

In one embodiment of the present invention, in step S3,obtaining the available offset error Δ x ═ x of xy^*-x、Δy＝y^*Y passes through the x-offset PI regulator and the y-offset PI regulator output, respectively.

In one embodiment of the present invention, in step S3,K_PMTandthe calculation method of (2) is as follows:

1) according to i_αTi_βTAnd theta_rThrough α_Tβ_TCoordinate direction d_T q_TCoordinate transformation link, output d_T q_TAxial current i_dTi_qT：

2) Handle i_dTi_qTFeeding into K_PMTComputing link, output K_PMT：

3) Handle i_dTi_qTIs sent intoAngle calculation link, output

4) Handle K_PMT、θ_rFeeding into matrix M_{S_PMT}Computing link, output M_{S_PMT}：

5) Solving the inverse matrix according to the matrix inversion formula

6) In one embodiment of the present invention, in step S6, the center point current i_OThe calculation formula of (a) is as follows:

in one embodiment of the present invention, the stepsIn step S6, inverter arm switching state quantity S_C～S_OIf the calculation adopts the current hysteresis control strategy, the calculation mode is as follows:

when in useWhen S is present_i＝1(i＝C,D,E,F,O)；

When in useWhen S is present_i＝0(i＝C,D,E,F,O)；

Where ε is the allowable current control error.

In an embodiment of the invention, in step S6, the inverter bridge arm switch state quantity S_C～S_OIf the calculation adopts the PWM control strategy, the calculation mode is as follows:

1) calculating the current control error of each branch

2) The current control error of each branch circuit passes through a PI regulator to output the given voltage of each branch circuit

3) Each branch voltage is given and compared with the same triangular carrier wave to output S_C～S_O。

Compared with the prior art, the invention has the following beneficial effects:

(1) an additional bridge arm is adopted to independently control the residual healthy 4-phase zero-sequence current, so that the current of the residual healthy 4-phase winding is independently controlled, and 4 controllable degrees of freedom are created; compared with a method for clamping the potential of the center point of the residual healthy 4 phases by the series capacitor, a complex series capacitor balance control algorithm is not needed;

(2) after two phases are lost, tangential rotation and radial suspension control of the rotor are decoupled with each other, so that fault-tolerant uninterrupted suspension rotation operation of the motor with two phases is realized, and the reliability of a six-phase single-winding bearingless flux switching motor driving system is improved.

Drawings

Fig. 1 is a block diagram of a suspension operation control structure of any adjacent two-phase rotor of a six-phase single-winding bearingless flux switching motor.

Fig. 2 is a schematic cross-sectional view of a six-phase single-winding bearingless flux-switching machine.

Fig. 3 shows a hardware structure of a driving system according to an embodiment of the present invention.

FIG. 4 illustrates the coordinate system and associated variable definitions of the present invention.

Detailed Description

The technical scheme of the invention is specifically explained below with reference to the accompanying drawings.

The invention relates to a suspension control method for a phase-lacking adjacent two-phase rotor of a single-winding bearingless flux switching motor, which aims to solve the problem that the controllable degree of freedom cannot meet the requirement of 4 degrees of freedom in total for tangential control and radial suspension control of a rotor after a winding lacks any adjacent two phases, and adds an inverter bridge arm to control the central point current of the remaining healthy 4 phases so as to realize the purpose of independently controlling the remaining healthy 4-phase current; in order to solve the problem that the rotation and suspension of a rotor are not decoupled after a winding lacks any two adjacent phases, a mathematical model of the torque and the suspension force of the motor lacking any two adjacent phases is provided, and a torque and suspension force decoupling control method is constructed. Because the mathematical model of the motor without any two adjacent phases in the 6-phase winding can be equivalent to the mathematical model without the AB two phases, the AB two phases are taken as an example in the invention. The specific explanation is as follows.

The block diagram of the suspension operation control structure of any adjacent two-phase rotor of the six-phase single-winding bearingless flux switching motor provided by the invention is shown in fig. 1. The control structure shown in figure 1 comprises a seven-phase inverter bridge arm, a six-phase single-winding bearingless flux switching motor and a current i_Ai_Bi_Ci_Di_Ei_FMeasuring link, rotor position angle theta_rMeasuring link, rotor radial displacement xy measuring link and torque plane alpha_Tβ_TCoordinate system current settingCalculating link, suspension plane alpha_sβ_sCoordinate system current settingCalculating link, central point current settingComputing link, remaining healthy four-phase current settingA calculation link, a current closed-loop control link and the like. Six-phase winding current i_A～i_FThrough T₆Matrix transformation to obtain a torque plane alpha_Tβ_TCoordinate system current i_αTi_βTAnd a suspension plane alpha_sβ_sCoordinate system current i_αSi_βS(ii) a According to rotor position angle theta_rAnd i_αTi_βTCalculating the torque plane d_Tq_TCoordinate system current i_dTi_qT(ii) a According to i_dTi_qTRespectively calculating the suspension force coefficient K of the bias magnetic field_PMTAnd offset angleAccording to K_PMT、And theta_rCalculating suspension current coefficient matrix M_{S_PMT}(ii) a Further calculate M_{S_PMT}Inverse matrix ofAccording to xy and given value x thereof^*y^*Obtaining xy suspension force given through xy offset regulatorAccording toi_αTi_βT、K_PMT、And theta_rCalculating the levitation plane alpha_sβ_sCoordinate system current settingObtaining a torque plane alpha by a vector control method_Tβ_TCoordinate system current settingAccording toRespectively calculating the remaining healthy four-phase current given valueAnd center point current is givenAccording to And the actual current i_Ci_Di_Ei_Fi_OControlling the switch variables S of the rest four-phase healthy bridge arm and the central point bridge arm through the current closed-loop control output_CS_DS_ES_FS_O. S when the upper tube of the corresponding bridge arm is switched on and the lower tube is switched off_i1(i ═ C, D, E, F, O); when the upper tube of the corresponding bridge arm is turned off and the lower tube is turned on S_i0(i ═ C, D, E, F, O); the upper pipe and the lower pipe are conducted complementarily. At S_iUnder the control action, the inverter outputs stator current meeting the requirement, and the fault-tolerant operation control of any two adjacent phases of the motor is realized. The current closed-loop control can adopt a current hysteresis comparator to obtain S_C～S_O(ii) a PWM can also be adoptedThe control method obtains S_C～S_O。

The cross-sectional structure of the motor studied by the present invention is shown in fig. 2, and the rotor is composed of a core having 10 teeth. The stator is laminated by 12U-shaped punching sheets, a permanent magnet magnetized along the tangential direction is clamped between two adjacent U-shaped punching sheets, and the magnetizing directions of the adjacent permanent magnets are opposite. Each stator tooth is sleeved with a coil, and 12 coils form 6-phase windings A-F according to the connection mode in figure 2. The axes of the coil A1 and the coil A2 are defined as the x axis and the y axis of a rectangular coordinate system xy respectively.

The invention provides a fault-tolerant uninterrupted operation control method aiming at the problem that a six-phase single-winding bearingless flux switching motor lacks any adjacent two-phase rotor suspension rotation control, which is technically characterized by sequentially comprising the following steps of:

1) stator six-phase winding current i is detected by using a current sensor and an AD conversion channel_A～i_FThe rotor position angle theta is detected by using the rotor position angle sensor and the detection channel_rAnd rotor electrical angular velocity ω_r(ii) a And detecting the radial displacement X and Y of the rotor in the X and Y directions by using the radial displacement sensors in the X and Y directions of the rotor and the detection channel.

2) Sampling current i_Ai_Bi_Ci_Di_Ei_FThrough T₆Transforming to obtain a torque plane alpha_Tβ_TCoordinate system current i_αTi_βTAnd a suspension plane alpha_sβ_sCoordinate system current i_αSi_βS：

Wherein the content of the first and second substances,

3) according to the rotation speedSpeed of rotation omega_rAnd theta_rCalculating the torque plane alpha_Tβ_TCoordinate system current setting

4) According toi_αTi_βT、K_PMT、And theta_rOutputting a suspension plane alpha through a suspension current given calculation link_sβ_sCoordinate system current setting

5) According toCalculating the current given at the center pointAnd remaining healthy four-phase current is given

6) Will be provided withAnd i_Ci_Di_Ei_Fi_OSending the current to a current closed-loop control link, and outputting a switching state S for controlling a healthy four-phase bridge arm and a current bridge arm with a central point_CS_DS_ES_FS_O。

Torque plane α in step 3) above_Tβ_TCoordinate system current settingThe calculation is as follows:

a (1.1) is given according to the rotational speedAnd the actual rotational speed omega_rObtaining q by a rotational speed PI regulator_TShaft current setting

A (1.2) controlling the link according to the magnetic field to obtain d_TShaft current setting

A (1.3) the rotor position angle theta_rAndis fed to d_T q_TCoordinate direction alpha_Tβ_TConversion link, output

In the above step 4)Is obtained byTo make use of the offset error Δ x of xy ═ x^*-x、Δy＝y^*Y passes through the x-offset PI regulator and the y-offset PI regulator output, respectively.

In the above step 4)K_PMTAndthe calculation method is as follows:

b (1.1) according to i_αTi_βTAnd theta_rThrough α_Tβ_TCoordinate direction d_T q_TCoordinate transformation link, output d_T q_TAxial current i_dTi_qT：

B (1.2) treating i_dTi_qTFeeding into K_PMTComputing link, output K_PMT：

B (1.3) treating i_dTi_qTIs sent intoAngle calculation link, output

B (1.4) K_PMT、θ_rFeeding into matrix M_{S_PMT}Computing link, output M_{S_PMT}：

B (1.5) solving the inverse matrix according to the matrix inversion formula

Center point current i in step 6)_OThe acquisition can be performed by the following method:

inverter bridge arm switch state quantity S in step 6)_C～S_OIf the calculation adopts a current hysteresis control strategy, the method comprises the following steps:

when in useWhen S is present_i＝1(i＝C,D,E,F,O)；

When in useWhen S is present_i＝0(i＝C,D,E,F,O)；

Where ε is the allowable current control error.

Inverter bridge arm switch state quantity S in step 6)_C～S_OIf the calculation adopts the PWM control strategy, the method comprises the following steps:

c (1.1) calculating the current control error of each branch

C (1.2) each branch current control error passes through PI regulator, and each branch voltage setting is output

C (1.3) each branch voltage given is compared with the same triangular carrier wave to output S_C～S_O。

The hardware structure of the driving system of the embodiment of the invention is shown in fig. 3. The method comprises the following steps: the device comprises a rectification circuit, a filter capacitor, a direct current bus voltage acquisition circuit, a seven-phase inverter, a bearingless flux switching motor, a six-phase winding current acquisition circuit, a motor rotor position angle acquisition circuit, a rotor radial xy offset acquisition circuit, an isolation drive, a central controller, a human-computer interface and the like. Wherein the six-phase inverter dc bus voltage may also be provided using a suitable dc power supply. The power tube in the inverter adopts IGBT or MOSFET, and the central controller adopts DSP or singlechip. The winding current acquisition circuit is formed by combining a Hall current sensor and an operational amplifier, and can also be formed by combining a branch circuit series power resistor and a differential operational amplifier. The Hall scheme can effectively realize the electrical isolation of the control loop and the main loop, and the branch power resistor scheme can reduce the cost of the driving system. The direct current bus voltage acquisition circuit is formed by combining a Hall voltage sensor and an operational amplifier, and can also be formed by combining a voltage follower formed by an operational amplifier after voltage division of a parallel resistor. The rotor position angle detection circuit can be formed by connecting a rotary encoder with a level conversion circuit and can also be formed by connecting a rotary transformer with a decoding circuit, wherein the cost of the former is lower, but the position angle sampling precision is limited by the number of lines of the encoder, and the cost of the latter is higher, but the position angle sampling precision is higher. The rotor radial xy offset acquisition circuit is formed by combining an eddy current sensor and a subsequent operational amplifier, and can also be formed by combining a linear optical coupler and a subsequent operational amplifier. Weak current signals output by the current detection circuit, the voltage sampling circuit and the rotor radial xy offset acquisition circuit are sent to the A/D conversion module of the central controller, and pulse signals output by the position angle detection circuit are sent to the QEP module of the central controller. According to the obtained signals and the rotor levitation operation control method, switching signals of an inverter bridge arm are output, and the switching action of a power switching tube in an inverter is controlled through an isolation driver.

The basic principle is described as follows:

for analytical convenience, coordinate systems and variables are defined as shown in fig. 4:

the A-F are six-phase winding axes respectively, and the A-phase winding axis leads the x-axis by 9-degree mechanical angle; alpha is alpha_sβ_sFor a stationary coordinate system of the levitation plane, d_sq_sIs a suspended plane rotating coordinate system, and the included angle between the d axis and the alpha axis isi_AS～i_FSIs a component of the levitation current in the six-phase winding, i_αS、i_βSIs a plane of suspension alpha_sβ_sCoordinate system levitation current, i_dS、i_qSIs a suspended plane d_sq_sThe coordinate system suspends the current. Alpha is alpha_Tβ_TFor the static coordinate system of the torque plane, the alpha T axis is coincident with the axis of the A-phase winding, and theta_rCoupling flux linkage vector psi for six-phase winding permanent magnet_fThe angle between the shaft and the alpha T axis is also called the rotor position angle. d_Tq_TFor the torque plane synchronous rotation of the coordinate system, d_TAxis and psi_fIn the same direction. Torque current vector i_sTAt α_Tβ_TAxial components are each i_αTi_βTAt d_Tq_TAxial components are each i_dTi_qT. In addition to the above shafting definitions, zero sequence shafting o1 and o2 also exists in normal six-phase drive systems. Because any two adjacent phase motor mathematical model in the 6-phase winding can be equivalent to an AB two-phase mathematical model, the AB two-phase is taken as an example in the invention.

Deducing suspension current i respectively introduced into A-F phases by using a magnetic circuit analysis method_AS～i_FSThen, the expression of the levitation force in the x and y directions generated under the bias magnetic field established by the permanent magnet and the torque current is as follows:

wherein the content of the first and second substances,is the bias magnetic field suspension force coefficient; k is a radical of_PMIs the amplitude of the suspension force generated by unit suspension current under the magnetic field of the permanent magnet; k is a radical of_{PM_DC}Is the suspension force direct current bias generated by unit suspension current under the magnetic field of the permanent magnet. k is a radical of_dTIs the unit alpha_SShaft levitation current or beta_SAxial levitation current in unit i_dTThe amplitude of the fundamental component of the suspension force in the x and y directions generated under the magnetic field established by the current. k is a radical of_qTIs the unit alpha_SAxis suspension current or beta S axis suspension current in unit i_qTThe fundamental component amplitude of the suspension force in the x and y directions generated under the magnetic field established by the current. K_PM、k_{PM_DC}、k_dT、k_qTCan be obtained by finite element simulation with segments.

T of the form₆The transformation matrix transforms 6 phasors to alpha_Tβ_Tα_Sβ_So1 o2 axis on:

thus at a known current i_αTi_βT i_αSi_βS i_o1i_o2The result of the 6-phase current can be found as follows:

in the absence of phase A, there is i_AWhen it is 0, the formula is obtained from formula (8)

Simplified to obtain zero sequence current i_o1The following were used:

substituting the formula (10) into the formula (8) to obtain the following six-phase winding current after the A phase is lacked:

and further analyzing the AB-lacking two-phase control method on the basis of the A-lacking phase current conclusion. At this time, the A, B phase currents are all equal to 0, and the zero-sequence current i is obtained by the formula (11)_o2

Substituting formula (12) into formula (11) to obtain A, B-two-phase-lacking six-phase winding current:

if the six-phase winding current is controlled according to the formula (13), the electromagnetic torque T generated by the motor_eThe following were used:

wherein n is_pThe number of the magnetic pole pairs of the motor is counted; e_m＝ω_rΨ_f. As can be seen from equation (14), in the absence of AB, the torque plane current i can be used_αT i_βTThe electromagnetic torque is controlled, and the suspension plane current has no influence on the electromagnetic torque.

Adding the formulas (3), (4), (5) and (6), substituting the specific expressions of the suspension current components of each phase in the formula (13), and deriving the suspension force expression of the rotor after two phases lack AB as follows:

wherein the content of the first and second substances,

according to equation (15), if the torque plane current i is known_αT i_βTSuspension force F_x F_yRotor position angle theta_rAngle of deviationAnd i_dTi_qTThen, the required current i of the suspension plane can be solved_αSi_βS. The following were used:

when the motor lacks the AB two-phase winding, the remaining healthy phases are only 4 phases. If the remaining 4 phases are still star-connected and no central point is led out, it is obviously impossible to try to control the 4-phase winding current by using 4 bridge arms. Therefore, the invention additionally adds a bridge arm for controlling the current of the central point, realizes the independent control of the current of the rest 4-phase winding, and meets the requirements of 4 control degrees of freedom of tangential rotation and radial suspension of the rotor.

The center point current i of the 4-phase healthy phase can be obtained from the formula (13)_OThe following were used:

according to the theoretical analysis, after two phases AB are absent, the suspension plane current has no influence on the torque, so the torque control can adopt vector control; after the torque plane current and the suspension force requirement are obtained, the suspension plane current can be further obtained according to the formula (16); thereafter, the remaining healthy 4-phase current is further obtained from the equation (13), and the center point current i is obtained from the equation (17)_O. And then 4-phase healthy phase current and central point current closed-loop control is realized by using a 4-phase bridge arm and a central point bridge arm.

The above are preferred embodiments of the present invention, and all changes made according to the technical scheme of the present invention that produce functional effects do not exceed the scope of the technical scheme of the present invention belong to the protection scope of the present invention.