阅读说明：本技术 基于非等长传输线的双频带异向功率放大器及其设计方法 (Dual-band different-direction power amplifier based on unequal-length transmission lines and design method thereof ) 是由 陈世昌 王伟伟 徐魁文 赵鹏 王高峰 于 2019-08-22 设计创作，主要内容包括：本发明公开基于非等长传输线的双频带异向功率放大器及其设计方法。本发明将传统的异向功率放大器的各结构包括输入匹配电路、偏置电路、以及非等长功率合成电路采用特殊的宽带以及双带结构来替换各部分进而实现双带射频工作。其中输入匹配电路利用切比雪夫阶跃式宽带匹配方法。非等长功率合成电路采用混合T/Pi型结构替换传统的非等长阻抗变换器实现双频带不同阻抗的变换功率合并以及虚部补偿。最后通过信号调理电路得到的等幅异向两路信号作为输入信号,从何实现了双频带异向射频功率放大器。本发明所提出的双频带射频异向功率放大器填补了非等长双频带异向功率放大器的设计空白,并且设计思路简单,便于推广。(The invention discloses a dual-band different-direction power amplifier based on unequal length transmission lines and a design method thereof. The invention replaces each part of the traditional structure of the different-direction power amplifier, including the input matching circuit, the bias circuit and the unequal length power synthesis circuit, with a special broadband and double-band structure to realize the double-band radio frequency work. Wherein the input matching circuit utilizes a Chebyshev step-wise broadband matching method. The unequal length power synthesis circuit adopts a mixed T/Pi type structure to replace a traditional unequal length impedance transformer to realize the transformation power combination and imaginary part compensation of different impedances of the dual-frequency band. And finally, the two paths of signals with the same amplitude and different directions obtained by the signal conditioning circuit are used as input signals, so that the dual-band different-direction radio frequency power amplifier is realized. The dual-band radio frequency anisotropic power amplifier provided by the invention fills the design blank of a non-isometric dual-band anisotropic power amplifier, and has the advantages of simple design idea and convenience for popularization.)

1. The dual-band different-direction power amplifier based on the unequal length transmission lines is characterized by comprising an input signal conditioning circuit and two paths of dual-band high-efficiency different-direction power amplifying circuits;

the input signal conditioning circuit adjusts the radio frequency modulation signal to be amplified into two paths of signals with equal amplitude and different directions, and the two paths of signals are respectively used as input signals of two paths of double-band input matching circuits;

the input signal conditioning circuit comprises two paths of digital-to-analog conversion chips, a low-pass filter, an orthogonal mixer and a self-adaptive gain controller which are sequentially connected in series;

each path of the two paths of double-frequency-band high-efficiency different-direction power amplifying circuits comprises a double-band input matching circuit, a bias circuit, a power supply circuit, a transistor, a double-band output matching circuit and a double-band unequal-length power synthesis circuit; the dual-band output matching circuit is fused into the dual-band unequal-length power synthesis circuit to form a new dual-band power synthesis circuit; the input end of the double-band input matching circuit is connected with the signal conditioning circuit, and the output end of the double-band input matching circuit is connected with the grid electrode of the transistor; the input end of the bias circuit is connected with a direct current power supply, and the output end of the bias circuit is connected with the grid electrode of the transistor; the input end of the power supply circuit is connected with a direct current power supply, and the output end of the power supply circuit is connected with the drain electrode of the transistor; the drain electrode of the transistor is connected with the input end of the new double-band power synthesis circuit, and the output ends of the two paths of new double-band power synthesis circuits are connected together and output to the load end;

due to parasitic capacitance C_dsAs a determinant factor of parasitic parameters of the transistor, the invention connects-C in parallel at the output end of the transistor_dsIn such a way as to cancel the parasitic capacitance, then-C can be added_dsFusing the two-band power synthesis circuit with unequal length at the rear end to form a new two-band power synthesis circuit;

the dual-band unequal-length power synthesis circuit is based on an optimal value obtained by the single-frequency lower power synthesis circuit and is used as a target value of a dual-band;

the power synthesis circuit under the single frequency band adopts the theory of non-equal length transmission lines to realize the design purpose, and specifically comprises the following steps:

the two paths of output currents are respectively as follows:

wherein i₅And i₆Respectively output current v for upper and lower circuits_opepFor maximum output voltage, theta is at a compensation angle, R, varying from 0 to 90 degrees_oIs the load impedance; j represents the imaginary symbol; in order to ensure the high efficiency of the anisotropic transistor, the output impedance which needs to be seen at the output end of the transistor is pure real, namely the influence of reactive impedance needs to be eliminated; knowing the current-voltage relationship of the output end, according to the ABCD transmission matrix, the following equation is obtained:

to obtain an output admittance of the transistor

Wherein G is₁、B₁Respectively conductance and susceptance; r_LIn order to be the characteristic impedance of the transmission line,

OBO＝20·log(sinθ).

formula (8)

Wherein t ═ R_L/R_OOBO is a backspacing range, and the final product is obtained

under the above single frequency belt

the mixed T/Pi type structure comprises a first T microstrip line, a second microstrip line and a third T microstrip line; the first T microstrip line can be further divided into three parts, one end of the first part of the first T microstrip line is connected with the drain electrode of the transistor and one end of the second T microstrip line, the other end of the first part is connected with one end of the second part and one end of the third part, the other end of the second part is suspended, and the other section of the third part is grounded or suspended; the first partIs divided into implementation f₁Imaginary part compensation at frequency, second part elimination f₂Imaginary part compensation pair f₁Influence of imaginary part compensation, third part implementing f₂Imaginary compensation at frequency; the straight line where the first part and the third part are located is perpendicular to the second part, so that a T-shaped structure is formed; the structure and the function of the third T microstrip line are the same as those of the first T microstrip line; the other end of the second microstrip line is connected with one end of the first part of the third T microstrip line and is used as a port B; the ports B of the two paths of second microstrip lines are connected to a combining point together, so that the purpose of combining and restoring signals is achieved; the three microstrip lines jointly form a mixed T/Pi type structure to meet the requirement of different impedance transformation of dual-frequency bands.

2. The dual-band anisotropic power amplifier based on unequal length transmission lines of claim 1, wherein the dual-band input matching circuit adopts a high-low impedance matching method.

3. The dual band differential power amplifier according to claim 1, wherein the bias circuit and the power supply circuit respectively provide voltages to the gate and the drain of the transistor and suitable gate voltage and power to the transistor; the bias circuit provides a dc power supply for the transistor and may be powered with a constant or dynamically variable voltage to provide the appropriate bias state and quiescent operating point.

4. The dual band differential power amplifier based on unequal length transmission lines according to claim 1, wherein in the bias circuit and the supply circuit, for f₁Second harmonic use at frequency f₁One end of a 90-degree impedance conversion line under the frequency is connected with a direct current power supply, and the other end of the impedance conversion line is connected with the drain electrode of a transistor, so that f is eliminated₁Influence of the second harmonic of frequency, on f₂Second harmonic of frequency, using parallel connection f₂And in the 90-degree impedance transformation line mode at the frequency, one end of the impedance transformation line is grounded through a direct current blocking capacitor, and the other end of the impedance transformation line is connected with the drain electrode of the transistor.

5. The design method of the dual-band different-direction power amplifier based on the unequal length transmission lines is characterized by comprising the following steps of:

the method comprises the following steps: the modulation signal to be transmitted is subjected to angle modulation at a PC end, a digital-to-analog conversion chip, a low-pass filter, a frequency mixer and a self-adaptive gain amplifier to obtain two paths of modulation signals to be input with equal amplitude and different directions; the digital-to-analog conversion chip converts the digital baseband signal into an analog modulation signal; the low-pass filter is used for eliminating clutter components of the baseband signal; the quadrature mixer is used for modulating the baseband signal to a carrier frequency; the adaptive gain controller is used for adjusting the amplitude of the radio frequency modulation signal to a proper size to be input into the back-end amplifier;

step two: selecting two same transistors according to design indexes, and determining corresponding input impedance and output impedance;

step three: designing an input matching circuit by adopting a high-low impedance matching method according to input impedance;

step four: debugging a bias circuit:

two ends of a 90-degree electric length transmission line are respectively connected with a power supply and a transistor grid to eliminate the influence of second harmonic;

step five: debugging the power supply circuit:

for the second harmonic wave under the frequency of f1, one end of a 90-degree impedance transformation line under the frequency of f1 is connected with a direct-current power supply, and the other end of the impedance transformation line is connected with the drain electrode of a transistor, so that the influence of the second harmonic wave of the frequency of f1 is eliminated; for the second harmonic wave under the frequency of f2, a 90-degree impedance transformation line mode under the frequency of f2 is adopted, wherein one end of the 90-degree impedance transformation line is grounded through a direct-current blocking capacitor, and the other end of the 90-degree impedance transformation line is connected with the drain electrode of the transistor;

step six: and debugging the single-band unequal-length transmission lines under two independent frequency bands according to the output impedance:

the two paths of output currents are respectively as follows:

wherein i₅And i₆Respectively output current v for upper and lower circuits_opepFor maximum output voltage, theta is at a compensation angle, R, varying from 0 to 90 degrees_oIs the load impedance, typically set at 50 ohms; j represents the imaginary symbol; in order to ensure the high efficiency of the anisotropic transistor, the output impedance which needs to be seen at the output end of the transistor is pure real, namely the influence of reactive impedance needs to be eliminated; knowing the current-voltage relationship of the output end, according to the ABCD transmission matrix, the following equation is obtained:

to obtain an output admittance of the transistor

Wherein G is₁、B₁Respectively conductance and susceptance; r_LIn order to be the characteristic impedance of the transmission line,

OBO＝20·log(sinθ).

formula (8)

Step seven: compensating the electrical length obtained according to the sixth stepAnd R_L、R_OThe relation between the two microstrip lines is that the single-strip unequal length transmission line is equivalently replaced by a mixed T/Pi dual-strip unequal length power synthesis circuit, wherein the mixed T/Pi dual-strip unequal length power synthesis circuit structure comprises a first T microstrip line, a second microstrip line and a third T microstrip line; the first T microstrip line can be further divided into three parts, one end of the first part of the first T microstrip line is connected with the drain electrode of the transistor and one end of the second T microstrip line, the other end of the first part is connected with one end of the second part and one end of the third part, the other end of the second part is suspended, and the other section of the third part is grounded or suspended; first part implementation f₁Imaginary part compensation at frequency, second part elimination f₂Imaginary part compensation pair f₁Influence of imaginary part compensation, third part implementing f₂Imaginary compensation at frequency; the straight line where the first part and the third part are located is perpendicular to the second part, so that a T-shaped structure is formed; the structure and the function of the third T microstrip line are the same as those of the first T microstrip line; the other end of the second microstrip line is connected with one end of the first part of the third T microstrip line and is used as a port B; the ports B of the two paths of second microstrip lines are connected to a combining point together, so that the purpose of combining and restoring signals is achieved;

step eight: designing an output matching circuit, the output matching aiming at combining resistanceImpedance matching to real impedance presented by the dual-band unequal-length power synthesis circuit; the output complex impedance presented by the transistor is mainly due to the influence of parasitic capacitance; thus compensating the parasitic capacitance-C at the output of the transistor by parallel connection_dsTo cause the transistor to appear as a real impedance; due to-C_dsIs a virtual circuit part, so that the capacitor-C will be compensated_dsIntegrating the power combining circuit with the designed dual-band unequal-length output power to finally form a new dual-band power combining circuit;

step nine: combining the input matching circuit, the bias circuit, the power supply circuit and the new dual-band power synthesis circuit which are debugged in the step to form one path of dual-band high-efficiency different-direction power amplification circuit; the two paths of double-frequency-band high-efficiency anisotropic power amplifying circuits respectively receive the two paths of modulated signals with equal amplitude and different directions after the first step of processing.

Technical Field

The invention relates to the field of radio frequency microwave communication, and provides a novel high-efficiency radio frequency anisotropic power amplifier capable of working under two independent frequency bands and a design method thereof.

Background

With the rapid development of communication technology, especially wireless communication, the rf microwave technology carrying the development thereof is becoming more and more important. But the limited frequency band resource becomes another challenge. Therefore, in order to transmit more communication information in a limited frequency band, the envelope of the modulation signal used in the current communication technology is changed dramatically, that is, the peak-to-average power ratio (abbreviated as "peak-to-average ratio") of the signal is large. The function of the radio frequency transmitter is to transmit radio frequency signals completely, accurately and losslessly, and the transmission distance is required to be as far as possible, so that as the most critical energy consumption element of the radio frequency transmitter, the performance of the radio frequency power amplifier greatly determines the performance of the radio frequency transmitter. The design goal of the rf power amplifier is to enable lossless long-distance transmission of signals and to minimize power consumption. Therefore, a linear and efficient power amplifier needs to be designed. However, in order to meet the requirement of modern communication technology for high peak-to-average ratio signal transmission, new requirements are put on the power back-off efficiency of the power amplifier based on the original design target. Power amplifiers can be classified as A, B, AB-class linear power amplifiers or C-class nonlinear power amplifiers. The linearity of the linear power amplifier is good but the saturation efficiency is not high, and the linearity of the nonlinear power amplifier is poor but the saturation efficiency is high. Therefore, class AB power amplifiers, which maintain high linearity while maintaining high efficiency to the maximum extent, are currently popular. On the basis of the class AB power amplifier, in order to further improve the power back-off efficiency, the current mainstream design methods include a Doherty power amplifier and an alien power amplifier. The anisotropic power amplifier is characterized in that a radio frequency signal to be transmitted is changed into two paths of constant-amplitude anisotropic radio frequency signals through a signal conditioning circuit and then is input into the radio frequency power amplifier. Since the amplitude is constant, the power amplifier is insensitive to amplitude information, and thus a non-linear power amplifier may be selected for amplification. The two paths of constant-amplitude anisotropic signals are subjected to power amplification and then are combined and output, so that the original signals are restored. The mainstream power synthesis circuit has two types, namely an isolated type and a non-isolated type. The isolated synthesis circuit has a low efficiency at the power back-off point. The non-isolated structure, taking a classic Chireix type as an example, eliminates reactive impedance by adding a parallel branch knot, can effectively improve output power and improve the efficiency of a back-off point. The classical Chireix architecture requires the addition of reactance compensation elements for the different sub-amplifiers to counteract the adverse effects of reactive impedance. However, the components of the reactance compensation element easily increase the complexity of the power amplifier circuit, and it is difficult to satisfy the requirement of multi-frequency point operation. In recent years, researchers find that the function of the type Chireix can be realized by using transmission lines with unequal lengths, the traditional reactance compensation structure is omitted, and the circuit structure is simplified on the basis of effectively ensuring the characteristics of the outphasing amplifier.

With the popularization of 5G communications and the arrival of future 6G communications, it is imperative that multi-band or broadband functionality be supported. It is therefore desirable to develop a high efficiency power amplifier that can support multi-band operation. Multiband anisotropic power amplifiers have also of course been the focus of academic and industrial research. However, due to the dispersion effect of the presence of microstrip lines and the strict angle control requirements required for the operation of the outphasing amplifier, it becomes very difficult to design a dual-band and wide-band anisotropic power amplifier. Due to the imperfection of the early design theory, the design scheme of the multi-frequency anisotropic power amplifier at home and abroad is basically blank at present.

In view of the deficiencies of the prior art, there is a need for a versatile solution for a compact outphasing amplifier with multiple frequencies.

Disclosure of Invention

In view of the above, the present invention provides a compact anisotropic high efficiency power amplifier capable of operating in two frequency bands. The dual-band function of the outphasing amplifier is effectively realized by perfecting the design theory of constructing a power synthesis network by applying the unequal length transmission lines and replacing the unequal length transmission lines under the condition of a single band by using a mixed T/Pi dual-band unequal length power synthesis circuit. Meanwhile, a parallel imaginary part compensation circuit required in the traditional design is omitted, so that the whole circuit is simplified, and the performance is greatly improved.

In order to fill the defects of the prior art, the invention adopts the following technical scheme:

a dual-band high-efficiency different-direction power amplifier comprises an input signal conditioning circuit and two paths of dual-band high-efficiency different-direction power amplifying circuits.

The input signal conditioning circuit adjusts the radio frequency modulation signal to be amplified into two paths of signals with equal amplitude and different directions, and the two paths of signals are respectively used as input signals of two paths of double-band input matching circuits; namely, the conversion from a baseband signal to a radio frequency signal is realized, and the necessary functions of pre-amplification, filtering and the like are completed at the same time.

Preferably, the input signal conditioning circuit includes two digital-to-analog conversion chips, a low-pass filter, a quadrature mixer, and a self-adaptive gain controller, which are sequentially connected in series, and finally outputs the radio frequency modulation signal to be input.

The digital-to-analog conversion chip converts the digital baseband signal into an analog modulation signal; the low-pass filter is used for eliminating clutter components of the baseband signal; the quadrature mixer is used for modulating the baseband signal to a carrier frequency; the adaptive gain controller is used for adjusting the amplitude of the radio frequency modulation signal to a proper size to be input into the back-end amplifier. The above techniques and devices are well known.

Each path of the two paths of double-frequency-band high-efficiency different-direction power amplifying circuits comprises a double-band input matching circuit, a bias circuit, a power supply circuit, a transistor, a double-band output matching circuit and a double-band unequal-length power synthesis circuit;

the output matching circuit can present a complex impedance form due to the influence of transistor parasitic parameters, so that the space and complexity are increased during output matching, and the dual-band output matching circuit is fused into a dual-band unequal-length power synthesis circuit to form a new dual-band power synthesis circuit so as to overcome the defects.

The input end of the double-band input matching circuit is connected with the signal conditioning circuit, and the output end of the double-band input matching circuit is connected with the grid electrode of the transistor; the input end of the bias circuit is connected with a direct current power supply, and the output end of the bias circuit is connected with the grid electrode of the transistor; the input end of the power supply circuit is connected with a direct current power supply, and the output end of the power supply circuit is connected with the drain electrode of the transistor; the drain electrode of the transistor is connected with the input end of the new double-band power synthesis circuit, and the output end of the new double-band power synthesis circuit is connected together and output to the load end.

The double-band input matching circuit is carried out by adopting a high-low impedance (step type broadband) matching method, the method is carried out by utilizing an alternative mode of a low-pass filter prototype, a series inductor and a parallel capacitor, and the numerical values of the series inductor and the parallel capacitor are determined by a Chebyshev filter design method. Lumped elements such as inductors and capacitors are suitable for low frequency environment, and microstrip lines are more suitable for high frequency environment, so that the lumped elements need to be converted into microstrip lines. The series inductance is approximately equal to the series high-impedance microstrip line, and the shunt capacitance is approximately equal to the series low-impedance microstrip line. Finally, high-low impedance alternative step type broadband matching is formed, and the aim of double-band matching is achieved. The chebyshev step-wise broadband matching described above is a well-known technique.

The two transistors are basic amplifiers, and can adopt various high-efficiency power amplifier forms such as AB type, B type and the like.

The bias circuit and the power supply circuit respectively provide voltage for a grid electrode and a drain electrode of the transistor and provide proper grid voltage and power for the transistor; preferably, the bias circuit provides a dc power supply to the transistor, which may be powered with a constant or dynamically variable voltage to provide the appropriate bias state and quiescent operating point.

Since the non-linearity of the transistor generates harmonics, the effect of which on the result is the greatest, particularly the second harmonic, it is desirable to try to eliminate the second harmonic and reduce the effect of the harmonics on the non-linearity. In the bias circuit and the power supply circuit, for f₁Second harmonic use at frequency f₁90 degree impedance transformation line at frequency, one endConnecting a direct current power supply, and connecting the other end of the direct current power supply with the drain electrode of the transistor to eliminate f₁Influence of the second harmonic of frequency, on f₂Second harmonic of frequency, using parallel connection f₂And in the 90-degree impedance transformation line mode at the frequency, one end of the impedance transformation line is grounded through a direct current blocking capacitor, and the other end of the impedance transformation line is connected with the drain electrode of the transistor.

Due to parasitic capacitance C_dsAs a determinant factor of parasitic parameters of the transistor, the invention connects-C in parallel at the output end of the transistor_dsIn such a way as to cancel the parasitic capacitance, then-C can be added_dsAnd the power is fused into a double-band unequal-length power synthesis circuit at the rear end to form a new double-band power synthesis circuit.

Preferably, the dual-band unequal-length power combining circuit realizes physical characteristics of different impedances and electrical lengths of dual bands by a T/PI type combined structure, so as to realize functions of imaginary part compensation, real impedance matching and power combining.

The dual-band unequal-length power synthesis circuit is based on an optimal value obtained by the single-frequency lower power synthesis circuit and is used as a target value of a dual-band.

The power synthesis circuit under the single frequency band adopts the theory of non-equal length transmission lines to realize the design purpose, and specifically comprises the following steps:

the two paths of output currents are respectively as follows:

wherein i₅And i₆Respectively output current v for upper and lower circuits_opepFor maximum output voltage, theta is at a compensation angle, R, varying from 0 to 90 degrees_oThe load impedance is generally set to 50 ohms. j denotes the imaginary sign. In order to ensure high efficiency of the non-inverting transistor, the output impedance seen at the output of the transistor needs to be purely real, i.e. the effect of the reactive impedance needs to be eliminated. Knowing the current-voltage relationship of the output end, according to the ABCD transmission matrix, the following equation is obtained:

to obtain an output admittance of the transistor

Wherein G is₁、B₁Respectively conductance and susceptance; r_LIn order to be the characteristic impedance of the transmission line,are of unequal electrical length. Eliminating the effect of reactive load so that B₁When the value is equal to 0, and in order to eliminate reactive impedance, a backoff value OBO is set artificially as a design target, a value theta corresponding to the backoff value can be obtained by using a formula (8) and then substituted into a formula (7) to obtain a formula (9);

OBO＝20·log(sinθ).

formula (8)

Wherein t ═ R_L/R_OOBO is a backspacing range, and the final product is obtained

And R_L、R_OThe relationship between them. Due to constant amplitude of input signalThe heterodromous relationship, therefore, the modulation signal for constant amplitude + theta input is required

Compensation of (2) and, similarly, the need for a modulation signal of constant amplitude-theta inputIs compensated for, finally formed

And

two transmission lines of unequal length.

Under the above single frequency beltAnd

the transmission line is replaced with the following hybrid T/Pi type structure.

The mixed T/Pi type structure comprises a first T microstrip line, a second microstrip line and a third T microstrip line; the first T microstrip line can be further divided into three parts, one end of the first part of the first T microstrip line is connected with the drain electrode of the transistor and one end of the second T microstrip line, the other end of the first part is connected with one end of the second part and one end of the third part, the other end of the second part is suspended, and the other section of the third part is grounded or suspended. First part implementation f₁Imaginary part compensation at frequency, second part elimination f₂Imaginary part compensation pair f₁Influence of imaginary part compensation, third part implementing f₂Imaginary compensation at frequency; the straight line of the first part and the third part is perpendicular to the second part, so that a T-shaped structure is formed. The third T microstrip line has the same structure and function as the first T microstrip line. The other end of the second microstrip line is connected with one end of the first part of the third T microstrip line to be used as a B port. The ports B of the two paths of second microstrip lines are connected to a combining point together, so that the purpose of combining and restoring signals is achieved. Three microThe strip lines together form a mixed T/Pi type structure to realize the requirement of dual-band different impedance transformation.

The double-band unequal-length power synthesis circuit with the mixed T/Pi type structure is obtained by calculation and design by using the method. And then designing an output matching circuit between the transistor and the dual-band unequal-length power synthesis circuit to achieve the maximum output power. The purpose of output matching is to match the complex impedance to the real impedance presented by the dual-band non-equal length power combining circuit. The output complex impedance presented by the transistor is mainly due to the influence of parasitic capacitance. Therefore, the invention compensates the parasitic capacitance-C at the output end of the transistor by parallel connection_dsTo make the transistor appear as a real impedance. Due to-C_dsIs a virtual circuit part, and the final circuit is realized by compensating the capacitor-C_dsThe power combining circuit is integrated into the designed dual-band unequal-length output power combining circuit. Finally, a new dual-band power synthesis circuit is formed.

Another object of the present invention is to provide a method for designing a dual-band differential power amplifier based on unequal length transmission lines, which specifically comprises:

the method comprises the following steps: the modulation signal to be transmitted is subjected to angle modulation at the PC end, a digital-to-analog conversion chip, a low-pass filter, a frequency mixer and a self-adaptive gain amplifier to obtain two paths of modulation signals to be input with equal amplitude and different directions. The digital-to-analog conversion chip converts the digital baseband signal into an analog modulation signal; the low-pass filter is used for eliminating clutter components of the baseband signal; the quadrature mixer is used for modulating the baseband signal to a carrier frequency; the adaptive gain controller is used for adjusting the amplitude of the radio frequency modulation signal to a proper size to be input into the back-end amplifier.

Step two: according to design indexes, two identical transistors are selected, and then corresponding input impedance and output impedance are determined.

Step three: according to input impedance, an input matching circuit is designed by adopting a high-low impedance (step-type broadband) matching method, specifically, a low-pass filter prototype, a series inductor and a parallel capacitor are alternatively used, and the numerical values of the series inductor and the parallel capacitor are determined by a Chebyshev filter design method. Since lumped elements such as inductors and capacitors are suitable for a low frequency environment, and microstrip lines are more suitable for a high frequency environment, the lumped elements need to be converted into microstrip lines. The series inductance is approximately equal to the series high-impedance microstrip line, and the shunt capacitance is approximately equal to the series low-impedance microstrip line. Finally, high-low impedance alternative step type broadband matching is formed, and the aim of double-band input matching is achieved.

Step four: debugging a bias circuit:

two ends of a 90-degree electric length transmission line are respectively connected with a power supply and a transistor grid to eliminate the influence of second harmonic.

Step five: debugging the power supply circuit:

for the second harmonic wave under the frequency of f1, one end of a 90-degree impedance transformation line under the frequency of f1 is connected with a direct-current power supply, and the other end of the impedance transformation line is connected with the drain electrode of a transistor, so that the influence of the second harmonic wave of the frequency of f1 is eliminated; for the second harmonic wave at the frequency of f2, a 90-degree impedance transformation line mode at the frequency of f2 is adopted, wherein one end of the 90-degree impedance transformation line is grounded through a direct-current blocking capacitor, and the other end of the 90-degree impedance transformation line is connected with the drain of the transistor.

Step six: and debugging the single-band unequal-length transmission lines under two independent frequency bands according to the output impedance:

the two paths of output currents are respectively as follows:

wherein i₅And i₆Respectively output current v for upper and lower circuits_opepFor maximum output voltage, theta is at a compensation angle, R, varying from 0 to 90 degrees_oThe load impedance is generally set to 50 ohms. j denotes the imaginary sign. In order to ensure high efficiency of the non-inverting transistor, the output impedance seen at the output of the transistor needs to be purely real, i.e. the effect of the reactive impedance needs to be eliminated. Knowing the current-voltage relationship at the output, according to the ABCD transmission matrix,the following equation is obtained:

to obtain an output admittance of the transistor

Wherein G is₁、B₁Respectively conductance and susceptance; r_LIn order to be the characteristic impedance of the transmission line,

are of unequal electrical length. Eliminating the effect of reactive load so that B₁And is equal to 0, and in order to eliminate reactive impedance, a backoff value OBO (a conventional backoff value OBO can be set to 6dB) is set artificially as a design target, and an angle θ corresponding to the backoff range OBO is obtained according to a formula (8). Then, the formula (7) is used to obtain the formula (9) compensation electrical length

And R_L、R_O(t＝R_L/R_O) The relation between the two signals can obtain the transmission line values with different lengths in the respective single frequency bands.

OBO＝20·log(sinθ).

Formula (8)

Step seven: compensating the electrical length obtained according to the sixth step

And R_L、R_OThe relation between the two microstrip lines is that the single-strip unequal length transmission line is equivalently replaced by a mixed T/Pi dual-strip unequal length power synthesis circuit, wherein the mixed T/Pi dual-strip unequal length power synthesis circuit structure comprises a first T microstrip line, a second microstrip line and a third T microstrip line; the first T microstrip line can be further divided into three parts, one end of the first part of the first T microstrip line is connected with the drain electrode of the transistor and one end of the second T microstrip line, the other end of the first part is connected with one end of the second part and one end of the third part, the other end of the second part is suspended, and the other section of the third part is grounded or suspended. First part implementation f₁Imaginary part compensation at frequency, second part elimination f₂Imaginary part compensation pair f₁Influence of imaginary part compensation, third part implementing f₂Imaginary compensation at frequency; the straight line of the first part and the third part is perpendicular to the second part, so that a T-shaped structure is formed. The third T microstrip line has the same structure and function as the first T microstrip line. The other end of the second microstrip line is connected with one end of the first part of the third T microstrip line to be used as a B port. The ports B of the two paths of second microstrip lines are connected to a combining point together, so that the purpose of combining and restoring signals is achieved.

Step eight: an output matching circuit is designed, and the purpose of output matching is to match the complex impedance to the real impedance presented by the dual-band unequal length power synthesis circuit. The output complex impedance presented by the transistor is mainly due to the influence of parasitic capacitance. Therefore, the invention compensates the parasitic capacitance-C at the output end of the transistor by parallel connection_dsTo make the transistor appear as a real impedance. Due to-C_dsIs a virtual circuit part, and the final circuit is realized by compensating the capacitor-C_dsAnd integrating the power combination circuit into the designed dual-band unequal-length output power combination circuit to finally form a new dual-band power combination circuit.

Step nine: and combining the input matching circuit, the bias circuit, the power supply circuit and the new double-band power synthesis circuit which are debugged in the steps to form one path of double-frequency-band high-efficiency different-direction power amplification circuit. The two paths of double-frequency-band high-efficiency anisotropic power amplifying circuits respectively receive the two paths of modulated signals with equal amplitude and different directions after the first step of processing.

The invention has the following effective effects: on the basis of a classic Chireix outphasing amplifier, the detailed theory of impedance matching, power combination and imaginary part compensation by using unequal-length transmission lines is perfected. Under the guidance of the theory, a double-band unequal-length power synthesis circuit is used for replacing an optimal value under a single band, and a key factor compensation parasitic capacitance-Cds of an output matching circuit is integrated into the synthesis circuit, so that a new double-band power synthesis circuit combined with the parasitic capacitance and the whole double-band high-efficiency outphasing amplifier circuit are finally obtained.

Drawings

Fig. 1 is a schematic structural diagram of a dual-band different-direction power amplifier based on unequal length transmission lines according to the present invention.

Fig. 2 is a schematic diagram of the output terminal of the single-band unidirectional power amplifier of the present invention.

FIG. 3 is a schematic diagram of a hybrid T/Pi circuit structure of a dual-band unequal-length power combining circuit according to the present invention.

FIG. 4 is a diagram showing the simulation results of the dual-band simulation of the present invention using ADS software, wherein (a) is 2.6GHz and (b) is 3.5 GHz.

Detailed Description

The following are specific embodiments of the present invention and are further described with reference to the drawings, but the present invention is not limited to these embodiments.

Aiming at the blank and the defects in the field of dual-band anisotropic power amplifiers, the invention carries out deep research on the structure of a classical anisotropic single-frequency band, finds that a power synthesis circuit can be realized by using unequal-length transmission lines under the single-frequency band, not only can the characteristics of the classical heterogeneous amplifier be effectively maintained, but also the dependence on a reactance compensation network can be effectively eliminated, thereby reducing the complexity of circuit design.

Fig. 1 is a block diagram of a dual-band high-efficiency anisotropic power amplifier according to the present invention, which includes a signal conditioning circuit and a microwave power amplifying circuit. The signal conditioning circuit comprises a baseband signal generation circuit, a DAC (digital-to-analog conversion chip), a filter, a mixer and a self-adaptive gain amplifier, and the structures are all known technologies. The microwave power amplifying circuit comprises an input matching circuit and an output matching circuit which are used for ensuring low-loss transmission of signals. The microwave power amplifier is a well-known power amplifier and has a load impedance of 50 ohms.

After the signal conditioning circuit is designed, the second step needs to determine the input and output resistances of the transistors for matching later. The specific method comprises the following steps: according to the transistor large-signal model provided by the trade company, ADS software is used for load traction and source traction, and the optimal input and output impedances of 3+ j × 5 and 15+ j10 are obtained.

And thirdly, using the optimal input impedance obtained in the second step as an input matching circuit, wherein the input matching circuit realizes double-band matching by using a step type broadband matching circuit formed by high impedance and low impedance, and the specific method is realized by using known matching technologies such as Chebyshev and the like. The methods such as chebyshev are known.

And designing a bias circuit, wherein the bias circuit firstly provides a proper bias voltage for the transistor to work in a proper amplification state, the CGH40010 transistor of the cree company is designed and biased in class AB, the bias voltage is set to be-2.7V, a 90-degree microstrip line is selected for connection of the bias voltage and the grid, and the effect of the bias circuit is to eliminate the influence of second harmonic.

And a fifth step of debugging a power supply circuit, wherein the core of the power supply circuit is used for supplying electric energy to the transistor. According to a data manual, a power supply voltage is set to be 28V, for the second harmonic wave under the frequency of 2.6GHz, one end of a 90-degree impedance conversion line under the frequency of 2.6GHz is connected with a direct-current power supply, and the other end of the 90-degree impedance conversion line is connected with a drain electrode of a transistor, so that the influence of the second harmonic wave under the frequency of 2.6GHz is eliminated; for the second harmonic wave under the frequency of 3.5GHz, a mode of connecting 90-degree impedance transformation lines under the frequency of 3.5GHz in parallel is adopted, wherein one end of the 90-degree impedance transformation line is grounded through a direct current blocking capacitor, and the other end of the 90-degree impedance transformation line is connected with the drain electrode of the transistor.

And sixthly, determining the optimal values of the circuits of each part under the 2.6GHz and the 3.5GHz frequency bands of two independent frequency bands. The method comprises the following steps:

the schematic diagram of the power combining circuit with unequal length at the output end is shown in fig. 2, and it is assumed that the total output voltage is v_oAnd the output voltage varies with the angle. The change relationship among the two is as follows:

v_o＝v_opepcos (θ) equation (1)

Wherein v is_opepθ varies from 0 to 90 degrees for maximum output voltage. The output voltage is maximum when the angle is 0 degrees, and 0 when the angle is 90 degrees. Because the upper path and the lower path are strictly symmetrical, the two paths of output currents respectively are as follows:

compensated electrical length obtained according to the above summary

And R_L、R_OThe relation between the voltage and the current is then used to obtain the input impedance Z before and after adding the compensation line₃、Z₁. The complex impedance Z is due to the phase difference between the current and the voltage₃、Z₁Is represented by R₃+jx₃And R₁+jx₁Similarly, the input impedance Z before and after the compensation line is added to the other power amplifier₄、Z₂Is represented by R₄+jx₄And R₂+jx₂. The optimal electrical length and impedance of each part for realizing the function of the high-efficiency power amplifier in the single frequency band can be obtained according to the formula (9). The characteristic impedance and the compensation electrical length at 2.6GHz single frequency band are (56 ohm, 42.14 degree), the characteristic impedance and the compensation electrical length at 3.5GHz single frequency band are (46 ohm, 56.95 degree)

And the seventh step is to replace the lower circuit part of the single frequency band with a mixed T/Pi type combined structure double-band unequal length power synthesis circuit. To implement a dual band hetero power amplifier. When the optimal values of each part of the single band are obtained, a dual-band unequal-length transmission line capable of meeting different electrical lengths and characteristic impedance under a dual-frequency band needs to be designed.

In a preferred embodiment, the present invention utilizes a hybrid T/Pi type configuration to achieve the above described functionality. The main body structure of the hybrid T/Pi type structure shown in FIG. 3 is the PI type. The branches connected in parallel at the two ends need to provide different susceptances under the double frequency bands.

cos(θ_T)＝cosθ_S-B_SR_Ssinθ_S

In the above formula, R_S、θ_SCharacteristic impedance and electrical length, R, of the second microstrip line, respectively_TFor optimum characteristic impedance of the transmission line at a single frequency, R_T1、θ_T1Respectively, a target optimum characteristic impedance and an electrical length, R, at a first frequency point_T2、θ_T2Respectively the target optimum characteristic impedance and the electrical length at the second frequency point, B_SFor the susceptance of the parallel branches, different input susceptances are required under the double frequency bands.

For this purpose, the parallel branches of the original Pi-type section need to satisfy the dual-band different input admittance. The T-shaped section is adopted to meet the requirements, and the specific requirements are as follows: first of all satisfy f₁Frequency imaginary part compensation and then f₂Compensating for frequency band to prevent f₂Imaginary part compensation pair f₁Influence of imaginary part compensation, at f₁The imaginary part compensation end is realized by adding a parallel 90-degree open-circuit branch. The three segments are together f₂Providing imaginary part compensation, and specifically comprising the following steps:

the dual-band imaginary part compensation circuit needs to implement different compensation in dual bands. As shown in fig. 3, the imaginary part compensation circuit is composed of three microstrip lines in total, and the first part realizes the pair f₁Imaginary compensation at frequency.

The second part is at f₁The end of imaginary part compensation is added at f₁Open-circuit microstrip line with an electrical length of 90 degrees at frequency, with the aim of making the first section pair f at the point of parallel connection₁The imaginary part of the frequency compensates the short circuit so as to add the microstrip line again and not to f₁The frequency compensation has an effect. Then adding a third part and combining the first part and the second part to form a pair f₂And (4) compensating the frequency.

Y_B(f₂)＝Y_A(f₂)-Y_C(f₂)

θ_B3＝tan^-1(R_B3img(Y_B(f₂) ) for open branches

Wherein theta is_B1,θ_B2,θ_B3The electrical lengths of the first part, the second part and the third part of the microstrip line are respectively; r_B1,R_B2，R_B3Characteristic impedances of the microstrip lines of the first part, the second part and the third part respectively; y is_A(f₂) Incorporating the second and third microstrip lines with a view into the input admittance, Y_B(f₂) Input admittance, Y, into which the third part of the microstrip line is looking_c(f₂) A second portion of the input admittance into which the microstrip line is to be viewed.

Wherein, the dual-band unequal length power synthesis circuit is realized by a mixed T/Pi type section to obtain R_SAnd theta_S44.13 ohms and 109.43 degrees.

The parallel branch node is realized by T-shaped node, wherein f₁Forward compensation or mapping

At this time correspond to f₂Negative compensation is corresponding to

At this time, 80 ohms are selected for the characteristic impedance of the first part, the second part and the third part of the input sub-amplifier circuit, and the corresponding electrical lengths are 10 degrees, 90 degrees and 38 degrees respectively at 3.5 GHz. Also, wherein f₁Negative compensation is corresponding to

At this time correspond to f₂Forward compensation or mapping

The characteristic impedance of the first part, the second part and the third part of the input sub-amplifier circuit is selected to be 80 ohms, and the corresponding electrical length is respectively 27 degrees, 90 degrees and 32 degrees under 2.6 GHz.

Step eight: an output matching circuit is designed to match the impedance presented by the hybrid T/Pi combination structure to the output complex impedance of the strip transistor. To further simplify the circuit structure, the output matching purpose is to match the complex impedance to the real impedance presented by the dual-band non-equal length power combining circuit. The output complex impedance presented by the transistor is mainly due to the influence of parasitic capacitance. The idea here is therefore to make the transistor appear as a real impedance at the output of the transistor by compensating the parasitic capacitance-Cds in parallel. Of course-Cds is a virtual circuit part, and the final circuit implementation method is that we integrate the compensation capacitor-Cds into the back-end power combining circuit. The parasitic capacitances Cds are 1.7pF and 2.5pF at 2.6GHz and 3.5GHz, respectively. And then the Cds is fused to a rear-end double-band unequal-length power synthesis circuit to form a double-band new power synthesis circuit.

Step nine: and combining the designed circuit parts together, and designing simulation optimization by using ADS software to obtain the finally designed dual-band high-efficiency anisotropic power amplifier.

Aiming at the blank and the defects of the existing dual-band anisotropic power amplifier, the invention respectively uses a Chebyshev structure and a mixed T/Pi type dual-band structure by inputting a matching network, a bias network and a dual-band unequal-length power synthesis circuit under two single frequency bands, thereby realizing the function of high-efficiency anisotropic power amplification under the dual frequency bands.

Fig. 4(a) (b) is a data diagram simulated by ADS software for two independent frequencies of 2.6GHz and 3.5GHz based on the method of the present invention. According to simulation results, the saturation efficiency can reach more than 70% under the frequencies of 2.6GHz and 3.5 GHz. The 6dB back-off efficiency is 68% and 59%. The result shows that the function of the dual-band high-efficiency anisotropic power amplifier based on the unequal length transmission lines is realized.

The above description of the embodiments is only intended to facilitate the understanding of the method of the invention and its core idea. It should be noted that, for those skilled in the art, it is possible to make various improvements and modifications to the present invention without departing from the principle of the present invention, and those improvements and modifications also fall within the scope of the claims of the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.