Method and apparatus for transmitting OFDM signal and method and apparatus for receiving OFDM signal
阅读说明：本技术 用于发送ofdm信号的方法及装置和用于接收ofdm信号的方法及装置 (Method and apparatus for transmitting OFDM signal and method and apparatus for receiving OFDM signal ) 是由 尹硕铉 高贤秀 金沂濬 金炳勋 朴昶焕 于 2018-12-04 设计创作，主要内容包括：一种在无线通信系统中由发送装置发送正交频分复用(OFDM)信号的方法,该方法包括：由发送装置的数字模块通过对第一信号执行载波频率f<Sub>0</Sub>与第一频率f<Sub>base</Sub>之差的频率上移,来生成频移的OFDM基带信号,其中,第一频率f<Sub>base</Sub>是与128Δf的整数倍相对应的频率当中最接近载波频率f<Sub>0</Sub>的,并且其中Δf是OFDM子载波间隔；由发送装置的模拟振荡器对频移的OFDM基带信号进行第一频率f<Sub>base</Sub>的上转换,以生成载波频率f<Sub>0</Sub>的OFDM符号信号；以及由发送装置的发送器发送载波频率f<Sub>0</Sub>的OFDM符号信号。(A method of transmitting an Orthogonal Frequency Division Multiplexing (OFDM) signal by a transmitting apparatus in a wireless communication system, the method comprising: by the digital module of the transmitting device performing the carrier frequency f on the first signal 0 And a first frequency f base The difference is shifted up in frequency to generate a frequency shifted OFDM baseband signal, wherein the first frequency f base Is the closest carrier frequency f among frequencies corresponding to integer multiples of 128 deltaf 0 And wherein Δ f is the OFDM subcarrier spacing; subjecting the frequency-shifted OFDM baseband signal to a first frequency f by an analog oscillator of a transmitting device base To generate a carrier frequency f 0 The OFDM symbol signal of (a); and transmitting carrier frequency f by transmitter of transmitting device 0 The OFDM symbol signal of (1).)
1. A method for transmitting an orthogonal frequency division multiplexing, OFDM, signal by a transmitting apparatus in a wireless communication system, the method comprising the steps of:
by the digital module of the transmitting device performing the carrier frequency f on the first signal0And a first frequency fbaseThe difference is frequency-shifted up to generate a frequency-shifted OFDM baseband signal, wherein the first frequency fbaseIs closest to the carrier frequency f among frequencies corresponding to integer multiples of 128 deltaf0Where Δ f is the OFDM subcarrier spacing;
subjecting the frequency-shifted OFDM baseband signal to the first frequency f by an analog oscillator of the transmitting devicebaseTo generate a carrier wave at said carrier frequency f0The OFDM symbol signal of (a); and
transmitting at the carrier frequency f by a transmitter of the transmitting device0The OFDM symbol signal of (a).
2. The method of claim 1, wherein the digital module is configured to perform an inverse fast fourier transform, IFFT, on the first signal.
3. The method of claim 2, wherein f is performed on the first signal0And fbaseSaid frequency up-shifting of the difference comprises the steps of:
up-shifting resource mapping of the first signal input to the IFFT by NfracIn which N isfracIs to satisfy f0-fbase＝NfracΔ f.
4. The method of claim 1, wherein the digital module comprises a digital oscillator, and
wherein f is performed on the first signal0And fbaseThe frequency up-shifting of the difference is performed by the digital oscillator.
5. The method of claim 4, further comprising the steps of:
resetting, by the digital oscillator and prior to transmitting the OFDM symbol signal, a phase of the OFDM symbol signal to a predetermined value at an end of a cyclic prefix of the OFDM symbol signal.
6. A method of receiving an orthogonal frequency division multiplexing, OFDM, signal by a receiving apparatus in a wireless communication system, the method comprising the steps of:
receiving at carrier frequency f0The OFDM symbol signal of (a);
subjecting the OFDM symbol signal to a first frequency f by an analog oscillator of the receiving devicebaseTo generate a down-converted OFDM symbol signal; and
by the digital module of the receiving device performing the carrier frequency f on the down-converted OFDM symbol signal0And fbaseThe difference is shifted down in frequency to generate an OFDM baseband signal,
wherein the first frequency fbaseIs closest to the carrier frequency f among frequencies corresponding to integer multiples of 128 deltaf0Where Δ f is the OFDM subcarrier spacing.
7. The method of claim 6, wherein the digital module is configured to perform a fast fourier transform, FFT, on the down-converted OFDM symbol signal.
8. The method of claim 7, wherein said f is performed on said down-converted OFDM symbol signal0And fbaseThe frequency downshifting of the difference comprises the steps of:
downshifting resource demapping from FFT used for the downconverted OFDM symbol signal by NfracWherein N isfracIs to satisfy f0-fbase＝NfracΔ f.
9. The method of claim 6, wherein the digital module comprises a digital oscillator, and
wherein the f is performed on the down-converted OFDM symbol signal0And fbaseThe frequency downshifting of the difference is performed by the digital oscillator.
10. The method of claim 9, further comprising the steps of:
resetting, by the digital oscillator, a phase of the down-converted OFDM symbol signal to a predetermined value at an end of a cyclic prefix of the down-converted OFDM symbol signal.
11. A transmission apparatus for transmitting an orthogonal frequency division multiplexing, OFDM, signal in a wireless communication system, the transmission apparatus comprising:
a digital module;
an analog oscillator;
at least one antenna;
at least one processor; and
at least one computer memory operatively connectable to the at least one processor and having instructions stored thereon that, when executed, cause the at least one processor to perform operations comprising:
by the digital module by performing the carrier frequency f on the first signal0And a first frequency fbaseThe difference is frequency-shifted up to generate a frequency-shifted OFDM baseband signal, wherein the first frequency fbaseIs closest to the carrier among frequencies corresponding to integer multiples of 128 afFrequency f0Where Δ f is the OFDM subcarrier spacing;
subjecting the frequency-shifted OFDM baseband signal to the first frequency f by the analog oscillatorbaseTo generate a carrier wave at said carrier frequency f0The OFDM symbol signal of (a); and
transmitting at the carrier frequency f using the at least one antenna0The OFDM symbol signal of (a).
12. The transmission apparatus of claim 11, wherein the digital module is configured to perform an inverse fast fourier transform, IFFT, on the first signal.
13. The transmission apparatus of claim 12, wherein the f is performed on the first signal0And fbaseThe frequency up-shifting of the difference comprises:
up-shifting resource mapping of the first signal input to the IFFT by NfracIn which N isfracIs to satisfy f0-fbase＝NfracΔ f.
14. The transmitting apparatus of claim 11, wherein the digital module comprises a digital oscillator, and
wherein the f performed on the first signal0And fbaseThe frequency up-shifting of the difference is performed by the digital oscillator.
15. The transmitting apparatus of claim 14, wherein the operations further comprise:
resetting, by the digital oscillator and prior to transmitting the OFDM symbol signal, a phase of the OFDM symbol signal to a predetermined value at an end of a cyclic prefix of the OFDM symbol signal.
16. A receiving apparatus for receiving an orthogonal frequency division multiplexing, OFDM, signal in a wireless communication system, the receiving apparatus comprising:
at least one antenna;
an analog oscillator;
a digital module;
at least one processor; and
at least one computer memory operatively connectable to the at least one processor and having instructions stored thereon that, when executed, cause the at least one processor to perform operations comprising:
receiving at a carrier frequency f using the at least one antenna0The OFDM symbol signal of (a);
subjecting the OFDM symbol signal to a first frequency f by the analog oscillatorbaseTo generate a down-converted OFDM symbol signal; and
performing the f by the digital module on the down-converted OFDM symbol signal0And fbaseThe difference is shifted down in frequency to generate an OFDM baseband signal,
wherein the first frequency fbaseIs closest to the carrier frequency f among frequencies corresponding to integer multiples of 128 deltaf0Where Δ f is the OFDM subcarrier spacing.
17. The receiving apparatus according to claim 16, wherein the digital module is configured to perform a fast fourier transform, FFT, on the down-converted OFDM symbol signal.
18. The receiving apparatus as claimed in claim 17, wherein the f is performed on the down-converted OFDM symbol signal0And fbaseThe frequency downshifting of the difference comprises:
downshifting resource demapping from FFT used for the downconverted OFDM symbol signal by NfracWherein N isfracIs to satisfy f0-fbase＝NfracΔ f.
19. The receiving device of claim 16, wherein the digital module comprises a digital oscillator, and
wherein the f is performed on the down-converted OFDM symbol signal0And fbaseThe frequency downshifting of the difference is performed by the digital oscillator.
20. The receiving device of claim 19, wherein the operations further comprise:
resetting, by the digital oscillator, a phase of the down-converted OFDM symbol signal to a predetermined value at an end of a cyclic prefix of the down-converted OFDM symbol signal.
Technical Field
The present disclosure relates to a wireless communication system. More particularly, the present disclosure relates to a method and apparatus for transmitting an OFDM signal, and a method and apparatus for receiving an OFDM signal.
Background
In a mobile communication system, a transmitting apparatus generally generates a baseband signal, up-converts the baseband signal to a higher carrier frequency, and transmits the up-converted radio signal at the carrier frequency. The receiving device then receives the radio signal and downconverts the received radio signal from the carrier frequency to a lower baseband frequency to obtain a baseband signal for further processing.
Disclosure of Invention
Technical problem
If information on the frequency used for the up-conversion is unknown to the transmitting apparatus and the receiving apparatus, a mismatch may occur between the up-conversion frequency used by the transmitting apparatus and the down-conversion frequency used by the receiving apparatus. A mismatch between the up-conversion frequency and the down-conversion frequency causes a sudden phase change per time symbol in the receiving device. Such sudden phase change greatly degrades the performance of signal recovery through channel estimation in the receiving apparatus. Therefore, a method for reducing a phase change of each symbol caused by a mismatch between an up-conversion frequency and a down-conversion frequency, a mismatch between a carrier frequency and a center frequency of a frequency band, or a mismatch between a carrier frequency and a center of an RF filter is required.
In addition, when the carrier frequency changes in the same frequency band, there is a need for a method of easily adjusting the carrier frequency without RF retuning.
Technical scheme
The object of the present disclosure can be achieved by the technology disclosed herein for transmitting an Orthogonal Frequency Division Multiplexing (OFDM) signal by a transmitting apparatus in a wireless communication system. In one aspect, a method of transmitting an Orthogonal Frequency Division Multiplexing (OFDM) signal by a transmitting apparatus in a wireless communication system is provided herein. The method comprises the following steps: by the digital module of the transmitting device performing the carrier frequency f on the first signal0And a first frequency fbaseThe difference is shifted up in frequency to generate a frequency shifted OFDM baseband signal; subjecting the frequency-shifted OFDM baseband signal to a first frequency f by an analog oscillator of a transmitting devicebaseTo generate a carrier frequency f0The OFDM symbol signal of (a); and at the carrier frequency f by the transmitter of the transmitting device0An OFDM symbol signal is transmitted. First frequency fbaseMay be the closest carrier frequency f among frequencies corresponding to integer multiples of 128 Δ f0In (1). Δ f is the OFDM subcarrier spacing.
In another aspect, provided herein is a method of receiving an Orthogonal Frequency Division Multiplexing (OFDM) signal by a device of a reception side in a wireless communication system.The method comprises the following steps: receiving carrier frequency f0The OFDM symbol signal of (a); subjecting an OFDM symbol signal to a first frequency f by an analog oscillator of the devicebaseTo generate a down-converted OFDM symbol signal; and performing carrier frequency f by digital module of the device on the down-converted OFDM symbol signal0And fbaseThe frequency of the difference is shifted down to generate an OFDM baseband signal. First frequency fbaseMay be the closest carrier frequency f among frequencies corresponding to integer multiples of 128 Δ f0In (1). Δ f is the OFDM subcarrier spacing.
In another aspect, provided herein is an apparatus at a transmitting side for transmitting an Orthogonal Frequency Division Multiplexing (OFDM) signal in a wireless communication system, the apparatus may include: a digital module; an analog oscillator; at least one antenna; at least one processor; and at least one computer memory operatively connected to the at least one processor and having instructions stored thereon that, when executed, cause the at least one processor to perform operations. The operations may include: by the digital module performing the carrier frequency f on the first signal0And a first frequency fbaseShifting up the frequency of the difference to generate a frequency shifted OFDM baseband signal; subjecting the frequency-shifted OFDM baseband signal to a first frequency f by an analog oscillatorbaseTo generate a carrier frequency f0The OFDM symbol signal of (a); and using at least one antenna at a carrier frequency f0An OFDM symbol signal is transmitted. First frequency fbaseMay be the closest carrier frequency f among frequencies corresponding to integer multiples of 128 Δ f0In (1). Δ f is the OFDM subcarrier spacing.
In yet another aspect, an apparatus at a receiving side for receiving an Orthogonal Frequency Division Multiplexing (OFDM) signal in a wireless communication system is provided herein. The apparatus may include: at least one antenna; an analog oscillator; a digital module; at least one processor; and at least one computer memory operatively connected to the at least one processor and having instructions stored thereon that, when executed, cause the at least one processor to perform operations. The operations may include: receiving a carrier frequency using at least one antennaf0An OFDM symbol signal; subjecting an OFDM symbol signal to a first frequency f by an analog oscillatorbaseTo generate a down-converted OFDM symbol signal; and by the digital module by performing the carrier frequency f on the down-converted OFDM symbol signal0And fbaseThe difference is shifted down in frequency to generate an OFDM baseband signal. First frequency fbaseMay be the closest carrier frequency f among frequencies corresponding to integer multiples of 128 Δ f0In (1). Δ f is the OFDM subcarrier spacing.
In each aspect of the transmitting side, the digital module may be configured to perform an Inverse Fast Fourier Transform (IFFT) on the first signal.
In each aspect of the transmitting side, f is performed on the first signal0And fbaseThe frequency up-shifting of the difference may comprise: upshifting a resource mapping of a first signal input to an IFFT by NfracIn which N isfracIs to satisfy f0-fbase＝NfracΔ f.
In each aspect of the transmitting side, the digital module may include a digital oscillator. Performing f on the first signal0And fbaseThe frequency up-shifting of the difference may be performed by a digital oscillator.
In each aspect of the transmitting side, the digital oscillator may reset the phase of the OFDM symbol signal to a predetermined value at the end of a cyclic prefix of the OFDM symbol signal before transmitting the OFDM symbol signal.
In each aspect of the receiving side, the digital module may be configured to implement a Fast Fourier Transformer (FFT) of the down-converted OFDM symbol signal.
In each aspect on the receiving side, f is performed on the down-converted OFDM symbol signal0And fbaseThe frequency downshifting of the difference may include: demapping resources from FFT of OFDM symbol signal for down-conversion by NfracWherein N isfracIs to satisfy f0-fbase＝NfracΔ f.
In each aspect of the receiving side, the digital module may include a digital oscillator. Performing on a down-converted OFDM symbol signalf0And fbaseThe frequency downshifting of the difference may be performed by a digital oscillator.
In each aspect of the receiving side, the digital oscillator may reset the phase of the down-converted OFDM symbol signal to a predetermined value at the end of the cyclic prefix of the down-converted OFDM symbol signal.
The above technical solutions are only some parts of the implementation of the present disclosure, and those skilled in the art can derive and understand various embodiments in which technical features of the present disclosure are incorporated from the following detailed description of the present disclosure.
Technical effects
According to the present invention, it is possible to easily minimize a phase variation according to a symbol occurring due to a mismatch between an up-conversion frequency and a down-conversion frequency. Therefore, a signal occurs even if the up-conversion frequency is unknown to the transmitting apparatus and the receiving apparatus, or there is no match between the up-conversion/down-conversion frequency and the center of the RF filter, or there is no match between the carrier frequency and the center frequency of the cell. The recovery performance of the receiving apparatus can be maintained.
In addition, when the carrier frequency changes in the same frequency band, the carrier frequency can be easily adjusted without RF retuning.
Drawings
Fig. 1A and 1B illustrate examples of modulation and up-conversion of a baseband signal to a carrier frequency;
fig. 2A and 2B are diagrams illustrating an example of a phase change according to a difference between an up-conversion frequency and a down-conversion frequency;
fig. 3 illustrates an example of resetting a phase at a symbol boundary;
fig. 4A and 4B illustrate examples of generation and modulation and upconversion of a baseband signal to its carrier frequency in accordance with some implementations of the present disclosure;
fig. 5A to 5C are diagrams illustrating an example of
FIGS. 6A and 6B are diagrams illustrating an example of an implementation 2-1 of the present disclosure;
FIGS. 7A and 7B are diagrams illustrating examples of resource mappings according to an implementation 2-1 of the present disclosure and resource mappings according to an implementation 2-2 of the present disclosure.
FIGS. 8A and 8B are diagrams illustrating an example of an implementation 2-2 of the present disclosure;
fig. 9A to 9C are diagrams illustrating an example of implementation 3 of the present disclosure;
10A and 10B are diagrams illustrating an example of implementation a2-1 of the present disclosure;
11A and 11B are diagrams illustrating an example of implementation a2-2 of the present disclosure;
fig. 12 is a diagram illustrating another use example of the present invention.
13A and 13B illustrate examples of transmitter structures and receiver structures according to some implementations of the present disclosure; and
fig. 14 is a block diagram illustrating examples of components of a transmitting device and a receiving device in accordance with some implementations of the present disclosure.
Detailed Description
Wireless communication systems typically communicate using a specific range of Radio Frequencies (RF). To ensure proper transmission in these RF ranges, wireless systems typically implement a technique called up-conversion at the transmitter to convert signals from a lower frequency range to a higher (RF) frequency range, and a technique called down-conversion at the receiver to convert signals from a higher (RF) frequency range to a lower frequency range.
However, difficulties may arise where information regarding frequency conversion is unknown to the transmitting device and/or the receiving device. In such a scenario, a mismatch may occur between the up-conversion frequency used by the transmitting device and the down-conversion frequency used by the receiving device. This mismatch between the up-conversion frequency and the down-conversion frequency may cause a phase offset in each time symbol received at the receiving device. The phase offset may degrade the performance of recovering the signal through channel estimation in the receiving apparatus.
Furthermore, in some scenarios, a mismatch may occur between the carrier frequency and the center frequency of the frequency band or between the carrier frequency and the center of the RF filter. Such a mismatch may also result in a phase offset in the received time symbol, which may degrade reception performance.
Thus, difficulties may arise in systems where such phase shifts occur due to a mismatch between the up-conversion frequency and the down-conversion frequency, or due to a mismatch between the carrier frequency and the center frequency of the frequency band, or due to a mismatch between the carrier frequency and the center of the RF filter. In addition, when the carrier frequency is changed in the same frequency band, difficulty may occur in adjusting the carrier frequency without performing RF retuning.
Implementations disclosed herein enable a transmitter to be configured to perform up-conversion in a manner that mitigates or removes such phase offsets. In some implementations, the transmitter up-converts from baseband to one of a predetermined limited number of frequencies, and is therefore configured not to cause a phase offset at the receiver. Since these limited number of frequencies may differ from the actual carrier frequency used by the transmitter, the transmitter may compensate for this difference by pre-shifting the baseband signal by any such difference.
In some implementations, the pre-shifting may be accomplished by performing a frequency domain shift (e.g., shifting the input of an Inverse Fast Fourier Transform (IFFT) at the transmitter) or may be accomplished by a time domain shift (e.g., shifting the output of an IFFT using, for example, a digital oscillator).
Similarly, in some implementations, the receiver is configured to perform a down-conversion from one of a limited number of predetermined frequencies down to baseband. Also, since a limited number of frequencies may differ from the actual carrier frequency used by the receiver, the receiver may compensate for this difference by post-offsetting the resulting baseband signal by any such difference.
Thus, implementations disclosed herein may mitigate or remove phase offsets that occur due to mismatches between the up-conversion frequency and the down-conversion frequency. Accordingly, even if the up-conversion frequency is unknown to the transmitting apparatus and the receiving apparatus, or even if a mismatch occurs between the up-conversion/down-conversion frequency and the center of the RF filter, or even if a mismatch occurs between the carrier frequency and the center frequency of the cell, the signal recovery performance at the receiving apparatus can be maintained.
In addition, in some scenarios, the carrier frequency can be easily adjusted without RF retuning when the carrier frequency changes in the same frequency band.
Reference will now be made in detail to various implementations of the present disclosure, examples of which are illustrated in the accompanying drawings. The detailed description given below with reference to the accompanying drawings is intended to explain exemplary implementations of the present disclosure, and does not show the only implementations that can be implemented according to the present disclosure. The following detailed description includes specific details to provide a thorough understanding of the present disclosure. However, it will be apparent to one skilled in the art that the present disclosure may be practiced without such specific details.
In some instances, well-known structures and devices are omitted or shown in block diagram form, focusing on important features of the structures and devices so as not to obscure the concepts of the present disclosure. Throughout the specification, the same reference numerals will be used to refer to the same or like parts.
The following techniques, devices, and systems may be applied to various wireless multiple access systems. Examples of multiple-access systems include Code Division Multiple Access (CDMA) systems, Frequency Division Multiple Access (FDMA) systems, Time Division Multiple Access (TDMA) systems, Orthogonal Frequency Division Multiple Access (OFDMA) systems, single carrier frequency division multiple access (SC-FDMA) systems, and multi-carrier frequency division multiple access (MC-FDMA) systems. CDMA may be embodied by a radio technology such as Universal Terrestrial Radio Access (UTRA) or CDMA 2000. TDMA may be embodied by radio technologies such as global system for mobile communications (GSM), General Packet Radio Service (GPRS), or enhanced data rates for GSM evolution (EDGE). OFDMA may be embodied by radio technologies such as Institute of Electrical and Electronics Engineers (IEEE)802.11(Wi-Fi), IEEE802.16(WiMAX), IEEE 802.20, or evolved UTRA (E-UTRA). UTRA is part of the Universal Mobile Telecommunications System (UMTS). The 3 rd generation partnership project (3GPP) Long Term Evolution (LTE) is part of an evolved UMTS (E-UMTS) that uses E-UTRA. 3GPP LTE employs OFDMA in the DL and SC-FDMA in the UL. LTE-advanced (LTE-A) is an evolved version of 3GPP LTE. For ease of description, implementations of the present disclosure are described herein as applied to 3 GPP-based communication systems, particularly NR systems. However, the technical features of the present disclosure are not limited thereto. Although the following detailed description is based on a mobile communication system corresponding to the 3GPP NR system, it is applicable to any other mobile communication system except for the unique function of the 3GPP NR. For example, the present disclosure is applicable to a communication technology in which an up-conversion frequency and a down-conversion frequency are not previously shared between a transmitting apparatus and a receiving apparatus, and a communication technology in which a mismatch may occur between the up-conversion frequency and the center of an RF filter or between the up-conversion/down-conversion frequency and the center frequency of a cell.
In the present disclosure, a User Equipment (UE) may be a fixed device or a mobile device. Examples of the UE include various apparatuses that transmit and receive user data and/or various control information to and from a Base Station (BS). A UE may be referred to as a Terminal Equipment (TE), a Mobile Station (MS), a Mobile Terminal (MT), a User Terminal (UT), a Subscriber Station (SS), a wireless device, a personal digital assistant, a wireless modem, a handheld device, etc. In addition, in the present disclosure, a Base Station (BS) generally refers to a fixed station that performs communication with a UE and/or another BS and exchanges various data and control information with the UE and/or another BS. The BS may be referred to as an Advanced Base Station (ABS), a node b (nb), an evolved node b (enb), a Base Transceiver System (BTS), an Access Point (AP), a Processing Server (PS), and the like. Specifically, the base station of the UTRAN is called Node-B, the base station of the E-UTRAN is called eNB, and the base station of the new radio access technology network is called gNB.
In the present disclosure, a node refers to a fixed point configured to transmit/receive a radio signal through communication with a UE. Regardless of its terminology, various types of enbs may be used as nodes. For example, a BS, node b (nb), electronic node b (enb), pico cell enb (penb), home enb (henb), relay, etc. may be a node. In addition, the node may not be a BS. For example, a node may be a Radio Remote Head (RRH) or a Radio Remote Unit (RRU). The RRH or RRU typically has a lower power level than the power level of the BS. Since an RRH or an RRU (hereinafter, RRH/RRU) is generally connected to a BS through a dedicated line such as an optical cable, cooperative communication between the RRH/RRU and the BS can be smoothly performed as compared with cooperative communication between BSRs connected through a radio line. Each node may be equipped with at least one antenna. The antenna may be a physical antenna or an antenna port or a virtual antenna.
In the present disclosure, a cell may refer to a designated geographical area in which one or more nodes provide communication services. Accordingly, in the present disclosure, communicating with a specific cell may include communicating with a BS or node providing a communication service for the specific cell. In addition, the DL/UL signal of a specific cell means a DL/UL signal from/to a BS or a node providing a communication service for the specific cell. A node providing the UL/DL communication service to the UE is referred to as a serving node, and a cell providing the UL/DL communication service by the serving node is particularly referred to as a serving cell.
3 GPP-based communication systems typically implement cells to manage radio resources, and the cells associated with radio resources are distinguished from cells of a geographic area. For example, a "cell" of a geographical area may be understood as a coverage area that a node can provide service using a carrier, while a "cell" of radio resources is associated with a Bandwidth (BW) that is a frequency range configured by the carrier. Since the DL coverage, which is the range where the node can transmit a valid signal, and the UL coverage, which is the range where the node can receive a valid signal from a UE, depend on the carrier carrying the signal, the coverage of the node may be associated with the coverage of the "cell" of the radio resource used by the node. Thus, the term "cell" may sometimes be used to indicate the service coverage of a node, to indicate radio resources at other times, or to indicate a range that a signal using radio resources can reach with a valid strength at other times. A "cell" associated with a radio resource is defined by a combination of downlink resources and uplink resources, i.e., a combination of DL Component Carriers (CCs) and UL CCs. The cell may be configured by only downlink resources or may be configured by downlink resources and uplink resources.
For terms and techniques not specifically described among the terms and techniques used in the present specification, reference can be made to 3GPP LTE/LTE-a standard documents such as 3GPP TS 36.211, 3GPP TS 36.212, 3GPP TS 36.213, 3GPP TS 36.321, and 3GPP TS36.331, and 3GPP NR standard documents such as 3GPP TS 38.211, 3GPP TS 38.212, 3GPP38.213, 3GPP 38.214, 3GPP 38.215, 3GPP TS 38.321, 3GPP TS 38.300, and 3GPP TS 38.331.
Referring to standard 3GPP TS 36.211, an OFDM symbol baseband signal, for example, a single-carrier frequency division multiple access (SC-FDMA) baseband signal, is generated as follows for all physical signals and physical channels except for a physical random access channel. In LTE systems, time interval 0 ≦ t < (N) by the following equationCP，l+N)×Ts(wherein Fast Fourier Transform (FFT) size N equals 2048) to define a time-continuous signal s in SC-FDMA symbol l in the uplink slotl(t)。
Wherein the content of the first and second substances,subcarrier spacing Δ f 15kHz, ak，lIs the content of the resource element (k, l). The index k is from 0 to N in the frequency domainUL RB×
In an LTE system, the signal utilization N of the uplink transmission in each time slotUL RB×NRB scSub-carriers and NUL symbA resource grid of OFDM symbols. Each resource element in the resource grid is uniquely defined by an index pair (k, l) in a slot, where k is 0UL RB×
TABLE 1
SC-FDMA symbols in a slot are transmitted in increasing order of l starting from l-0, wherein the SC-FDMA symbol l>0 is expressed in time slot
For 0 ≦ t < (N) by the following formulaCP，l+N)×TsDefining a time-continuous signal on an antenna port p in an Orthogonal Frequency Division Multiplexing (OFDM) symbol l in a downlink time slot
Formula 2
Wherein the content of the first and second substances,and is
In the LTE system, the downlink transmission signal of each time slot is composed of NDL RB×NRB scSub-carriers and NDL symbA resource grid description of a number of OFDM symbols. Each resource element in the resource network is uniquely identified in a time slot by an index pair (k, l), where k is 0DL RB×
TABLE 2
Fig. 1A and 1B illustrate examples of modulation and upconversion of a baseband signal to a carrier frequency that may be implemented in some systems (e.g., LTE systems). Specifically, fig. 1A illustrates an example of modulation and up-conversion of a complex-valued SC-FDMA baseband signal to a carrier frequency for each antenna port, and fig. 1B illustrates an example of modulation and up-conversion of a complex-valued OFDM baseband signal to a carrier frequency for each antenna port.
Filtering may be performed prior to uplink transmission, for example as specified in standard 3GPP TS 36.101, and may likewise be performed prior to downlink transmission, for example as specified in standard 3GPP TS 36.104. In the example of fig. 1A and 1B, the frequency f0Is the upconversion frequency. In some scenarios (e.g., in LTE systems), the up-conversion frequency may be set equal to the carrier frequency of the cell.
In some systems (e.g., in LTE systems), since the synchronization signal of a cell and the Physical Broadcast Channel (PBCH) of the cell are transmitted in a total of six RBs around the carrier frequency of the cell, a User Equipment (UE) of the LTE system can know the downlink carrier frequency of the cell by acquiring the synchronization signal and the PBCH. In such a scenario, if the UE and the Base Station (BS) know the downlink carrier frequency, they may also know the uplink carrier frequency in the following scenario: (i) the downlink carrier frequency and the uplink carrier frequency are the same, e.g., in the case of Time Division Duplex (TDD); or (ii) an uplink carrier frequency used with a downlink carrier frequency is predetermined, e.g., in the case of Frequency Division Duplex (FDD); or (iii) the uplink carrier frequency is explicitly broadcast by system information of the cell, etc. As a result, in this scenario of the LTE system, both the UE and the Base Station (BS) can know the carrier frequency used by the cell to transmit/receive radio signals.
In the conventional LTE system, the following frequencies are configured to be the same: (i) the center of a Radio Frequency (RF) filter (e.g., a filter between the IFFT and the up-conversion, a filter applied after the up-conversion, etc.); (ii) a center frequency of a carrier bandwidth; and (iii) an upconversion frequency f0. In addition, the same frequency is used for up-converting the baseband into a carrier frequency signal and for down-converting the radio signal into a baseband signal.
However, with the increasing various utilizations of Machine Type Communication (MTC), internet of things (IoT) communication, and ultra-reliable low latency communication (URLLC), new radio access technologies (NRs) different from the conventional LTE communication technologies are being developed. The NR system considers using frequencies on a frequency band used in a conventional communication system and also considers supporting a much wider bandwidth than the frequency band used in the conventional communication system. In view of the disadvantage that the conventional LTE system is difficult to introduce a communication technology having forward compatibility due to various constraints, NR systems are being developed to reduce such constraints, thereby facilitating the introduction of future communication technologies having forward compatibility with NR systems.
Therefore, in the NR system, the frequency for up-conversion of the baseband signal is not necessarily limited to the center frequency of the cell. In addition, in the NR system, the frequency resource for transmitting the synchronization signal is not necessarily limited to the center of the frequency band of the cell.
Considering that a UE may not be able to support a wide bandwidth to be supported in an NR system at once, the UE may be configured to operate in a portion of a frequency bandwidth of a cell (hereinafter referred to as a bandwidth portion (BWP)). The BWP may be assigned based on any reference point. The reference point need not be limited to the center frequency of the cell. If only a portion of the frequency bandwidth of a cell is used for communications such as BWP-based communications and NB-IoT, the receiving device may not know the upconversion frequency used by the transmitting device prior to downconverting the received signal.
Therefore, the up-conversion frequency for the baseband signal may be different from the down-conversion frequency for the radio signal, and the up-conversion frequency is not necessarily limited to the center of the RF filter.
Furthermore, it is expected that various parameter sets will be supported in the NR system. The subcarrier spacing may change if the set of parameters for the same frequency band changes. Such a change in subcarrier spacing may result in a change in up-conversion frequency or down-conversion frequency. Therefore, there is a need for a technique in which a transmitting apparatus and a receiving apparatus can easily adjust an up-conversion frequency and a down-conversion frequency, respectively.
Before explaining the implementation of the present disclosure in more detail, the basic frame structure and physical resources of the NR system discussed so far will be described to facilitate understanding of the present disclosure.
In this disclosureIn the description, unless otherwise stated, the size of each field in the time domain is in time units Tc＝1/(Δfmax*Nf) Is represented by, wherein Δ fmax＝480*103Hz, and Nf4096 or in time units Ts。TcIs the basic time unit of NR. Constant k ═ Ts/Tc64 where Ts＝1/(Δfref*Nf，ref)，Δfref＝15*103Hz，Nf，ref＝2048。TsIs the basic time unit of LTE. In NR, various OFDM parameter sets are supported, as given in the table below, where the cyclic prefix of the μ and bandwidth parts is given by higher layer parameters provided by the BS.
TABLE 3
μ
Δf＝2μ*15[kHz]
Cyclic prefix
0
15
Is normal
1
30
Is normal
2
60
Normal, extended
3
120
Is normal
4
240
Is normal
The downlink and uplink transmissions are organized to have a Tf＝(ΔfmaxNf/100)*TcFrames of 10ms duration, each frame consisting of Tsf＝(ΔfmaxNf/1000)*Tc1ms duration. The number of consecutive OFDM symbols per sub-frame is
For subcarrier spacing configuration mu, slots are numbered in increasing order within a subframe
TABLE 4
TABLE 5
In the case of tables 4 and 5,indicating the number of symbols per slot and,is the number of slots per frame for the subcarrier configuration mu,is the number of slots per subframe for which the subcarrier configuration μ.
For each parameter set and carrier, define
Resource Blocks (RBs) are defined as N in the frequency domainRBsc is 12 contiguous subcarriers. The reference resource blocks are numbered from 0 up in the frequency domain. Subcarrier 0 of reference resource block 0 is common to all subcarrier spacing configurations μ, also denoted as "reference point a" or "point a", and serves as a common reference point for other resource block grids. Reference point a is obtained from higher layer parameters provided by the BS. For the subcarrier spacing configuration μ, Common Resource Blocks (CRBs) are numbered from 0 up in the frequency domain. Subcarrier 0 of common resource block 0 for which μ is configured for subcarrier spacing coincides with reference point a. Common resource block number n in frequency domainCRBThe relationship with the resource elements (k, l) configuring μ for the subcarrier spacing is given by the following equation.
Formula 3
Where k is defined relative to subcarrier 0 of the resource grid for the subcarrier spacing configuration μ.
In NR systems, Physical Resource Blocks (PRBs) are defined within a carrier bandwidth portion and range from 0 to
Formula 4
Wherein the content of the first and second substances,is the common resource block with the carrier bandwidth part starting with respect to common resource block 0.
The bandwidth part is a bandwidth part i for a given set of parameters mu in a given carrieriA subset of defined contiguous common resource blocks. Starting position of resource block in bandwidth partAnd quantity ofShould satisfy
In some wireless communication systems, the carrier frequencies used by the transmitter and the receiver are known to each other, and the transmitter and the receiver set the same carrier frequency as the up-conversion frequency and the down-conversion frequency, respectively. However, due to inaccuracies in the analog oscillator or Phase Locked Loop (PLL), an error, i.e., a frequency offset, occurs between the frequencies generated by the transmitter and receiver. In this case, the signal phase varies depending on the symbol of the receiving end. However, in general, phase variations due to inaccuracies of the analog module are not so severe as to make channel estimation with Reference Signals (RSs) useless, and in general, such phase variations do not greatly affect the recovery of the received signal.
On the other hand, in a radio communication system such as an NR system supporting a broadband cell, the UE and the BS may have to operate without information on carrier frequencies for up-conversion known to the UE and the BS. Therefore, when the UE and the BS perform up-conversion and down-conversion using different carrier frequencies, the phase of the receiving apparatus may abruptly change in each symbol even if it is assumed that there is no frequency offset (i.e., frequency error) due to the inaccuracy of the analog oscillator or the PLL, as will be described later.
For any physical channel or signal other than the Physical Random Access Channel (PRACH), a time-continuous signal of μ is configured on an antenna port p and subcarrier spacing for an OFDM symbol l in a subframe
Formula 5
Wherein the content of the first and second substances,
formula 6
Wherein the content of the first and second substances,
In the case of the formulas 5 and 6,
Formula 7
Here, the effective symbol length N of the OFDM symbol lμ uAnd Cyclic Prefix (CP) length N of OFDM symbol lμ CP，lComprises the following steps:
formula 8
Time-continuous signal on antenna port p of PRACHIs defined by the following formula.
Formula 9
K＝Δf/ΔfRA
Wherein the content of the first and second substances,a detailed description of each parameter in equation 9 can be found in 3GPP TS 38.211.
The frequency of use of the transmitter is fTxWill configure the OFDM symbol baseband signal of mu for antenna port p and subcarrier spacingUp-conversion to up-frequency fTx. OFDM symbol baseband signal with mu configured for antenna port p and subcarrier spacingTo an up-conversion frequency fTxThe up-conversion of (a) can be expressed as:
In the
Formula 11
Even if the transmitting apparatus transmits only the real value of the complex signal, the receiving apparatus applies the FFT after the received signal converts the reply signal. Therefore, in the description of the present disclosure, a transmission signal is represented as a complex signal for convenience, and is equivalent to a real signal in modeling. The same applies to the receive operation.
When the receiving device receives the radio signal x(p,μ)(t) time, the receiving device is paired with x(p,μ)(t) performing frequency down-conversion to obtain a baseband signal
To show the received signalOf phase change, frequency fTxAnd fRxCan be expressed as fTx＝NTx*Δf+ΔoffsetAnd fRx＝NRx*Δf+ΔoffsetWhere Δ f is the subcarrier spacing, term NTxIs closest to fTxPositive integer of/[ delta ] f (e.g., floor { f)Tx/. DELTA.f or ceil { f)Tx/Δ f }), the term NRxIs closest to fRxPositive integer of/[ delta ] f (e.g., floor { f)Rx(Δ f) or ceilfRx/Δ f }) and ΔoffsetIs a real number, whose magnitude is less than Δ f. In the description of the present disclosure, the same Δ is used for simplicityoffsetTo represent fTxAnd fRxBut at fTxAnd fRxDelta betweenoffsetMay be different.
Using these expressions,
Even in an environment without frequency offset, which is a frequency error unintentionally generated by the characteristics of the transmitter/receiver component, if f is due to the following reasonTxIs not equal to fRxThen receiving the signal
Fig. 2A and 2B are diagrams illustrating examples of phase changes according to the difference between an up-conversion frequency and a down-conversion frequency in terms of devices and signal waveforms.
Referring to fig. 2A, an information symbol a intended to be transmitted by a transmitting apparatuskIs converted into an OFDM baseband signal s (t) by IFFT. The frequency of use of the transmitter is fTxUp-converting s (t) toWhen in use
Referring to fig. 2B, when a Cyclic Prefix (CP) is added to an IFFT signal obtained by performing IFFT on an information symbol, an OFDM symbol signal is obtained. The CP added to the IFFT signal causes a transition in the time domain with respect to the IFFT signal waveform. As a result, when the OFDM symbol signal is loaded onto the signal of the free-running OSC, the phase of the transmitted signal may not be zero at the beginning of the OFDM symbol. In addition, the phase of the transmission/reception signal may be different in the beginning of the OFDM symbol.
Therefore, if fTxIs not equal to fRxThen due to fTxAnd fRxSudden phase change between symbols caused by a difference between themIn the course of signal recovery through channel estimation at the receiving end, performance may be significantly degraded. If the phase abruptly changes between OFDM symbols, the receiver cannot apply a channel estimation value obtained using a Reference Signal (RS) of a specific OFDM symbol to other OFDM symbols, or the received signal may not be properly recovered when the channel estimation value is used. It is not proper for the transmitter to insert an RS in each OFDM symbol so that the receiver correctly estimates the channel state of each symbol because RS overhead becomes excessive.
Various types of techniques may be used in NR systems to mitigate this phase discontinuity/mismatch between symbols. Some examples of this technique, as well as potential drawbacks of each technique, are described below.
Technique a: the gNB informs the UE of the carrier frequency used by the gNB, and the UE compensates for the corresponding phase discontinuity.
According to this scheme, if the BS transmits a transmission signal without separately performing pre-compensation, the UE performs compensation of phase discontinuity for each symbol using carrier frequency information of the BS. For example, the UE as a receiver performs phase compensation symbol by symbol to eliminate the phase error due to
Technique B: the BS as a transmitter performs phase pre-compensation using DL carrier frequency information of the UE.
For example, the techniques may be implemented in an NB-IOT system as operations before the receiver receives the information about the carrier frequency in implementation a. For example, the transmitter symbols by symbolsPerforming phase pre-compensation to eliminate the phase error due to
Technique C: the transmitter and the receiver perform phase pre-compensation on the premise of a common reference point.
In this technique, the transmitter does not use information about the carrier frequency of the receiver (and the receiver does not use information about the carrier frequency of the transmitter). Instead, a common reference point is predefined between the transmitter and the receiver, and phase pre-compensation with respect to this reference point is performed symbol by symbol. For example, the transmitter is paired withThe phase discontinuity that occurs is compensated for, and the receiver pair is due toThe phase discontinuity that occurs performs the phase (pre-) compensation. As a specific example, in some scenarios, equation 5 above may be modified as follows for phase pre-compensation.
Formula 14
Here,. DELTA.fref15kHz, and
For reference, the NR standard does not explicitly specify a specific technique for implementation. In the NR standard, modulation and upconversion techniques are defined as shown in the following table such that a transmitting side and a receiving side reset carrier frequencies to zero phase symbol by symbol, respectively, to maintain a certain value of phase of the carrier frequency at each symbol start point (see section 5.4 of 3GPP TS 38.211). This is specified in the standard document 3GPP TS 38.211 V15.1.0 as follows.
TABLE 6
Fig. 3 illustrates an example of resetting the phase at a symbol boundary. That is, fig. 3 is a diagram illustrating an example of phase compensation defined in the NR standard. In FIG. 3, TCPCorresponding to Table 6
Formula 15
When equation 15 is rearranged to explain how equation 15 occurs in an actual implementation, the following equation can be obtained.
Formula 16
The techniques described so far to increase phase discontinuity/mismatch define a phase reset at the carrier frequency level. For the actual conversion (or down conversion) of the carrier frequency stage components such as Phase Locked Loops (PLLs) and mixers are used.
Fig. 4A and 4B are examples of generation and modulation and upconversion of a baseband signal to its carrier frequency according to some implementations of the present disclosure.
Referring to fig. 4A and 4B, for example, a PLL is used to generate a carrier frequency for actual conversion (or down conversion), and a mixer or the like is used for up conversion to the carrier frequency. Components such as PLLs and mixers are implemented as analog devices or operate at extremely high speeds, so in some scenarios it may be difficult to implement phase resetting of the carrier frequency level at the transmit and receive sides.
In other words, referring to table 6, the NR standard specifies that the phase reset should be achieved by directly controlling the phase of the carrier frequency. However, in some scenarios, it may be difficult to directly control the phase of the carrier frequency in practice. Therefore, since it is practically difficult to directly control the phase at the carrier frequency, some systems perform up-conversion and down-conversion using a carrier frequency having a continuous phase at the carrier frequency stage, and further implement a phase reset function at the baseband stage to eliminate phase discontinuity/mismatch between symbols caused by up-conversion/down-conversion at the analog stage.
In such a system, referring to equation 16, up-conversion and down-conversion at a carrier frequency level using carrier frequencies having continuous phases respectively correspond to
therefore, in such a system, the transmitting side and the receiving side compensate for the phase using only their respective carrier frequencies, respectively corresponding to those in equation 16
Therefore, the current NR standard (3GPP TS 38.211 V15.1.0) can be understood as specifying that the transmitting side utilizes the carrier frequency for down-conversion to 0
As discussed above (with respect to techniques a-C), some wireless communication systems based on the NR standard may apply a phase compensation term by calculating the phase in each symbol based on the carrier frequency. However, consider as fXX(where XX is Tx or Rx) is a myriad of applicable or available frequencies, this phase compensation term can become a function of very high resolution and very long periods, potentially imposing very complex implementations.
To address such challenges, the present disclosure describes implementations for addressing scenarios in which transmitters and receivers operate without knowledge of the carrier frequency used for transmission or knowledge of the up/down conversion frequency.
Techniques to reduce complexity
The current NR standard defines the following parameter sets (see, e.g., section 5.4 of 3GPP TS 38.101-1 and section 5.4 of 3GPP TS 38.101-2).
TABLE 7
TABLE 8
The NR standard has two main Frequency Ranges (FR) specified in 3 GPP. One is generally referred to as 6GHz or less and corresponds to a frequency range FR1 in tables 7 and 8, and the other is referred to as millimeter wave and corresponds to a frequency range FR2 in tables 7 and 8. The maximum bandwidth and the available subcarrier spacing are different depending on the frequency range.
Table 7 shows the channel trellis, i.e., NR-ARFCH definition, and table 8 shows the synchronization trellis.
The channel grid defines a set of Radio Frequency (RF) reference frequencies that are used to identify the location of RF channels. The RF reference frequencies for the RF channels are mapped to resource elements on the carrier. A global frequency grid is defined for all frequencies from 0 to 100GHz and is used to define a set of allowed RF reference frequencies. Granularity of the global frequency grid is Δ FGlobal. For each operating band, a subset of frequencies from the global frequency grid is applied to that band, with a granularity Δ FGlobalA channel grid for the frequency band is formed.
The synchronization grid represents the frequency location of synchronization (SS) blocks that the UE can use for system acquisition when there is no explicit signaling about SS block location. A global synchronization grid is defined for all frequencies, and the frequency positions of the SS blocks are definedFor an SS having a corresponding Global Synchronization Channel Number (GSCN)REF。
The mapping between the synchronization grid and the corresponding resource elements of the SS block is given in table 8. The mapping depends on the total number of RBs allocated in the channel and applies to both UL and DL. Table 8 shows the positions of Resource Elements (REs) #0 (i.e., subcarriers #0) of
As shown in tables 7 and 8, the channel trellis and the SS trellis are fixed to a certain value. Thus, if the carrier frequency is denoted as fTx＝NTx*Δf+ΔoffsetThen the offset term ΔoffsetMay be used for the frequency range FR1(<3GHz) may be limited to some specific values (e.g., -5kHz, 0, or 5kHz) and may be limited to 0 for the remaining frequency bands. In the above expression, as discussed with respect to
In addition, for a sampling time determined based on Δ f, the number of samples including the number of samples for a Cyclic Prefix (CP) used in the current LTE/NR communication system is an integer multiple of 16 for each subcarrier interval, that is, the CP length is 144-16 × 9 or 160-16 × 10, and the length of the signal part of the OFDM symbol other than the CP part is 2048-16 × 128 for example, for a bandwidth of 20MHz, which is a subcarrier interval of Δ f-15 kHz in the LTE or NR standard, the sampling frequency is 30.72MHz, one subframe (or one slot) is composed of 30720 samples, and each OFDM symbol is composed of 2048+144 sampling times or 2048+160 sampling times for referencesIs 1/(30.72MHz), i.e., Ts＝1/(2048×15×103kHz)。
In some scenarios, NR and LTE systems may use as parameter sets values that are proportional to parameter integration corresponding to a bandwidth of 20MHz as a subcarrier spacing of 15kHz, so it should be noted that all frequencies described in this disclosure are based on parameter sets corresponding to a bandwidth of 20MHz as a subcarrier spacing of 15 kHz. Here, 2048 is a signal length (i.e., an effective symbol length of an OFDM symbol) defined by an FFT size when the above parameter sets (e.g., 15kHz subcarrier spacing and 20MHz bandwidth) are used, and 144 and 160 correspond to a Cyclic Prefix (CP) length when the above parameter sets (e.g., Δ f ═ 15kHz subcarrier spacing and 20MHz bandwidth) are used.
For example, phase resetting may be implemented for the transmit and receive signals to account for scenarios in which the signal period according to the upconversion frequency is not an integer multiple of the OFDM symbol length, scenarios in which the OFDM symbol length is equal to the length of the Cyclic Prefix (CP) portion plus the length of the signal portion. Therefore, in some scenarios, if a carrier frequency is used in which an integer multiple of the signal period has a period corresponding to the OFDM symbol length, phase resetting may not be achieved.
For example, consider the following communication system: the Cyclic Prefix (CP) portion of the OFDM symbol consists of 144-16-9 samples or 160-16-10 samples, while the signal portion of the OFDM symbol consists of 2048-16-128 samples and the IFFT/FFT size is 2048. In such a system, if the up-conversion frequency is set to a frequency of 16 samples in period or 1/(16 × T)s) Then no phase reset will be required where 16 is the greatest common divisor of 144, 160, 2048. Instead of a sampling time TsThen this corresponds to equal 1/(16 × T)s) An upconversion frequency of 1/{16 × 1/(FFT size × Δ f) } ═ 1/{16 × 1/(2048 × Δ f) } ═ 128 Δ f. Thus, if the up-conversion frequency is set to the
The reason for this is that the frequency with a periodicity of 16 samples (16 being the greatest common divisor of 144, 160 and 2048) has the same phase at the beginning of each signal portion of the OFDM symbol. In particular, this is because for a period of 16 × TsAt a length of 144TsCP ofThe part comprises 9 sine waves with the length of 160TsIncludes 10 such sine waves and has a length of 2048TsIncludes 128 such sinusoids in the CP portion of (a). For example, considering that the minimum subcarrier spacing Δ f supported by the NR system is 15kHz, if carrier frequencies corresponding to an integer multiple of 15kHz 2048/16 15kHz 128 1.92MHz are used, the phase naturally becomes 0 at the beginning of the signal portion of each OFDM symbol, and thus the phase shift problem does not occur.
In addition, according to subcarrier spacing Δ f of 15kHz, Cyclic Prefix (CP) length is 144TsAnd 160TsAnd the signal part (i.e., effective symbol) length of the OFDM is 2048TsThe phase will be 0 even at the CP starting point in case the frequency corresponds to an integer multiple of 1.92 MHz. More generally, when multiple CP lengths (e.g., N) are defined for OFDM symbol signal generationCP，1、NCP，2… …) and the effective number of samples per OFDM symbol (i.e., the number of samples of the signal portion in the OFDM symbol other than the CP (i.e., the IFFT/FFT size)) is NsampleThen the frequency that does not cause phase discontinuity for each symbol will be one period corresponding to the number and NCP，1、NCP，2、…、NsampleThe frequency of samples corresponding to the largest common divisor of.
Such frequencies that do not cause phase discontinuity of each symbol can be represented using subcarrier spacing as follows:here, gcd { N {CP，1、NCP，2、…、NsampleIs NCP，1、NCP，2… …, and NsampleThe greatest common divisor of (c).
This is applied to the parameter sets 2048, 160 and 144 above, resulting in NbaseΔ f ═ 128 Δ f. If FFT size 4096 is used, the Cyclic Prefix (CP) length is changed to 144 x 2 and 160 x 2 in the NR standard. Thus, the modified CP length applies toTo obtain NbaseThe same result as 128.
As another example, even when the FFT size is reduced from 2048 to, for example, 1024, the CP length is changed to 144/2-72 and 160/2-80. Thus, the modified CP length applies toTo obtain NbaseThe same result as 128.
In the following, as an example, the present disclosure describes an implementation of a communication system (e.g., an LTE system or an NR system) in which a length of a Cyclic Prefix (CP) portion and a length of a signal portion are 144 x 2μOr 160 x 2μAnd the length of the signal portion of the OFDM symbol is 2048 × 2μ(where μ is an integer).
In such a scenario, 128 Δ f may be used to represent frequencies that do not cause phase discontinuities. However, the implementation is not limited thereto, and the present disclosure is applicable even in a scenario where a CP length and a signal part length different from those illustrated are used.
For example, implementations of the present disclosure may be applied where one cycle corresponds to a number and { N }CP，1、NCP，2、……、NsampleThe frequency of samples corresponding to the greatest common divisor of is used as the fundamental carrier frequency fbase(i.e., useInteger multiple of fbase) The scene (2).
Thus, in the following description, the basic carrier frequency f described above as an integer multiple of 128 Δ fbaseCan be summarized as one period corresponding to the number and NCP，1、NCP，2、……、NsampleInteger multiples of the frequency of the samples corresponding to the greatest common divisor of }, or
For the scenario where a basic carrier frequency of 128 Δ f is used, fTxCan be expressed as:
formula 17
fTx＝NTx·Δf+Δoffset＝Nint·128Δf+Nfrac·Δf+Δoffset＝fbase+ffrac+Δoffset
Here, the itemItem Nfrac＝modulo(NTx128), term fbaseIs a quantized (down-quantized, e.g., with a rounding down function) version of the carrier frequency obtained at a resolution of 128 af (e.g., a resolution of 1.92MHz at 15 kHz), and the term ffracIs directed to fbaseAnd fTxThe difference is a quantized version of Δ f.
Term ΔoffsetDenotes the deviation from frequency in units of Δ f 15 kHz. In NR systems, e.g. ΔoffsetMay be set to +/-5, 0 kHz. Specifically, based on the subcarrier spacing of Δ f ═ 15kHz, ΔoffsetMay be defined as one of-5 kHz, 5kHz and 0 kHz. In some implementations, N may be replaced with a rounding function instead of a rounding down functionint. In this case, NfracCan be defined as Nfrac＝NTx–128*round(NTx/128). When N is replaced by a rounding function instead of a rounding-down functionintThen, remove Nfrac＝NTx–128*round(NTx/128) and operation at NintThe same is defined for the floor function.
In equation 17, the fundamental carrier frequency fbaseIs a frequency for always resetting the phase to a certain value in units of OFDM symbols. Thus, the expression corresponding to the phase compensationHas the following advantagesThe same value.
Phase compensation term to be applied to one symbol for carrier frequency difference between transmitter and receiver considering only normal Cyclic Prefix (CP)
Thus, if a set of phase compensation values for a plurality of symbols is represented as a sequence, considering only the normal Cyclic Prefix (CP), then according to Nfrac0, …, 127 and Δoffset-5kHz, 0kHz, 5kHz combination, phase compensation term128 x 3 sequences are required. If will be deltaoffsetB instead of 3, 128 × b sequences may be required for phase compensation. For having aoffsetFrequency range FR1 (> 3GHz) or FR2 of 0, only 128 sequences may be required (e.g., NfracI.e. assuming that one period of the signal part of each OFDM symbol is 2048 sample times, if the phase value for a particular carrier frequency at any OFDM symbol boundary is α, because of the sample time T, a phase compensation term has a period of up to
Formula 18
Using carrier frequency fTx＝NTx*Δf+ΔoffsetThe final signal on the transmitting side can be given as follows.
Formula 19
In
Fig. 5A to 5C are diagrams illustrating an example of
Referring to fig. 5A, the transmitting side aims at a carrier frequency f by using 128 complex-valued sequences or 128 × 3 complex-valued sequences before up-converting the OFDM baseband signal to the carrier frequencyTxOne complex-valued sequence is computed to perform multiplication (i.e., perform phase reset) on the signal at each symbol. Then, the transmitting side uses fTxAn up-conversion is performed. Using for the carrier frequency fTxOne of the 128 complex-valued sequences (or one of the 128 x 3 complex-valued sequences) performs phase compensation and applies a plurality of elements constituting the respective complex-valued sequence to the plurality of OFDM symbols in a one-to-one correspondence.
As described above for the transmitter architectureAt carrier frequency denoted fRx＝NRx*Δf+ΔoffsetIn the frequency range FR1 (f), the receiver is located<3GHz) inner, ΔoffsetMay be-5 kHz, 0kHz or 5kHz, and in other frequency bands is 0 kHz. In addition, for the sampling time determined based on Δ f, the number of samples including the number of samples for a Cyclic Prefix (CP) used for each symbol in the current LTE/NR communication system is an integer multiple of 16 for each subcarrier interval. Therefore, in this case, when Δ f is 15kHz, if carrier frequencies corresponding to an integral multiple of 15kHz 2048/16 15kHz 128 1.92MHz are used for down-conversion, the phase of the signal portion (i.e., the effective symbol) naturally starts from 0 at each OFDM symbol, and thus the above-described problem may not be caused. Further, assume that the subcarrier spacing Δ f is 15kHz and the CP length is 160TsAnd 144TsAnd the effective symbol length is 2048TsThis corresponds to the set of parameters currently available in LTE and NR systems, with a phase of 0 even at the CP starting point, if the frequency corresponds to an integer multiple of 1.92 MHz.
More generally, when multiple Cyclic Prefix (CP) lengths (e.g., N) are defined for OFDM symbol signal generationCP，1，NCP，2…) and the effective number of samples per OFDM (i.e., the number of samples in the signal portion other than the CP in the OFDM symbol (i.e., IFFT/FFT size)) is NsampleThen one period corresponds to the number and NCP，1，NCP，2，…，NsampleThe frequency of samples corresponding to the greatest common divisor of is the frequency that does not cause phase discontinuity of each symbol. Such frequencies that do not cause phase discontinuity bits per symbol can be represented using subcarrier spacing as follows:
It is applied to the parameter sets (i.e., 2048, 160, 144) described above, resulting inTo NbaseΔ f ═ 128 Δ f. If the FFT size is 4096 in the example of 20MHz bandwidth as a sub-carrier spacing of Δ f 15kHz, the Cyclic Prefix (CP) length is changed to 144 x 2 and 160 x 2 in the NR standard. Thus, the modified CP length applies toTo obtain NbaseThe same result as 128. As another example, even when the FFT size is reduced from 2048 to, for example, 1024, the CP length is changed to 144/2-72 and 160/2-80. Thus, the modified CP length applies toTo obtain NbaseThe same result as 128. As described above, the following 128 Δ f will be used to represent the frequency that does not cause phase discontinuity.
Using 128 Δ f, fRxCan be expressed as:
fRx＝NRx·Δf+Δoffset＝Nint·128Δf+Nfrac·Δf+Δoffset＝fbase+ffrac+Δoffset
Here, the first and second liquid crystal display panels are,Nfrac＝modulo(NRx，128)，fbaseis a quantized (down-quantized, i.e. applying a down-rounding function) version among the carrier frequencies obtained at a resolution of 128 af (e.g. a resolution of 1.92MHz when af is 15 kHz), and ffracIs directed to fbaseAnd fTxThe difference is a quantized version of Δ f. DeltaoffsetRepresents the amount of distance frequency in units of Δ f-15 kHz. In NR systems, e.g. ΔoffsetMay be set to +/-5, 0 kHz. Specifically, based on the subcarrier spacing of Δ f ═ 15kHz, ΔoffsetMay be defined as one of-5 kHz, 5kHz and 0 kHz. In some implementations, N may be replaced with a rounding function instead of a rounding down functionint. In this case，NfracCan be defined as Nfrac＝NTx–128*round(NTx/128). When N is implemented by rounding functions instead of rounding functionsintThen, remove Nfrac＝NTx–128*round(NTx/128) and operation at NintThe same is defined for the floor function.
In
When using carrier frequency fRx＝NRx*Δf+ΔoffsetWhen the receiving-side operation is defined according to the original definition of (1), then the receiving-side operation according to the
In
The difference between the examples of fig. 5B and 5C is the location of the phase reset function, and the other functions are the same.
In use fRxPerforming down-conversion or using f on the received signalRxAfter down-converting and FFT are performed on the received signal, the receiving side performs down-conversion and FFT on the received signal by using a carrier frequency f among 128 complex sequences or among 128 x 3 complex sequencesRxA complex-valued sequence is computed, and phase-resetting is performed by multiplying the signal of each symbol.
The transmitting apparatus and the receiving apparatus according to
As such, according to some implementations of the present disclosure, when the carrier frequency changes, the sequence used for phase compensation may also be changed, but the changed sequence is one of only 128 possible sequences or 128 × 3 possible sequences. Thus, in some implementations of the present disclosure, each of the transmitting side and the receiving side may store, for each available carrier frequency, a sequence consisting of phase compensation values to be applied to OFDM symbols corresponding to positive integer multiples of a period of a phase change, and perform phase compensation by applying the sequence corresponding to a specific carrier frequency in each period during processing of an OFDM symbol signal using the specific carrier frequency.
For example, if the phase of an OFDM symbol varies with a periodicity of 1ms and 14 OFDM symbols are included in a period of 1ms, a phase compensation value sequence for a specific carrier frequency consists of 14 phase compensation values for each of the 14 OFDM symbols. The transmitting side and the receiving side may store a phase compensation value sequence to be applied at an interval of 1ms for each carrier frequency, and may perform phase compensation on the corresponding carrier frequency using the stored phase compensation value sequence.
Implementation 2
Similar to
By comparison, in the
In contrast, in implementation 2, f is usedXXOf the basic carrier frequency fbase＝Nint128 Δ f to perform frequency shifting at the analog stage (e.g., frequency shifting with a free-running OSC), and using fXXOf (f)XX–NintThe 128 Δ f "performs frequency shifting at the digital level. When N is presentint128 deltaf, when used as up/down conversion frequency in a free running OSC, gives the same phase shift value to the OFDM symbol (i.e. the phase at the beginning of the signal part of the OFDM symbol is the same). Therefore, when N isintWhen 128 Δ f is used as the up/down conversion frequency in the free running OSC, it may not be necessary to calculate and apply a phase shift value every OFDM symbol for phase compensation.
Hereinafter, the frequency shift will be based on the frequency to be used for "f" in the digital stageXX-NintThe module of 128 Δ f "describes two examples of implementation 2 as implementation 2-1 and implementation 2-2.
Realization of 2-1
Fig. 6A and 6B are diagrams illustrating an example of an implementation 2-1 of the present disclosure. Specifically, fig. 6A shows an example of a part of a transmitting-side configuration according to implementation 2-1, and fig. 6B shows an example of a part of a receiving-side configuration according to implementation 2-1. In FIGS. 6A and 6B, tlRepresents the starting position of the signal part of the OFDM symbol l in the time domain and can be expressed as
When the final transmitter signal is summarized using equation 17, the transmission signal of the transmitter can be expressed by the following equation.
Formula 24
Referring to the last row of equation 24, Nfrac＝modulo(NTx128) is an entry for changing the mapping position of the resource (i.e., the mapping position of the subcarrier) in the IFFT entry. I.e. in terms corresponding to IFFT terms
Thus, in implementation 2-1, the mapping position of the resources with respect to the IFFT is changed by NfracFrequency shifting the baseband signal by fTxOf (f)Tx-Nint128 Δ f "(if Δ)offsetIf not 0, then "fTx-NintA portion of 128 Δ f "). Since the IFFT itself has a function of resetting the phase, the frequency shift performed by changing the resource mapping position with respect to the IFFT does not cause an OFDM symbolPhase mismatch between the signs. The term in the last row of equation 24
In some scenarios, it is often difficult to digitally implement large frequency shifts. In contrast, due to ΔoffsetIs small, so Δ can be easily realized by digital OSCoffsetThe frequency shift of (2). Performing delta after IFFToffsetThe frequency shift of (2). In the last line of equation 24, the expressionFrequency f inbaseAmong frequencies corresponding to integer multiples of 128 Δ f (e.g., less than or equal to fTxOf frequencies of (a) or greater than or equal to fTxOf frequencies of (a) or fTxOf the frequencies on both sides) closest to fTxOf (c) is detected. As an example, when Δ f is 15kHz, the frequency fbaseMay be an integer multiple of 1.92MHz, and when Δ f is 30kHz, the frequency fbaseAnd may be an integer multiple of 3.84 MHz. Expression formula
Referring to FIG. 6A, consider ΔoffsetIn addition to the IFFT and upconversion functions, a function corresponding to Δ may be performedoffsetThe frequency shift operation of (1). In some implementations, ΔoffsetMay have a variance of +/-5kHz, although implementations are not so limited. Furthermore, in the frequency ranges FR2 and FR1(>3GHz), or in the frequency range FR1 (R) ((R)<3GHz) mediumoffsetIn the case of 0, if Δoffset0, then the frequency shift operation may be performed without any additional operation (e.g., Δ)offsetFrequency shift of (d). In this case, the group shown in FIG. 6A may be omitted
Corresponding operations can be configured on the receiving side. The receiving side operation according to the implementation 2 will be described in detail below.
The final receiver signal summarized by
Formula 25
Referring to the last row of equation 25, Nfrac＝modulo(NRxAnd 128) is a term for changing a demapping position of the resource among FFT terms. I.e. in the FFT expressionItem of Chinese medicine
In some scenarios, it may be difficult to digitally implement very large frequency shifts. In contrast, due to ΔoffsetIs small, so Δ can be easily realized by digital OSCoffsetThe frequency shift of (2). Performing delta before FFToffsetThe frequency shift of (2). In the last line of equation 25, the expressionFrequency f inbaseAmong frequencies corresponding to integer multiples of 128 Δ f (e.g., less than or equal to fRxOf frequencies of (a) or greater than or equal to fRxOf frequencies of (a) or (f)RxOf the frequencies on both sides) closest to fRxOf (c) is detected. As an example, when Δ f is 15kHz, the frequency fbaseMay be an integer multiple of 1.92MHz, and when Δ f is 30kHz, the frequency fbaseAnd may be an integer multiple of 3.84 MHz. Expression formulaDenotes from fbaseThe down-conversion operation of (1). In implementation 2-1, f may be performed using analog OSCbaseThe frequency shift of (2).
In contrast, in implementation 2-1 described herein, by using "f" of the FFT as a digital blockRx-NintFrequency shift of 128 Δ f "and f using analog OSCbase＝NintFrequency shift of 128 Δ f, actually performed from the carrier frequency fRxIs shifted down in frequency.
Referring to FIG. 6B, in addition to FFT and down conversion functions, AND Δ may be performedoffsetCorresponding frequency shift operation. In addition, Δ in the frequency ranges FR2 and FR1 (> 3GHz), or in the frequency range FR1 (< 3GHz)offsetIn the case of 0, if ΔoffsetIf 0, then the frequency shift operation (e.g., Δ) may be performed without any additional operationoffsetFrequency shift of (d). In this case, the structure shown in FIG. 6B can be omittedA frequency shift module is shown.
Realization of 2-2
In implementation 2-2, carrier frequency f is performed using an analog OSC as in implementation 2-1 described aboveXXF of (a)baseThe frequency shift of (2). However, in implementation 2-2, f is performed by the digital OSCXXOf (f)XX-fbase"is performed using IFFT/FFT in implementation 2-1.
For example, the frequency shifting of the digital OSC may be performed by multiplying the signal by a cosine value or a sine value through the digital OSC. In this case, the digital oscillator may obtain the cosine/sine value by any suitable technique, for example by reading the cosine/sine value from a computer memory, or by calculating the cosine/sine value. In order for the phase of the signal to have a specific value at a specific point in time, the digital OSC only needs to be configured such that the address of the memory for a specific sample read by the digital OSC is the address of the memory storing the cosine/sine value that makes the phase a certain value. Alternatively, if the digital OSC is implemented to calculate the cosine/sine value instead of reading it from memory, the digital oscillator only needs to adjust the phase to the required value at a specific point in time. That is, the digital OSC may be implemented to read a memory address storing a frequency shift value having a specific phase value for a specific time/sampling point, or adjust the phase to a specific phase value for a specific time/sampling point. Such an implementation may therefore simplify the phase reset function based on OFDM symbol boundaries for frequency shifting by the digital OSC. In this scenario, some implementations may not require phase pre-compensation to be performed at the transmit end.
For reference, the phase mismatch between OFDM symbols may be due to the difference between the time interval in which IFFT/FFT is applied and the time interval during the free-running OSC operation. Implementing the up/down conversion frequency with a digital OSC, rather than a free running OSC that is an analog OSC, may make phase resetting at the boundaries of OFDM symbols easier and simpler. However, up/down conversion by the digital OSC to the extent of the carrier frequency may result in very high complexity, since the transmitter and receiver may need to perform multiplication in units of several GHz. For example, to perform up-conversion/down-conversion to/from 2GHz using a digital OSC, sampling should be performed in units of at least 4GHz according to the Nyquist sampling theorem. Therefore, the digital OSC must be implemented to multiply the input signal by the up-converted/down-converted cosine/sine value on a sample unit basis of 4 GHz. Implementing such a large number of multiplication operations with digital modules may be very complex in some scenarios and may result in increased manufacturing costs for the transmitter and receiver. Thus, in some scenarios, the overall amplitude of the carrier frequency is not up/down converted by the digital OSC.
By comparison, in implementation 2-1 above, the transmit side is configured to perform an up-conversion from the OSC of the RF stage (i.e., the free-running OSC) to fbaseProcessing and using of corresponding frequenciesfrac(or "f)Tx-fbase") the process of determining the resource mapping location in the IFFT. At aoffsetIn scenarios other than 0, the use of a digital OSC pair may also be performed for the transmit side according to implementation 2-1, as described aboveΔoffsetA step of resetting the phase (e.g., having zero phase at the end time of the cyclic prefix CP) symbol by OFDM symbol. In some scenarios of implementation 2-1 above, the signal output at baseband is in accordance with ffracValue of (or "f)Tx-fbase") may be asymmetrical with respect to DC. Spectral efficiency may be limited in the transmitter by filtering after the IFFT output (or in the receiver by before the FFT input). For example, a portion of the output of the IFFT (or the input of the FFT) may be outside the filtering region of the transmitter (or receiver), thereby causing signal distortion at the band edge due to filtering.
Thus, to address these issues, in some scenarios, it may be desirable to modify the resource mapping location implementation and f for through IFFT of implementation 2-1frac(or "f)Tx-fbase") the corresponding frequency up-conversion (i.e., frequency up-shift) and the transmit side operation of the phase reset function.
Thus, in implementation 2-2 described herein, to this end, on the transmit side, as in the case of frequency shifting in fig. 6A (i.e., delta is performed using a digital OSC)offsetFrequency shift of) and f are configured with a digital OSCfrac(or "f)Tx-fbase") the function of frequency shifting and phase resetting of the corresponding frequency.
Further examples of the above scenarios are provided below with reference to fig. 7A and 7B.
7A and 7B are diagrams illustrating examples of resource mapping for implementations 2-1 and 2-2, respectively, according to the present disclosure. In particular, FIG. 7A illustrates an example of resource mapping and upconversion according to implementation 2-1, and FIG. 7B illustrates an example of resource mapping and upconversion according to implementation 2-2.
Referring to the left part of fig. 7A, in some wireless communication systems, the information symbol a of a subcarrier in an OFDM symbolk(where k is a subcarrier index) is mapped to the IFFT module, and information symbols mapped to the IFFT module are distributed symmetrically with respect to the center or DC (approximately) of the IFFT module.
In contrast, referring to the right-hand portion of FIG. 7A, in implementation 2-1 of the present disclosure, akMapping at IFFTThe position being offset by, e.g., an amount Nfrac. In this case, in such a scenario that the resource mapping position of the IFFT changes as shown in the right part of fig. 7A, then a part of the output of the IFFT may be outside the filtering region of the transmitter. Since the portion of the output of the IFFT outside the filtering region will not be filtered, the signal may be distorted at the band edges.
To solve this problem, implementation 2-2, as shown in fig. 7B, follows the digital oscillation of the output of IFFT by f with the IFFT resource mapping positions unchangedfrac+Δoffset(i.e., "fTx-fbase") to a desired position (after appropriate filtering has been performed, if any).
Fig. 8A and 8B are diagrams illustrating an example of implementation 2-2 of the present disclosure. Specifically, fig. 8A shows an example of a part of a transmission-side structure according to implementation 2-2, and fig. 8B shows an example of a part of a reception-side structure according to implementation 2-2. In FIGS. 8A and 8B, the term tlRepresents the starting position of the signal part of the OFDM symbol l in the time domain and can be expressed as
Referring first to FIG. 8A, to follow fTxPerforming frequency up-conversion according to the transmitter pair of implementation 2-2TxF of (a)baseThe corresponding frequency is up-converted by an oscillator (i.e., an analog OSC) in the RF stage and is referred to ffrac+Δoffset(i.e., "fTx-fbase") the corresponding frequency is frequency up-converted and phase reset by the digital OSC OFDM symbol by OFDM symbol (e.g., to have a zero phase at the end time of the cyclic prefix CP). The operation of the receiving side according to implementation 2-2 described below may be performed in a similar manner as the transmitting side.
Referring to FIG. 8B, in order to follow fRxPerforming frequency down-conversion according to implementation 2-2 receiver pair andRxf of (a)baseThe corresponding frequency is down-converted by an oscillator (i.e., an analog OSC) in the RF stage andand is directed tofrac+Δoffset(i.e., "fRx-fbase") the corresponding frequency is frequency down-converted and phase reset by the digital OSC OFDM symbol by OFDM symbol (e.g., to have a zero phase at the end time of the cyclic prefix CP).
In contrast, in the above implementation 2-1, the receiving side is configured to perform: by the OSC of the RF stage (i.e., free-running OSC) to be coupled with fbaseProcedure for down-conversion of corresponding frequency and determination of use ffrac(or "f)Rx-fbase") the location of the resource demapping in the FFT. If ΔoffsetIs not 0, the receiving side according to the above implementation 2-1 may perform for deltaoffsetA process of resetting phase (e.g., to have zero phase at the end time of the cyclic prefix CP) OFDM symbol by OFDM symbol using digital OSC. In some scenarios of implementation 2-1 above, according to ffracValue of (or "f)Rx-fbase"), the degree of asymmetry of the signal output in the RF down-conversion step can be large relative to DC. This may limit spectral efficiency by filtering at the down-conversion output stage of the receiver.
Thus, to address these issues, in some scenarios, it may be desirable to modify the mapping of location realizations with f by FFT for implementation 2-1frac(or "f)Rx-fbase") corresponding frequency down conversion (i.e., frequency down shift) and phase reset.
Thus, in implementation 2-2, to this end, on the receive side, as in the case of frequency shifting in fig. 6B (i.e., delta is performed using a digital OSC)offsetFrequency shift of) and f are configured with a digital OSCfrac(or "f)Rx-fbase") the function of frequency shifting and phase resetting of the corresponding frequency. Here, the frequency f for frequency down-conversion and phase resetfrac(or "f)Rx-fbase") may be 128 (if Δ)offset128 by 3 depending on the frequency band instead of 0).
Realization 3
Fig. 9A to 9C are diagrams illustrating an example of implementation 3 of the present disclosure. Specifically, fig. 9A shows a diagram of an example of a part of the transmission-side configuration according to implementation 3, and fig. 9B and 9C show an example of a part of the reception-side configuration according to implementation 3.
The above equation 24 may be modified as follows.
Formula 26
Based on the last line of equation 26, implementation 3 of the present disclosure will be described. Implementation 3 is similar to implementation 2-1 in that, for example, N is implemented in implementation 3frac＝modulo(NTx128) performs the same function as in implementation 2-1. Referring to FIG. 9A, item Nfrac＝modulo(NTx128) is an entry for changing the mapping position of the resource (i.e., the mapping position of the subcarrier) in the IFFT entry. As described in the above implementation 2-1,corresponding to the IFFT entries. In the IFFT item, the IFFT is composed of,indicating the mapping location of the resources for the IFFT.
In the last row with reference to equation 26Implementation 3 differs from implementation 2-1 as follows. In the expressionOf medium frequency fbaseAmong frequencies corresponding to integer multiples of 128 Δ f (e.g., at or below f)TxOf frequencies greater than or equal to fTxOf frequencies of (a), or in fTxOf the two side frequencies) closest to fTxOf (c) is detected. As an example, when Δ f is 15kHz, the frequency fbaseMay equal an integer multiple of 1.92 MHz; when Δ f is 30kHz, frequency fbaseMay equal an integer multiple of 3.84 MHz; when Δ f ═At 60kHz, frequency fbaseMay be equal to an integer multiple of 7.68 MHz; and when Δ f is 120kHz, the frequency fbaseMay equal an integer multiple of 15.36 MHz. Frequency fbaseCan be expressed as Nint128 Δ f (wherein, NintIs an integer).
Expression formula
By comparison, the above implementation 2 is by targeting ΔoffsetThe frequency shift of (a) rotates the phase (i.e., shifts the phase) sample by sample to shift the frequency sample by sample. That is, implementation 2 above utilizes a complex value derived sample-by-sample to rotate the phase sample-by-sample. In contrast, implementation 3, as shown in fig. 9A, multiplies the output of the IFFT by a fixed complex value only OFDM symbol by OFDM symbol.
In addition, by comparison,
Implementation 3 applies to the receiving side as well as the transmitting side. Formula 25 may be modified as follows.
Formula 27
Based on the last line of equation 27, implementation 3 of the present disclosure will be described. Implementation 3 is similar to implementation 2-1 in that, for example, N is in both implementation 3 and implementation 2-1frac＝modulo(NRx128) serve the same purpose. Referring to FIGS. 9B and 9C, item Nfrac＝modulo(NRx128) is a term for changing the demapping position of the resource (i.e., the demapping position to the subcarrier) in the FFT term. As previously described with respect to implementation 2-1 above,corresponding to the FFT term. In the FFT term, it is possible to,indicating the demapping position of the resources (e.g., subcarriers) according to the FFT. In this way it is possible to obtain,
In the last row with reference to equation 27
Expression formula
By comparison, 2 passes for ΔoffsetThe frequency shift of (a) is sample-by-sample rotated in phase to shift the frequency sample-by-sample. That is, implementation 2 implements a complex value derived sample-by-sample to rotate the phase sample-by-sample. In contrast, implementation 3 simply multiplies the input of the FFT by a fixed complex value, OFDM symbol by symbol.
In addition, by comparison,
The above implementations (
Formula 28
Formula 29
In equation 28, the following equation holds: f. ofTx＝NTx*Δf+Δoffset＝Nint*128Δf+Nfrac*Δf+Δoffset＝fbase+Nfrac*Δf+Δoffset＝fbase+ffrac+Δoffset. Further, in equation 29, the following equation holds: f. ofRx＝NRx*Δf+Δoffset＝Nint*128Δf+Nfrac*Δf+Δoffset＝fbase+Nfrac*Δf+Δoffset＝fbase+ffrac+Δoffset. Let Δ beoffsetOFDM symbol signal generation/recovery according to the present disclosure performed for 0, except assuming ΔoffsetIs 0 without performing a pair of ΔoffsetMay be similarly applied to
In this scenario,
Implementation a1
Equation 19 associated with the transmitting side of
Formula 30
The phase compensation term of 1 is realized byGiven, but without taking Δ into accountoffsetThe phase compensation term for implementation a1 consists ofIt is given.
Among
Formula 31
The phase compensation term of 1 is realized by
E.g. for deltaoffsetWhich may be a band of-5 kHz, 0, or +5kHz, phase compensation to achieve 1 is performed using one of 128 x 3 complex-valued sequences. In contrast, the phase compensation to achieve a1 is performed using one of 128 complex-valued sequences, and ΔoffsetIs irrelevant. In implementation a1, Δ is performed even by a simulated free running OSCoffsetDoes not perform phase compensation to correct for ΔoffsetPossibly causing phase mismatch.
Implementation a2-1
Fig. 10A and 10B are diagrams illustrating an example of implementation a2-1 of the present disclosure. Specifically, FIG. 10A shows a portion of a transmit side structure according to implementation a2-1, while FIG. 10B shows a portion of a receive side structure according to implementation a 2-1.
Considering only dividing by ΔoffsetThe other components perform phase compensation, and equation 24 related to the transmission side implementing 2-1 may be modified to the following equation.
Formula 32
By mixing fbase+ffrac+ΔoffsetF in the first row of the formula 24TxThe first row of equation 32 may be obtained. As can be seen from equation 32, the phase compensation for a2-1 can be achieved by changing the IFFT termsLocation mapping of resources in (corresponding to items)) And is carried out by convertingoffsetIs added to fbaseThe frequency (i.e., f) obtainedbase+Δoffset) Up-conversion of (corresponding to item))，fbaseAmong frequencies corresponding to integer multiples of 128 Δ f (e.g., less than or equal to fTxOf frequencies of (a) or greater than or equal to fTxOf frequencies of (A), orTxFrequency of both sides of (d) is closest to fTxOf (c) is detected. Therefore, if the pair Δ is not performedoffsetThe up-conversion to the carrier frequency can be performed by resource mapping offset and up-conversion for IFFT (using analog OSC) as shown in fig. 10A. In implementation a2-2, Δ is performed even by a simulated free-running OSCoffsetFrequency shift of, ΔoffsetAlso considered to be 0, i.e. not performing a pair ΔoffsetPhase correction/compensation.
Similarly, consider the pair onlyDivided by DeltaoffsetThe other components are phase-compensated, and thus equation 25 related to realizing the receiving side in 2-1 can be modified as follows.
Formula 33
Can be obtained by mixing fbase+ffrac+ΔoffsetF in the first row of the formula 25RxTo obtain the first row of equation 33. As can be seen from equation 33, implementing phase compensation for 2-1a involves changing the FFT term
Implementation a2-2
Fig. 11A and 11B are diagrams illustrating an example of implementation a2-2 of the present disclosure. Specifically, FIG. 11A shows a portion of a transmit side structure according to implementation a2-2, while FIG. 11B shows a portion of a receive side structure according to implementation a 2-2. In FIG. 11A and FIG. 11B, the term tlRepresents the starting position of the signal part of the OFDM symbol l in the time domain and can be expressed as
With reference first to figure 11A,showing a portion of the transmission in implementation a2-2, offset ΔoffsetIs considered to be 0, i.e. does not perform a pair of ΔoffsetPhase correction/compensation. That is, even with a simulated free running OSC in accordance with fbase+ΔoffsetCarry out frequency shift, and do not carry out aoffsetPhase correction/compensation.
By comparison, in the implementation a2-1 described above, on the transmitter side, fbaseAnd ΔoffsetThe corresponding frequency is used for up-conversion processing by the RF stage OSC (free running OSC) of the transmitting side, and is equal to ffracThe corresponding part is used in the process of determining the resource mapping position with respect to the IFFT. However, in some scenarios of implementation a2-1 described above, according to ffracThe asymmetry of the signal output at baseband may increase relative to DC. This may limit spectral efficiency by filtering after the IFFT output in the transmitter (or before the FFT input in the receiver).
Therefore, to address this issue, in some scenarios, it may be desirable to modify the resource mapping location implementation to f by adjusting the IFFT of implementation a2-1 abovefracFrequency up-conversion (i.e., f)fracUp shifting the frequency) and phase reset.
Thus, in implementation a2-2, f can be implemented by a digital OSC for this purpose on the transmitter sidefracFrequency shift and phase reset functions. Thus, as shown in FIG. 11A, in order to follow fTxPerform frequency up-conversion to achieve a2-2 to be equal to fTxF in (1)baseAnd ΔoffsetThe corresponding frequency is used for up-conversion performed by the free-running OSC of the RF stage, and f is convertedfracFor performing frequency up-conversion and reset phase (e.g., to have zero phase at the end time of the cyclic prefix CP) on an OFDM symbol-by-OFDM symbol basis through digital OSC.
Referring to FIG. 11B, a portion of the receive side in implementation a2-2 is shown with an offset ΔoffsetIs considered to be 0, i.e. does not perform a pair of ΔoffsetPhase correction/compensation. Thus, even if f is performed by a simulated free running OSCbase+ΔoffsetNor does it perform a shift of deltaoffsetPhase correction/compensation.
By comparison, in the implementation a2-1 described above, on the receiving side, fbaseAnd ΔoffsetThe corresponding frequency is used for the down-conversion process by the OSC (free running OSC) of the RF stage on the receiving side, andfracthe corresponding part is used for the process of determining the resource demapping position with respect to the FFT. However, in some scenarios of implementation a2-1 described above, f is dependentfracThe asymmetry of the output signal (i.e., the RF output) in the down-conversion may increase relative to DC. This may limit the efficiency of the spectrum by filtering after the down-converted output (or before the FFT input in the receiver).
Therefore, to address these issues, in some scenarios it may be desirable to modify the implementation of a2-1 above for implementing and f by adjusting the FFT resource mapping locationfracThe receiving side of the corresponding function of frequency down-conversion (i.e., frequency down-shift) and phase reset operates.
Thus, in implementation a2-2, to this end, on the receive side, the implementation by f may be through a digital OSCfracFrequency shift and phase reset functions. Thus, as shown in FIG. 11B, in order to follow fRxPerform frequency down-conversion to achieve a2-2 to be in common with fTxF in (1)baseAnd ΔoffsetThe corresponding frequency is used for down-conversion by a free running OSC (analog OSC) of the RF stage, and f is convertedfracFor performing frequency down-conversion and resetting phase (e.g., to have zero phase at the end time of the cyclic prefix CP) OFDM symbol by digital OSC.
To summarize,
In implementation 2, in implementation 2-1And use of integer multiples of 128 deltaf as the base carrier frequency f in both implementations 2-2baseAnd is regulated by the free-running OSC of the RF stage according to fbaseA frequency up/down shift is performed. In implementation 2, the transmitting side compensates f by resource mapping for IFFTTxAnd fbaseThe frequency difference between (implementation 2-1) or by frequency shifting using a digital OSC after IFFT (implementation 2-2). Implementation 2's receive side compensates for f by resource demapping for FFTTxAnd fbaseThe frequency difference between (implementation 2-1) or by frequency shifting using a digital OSC before the FFT (implementation 2-2).
Except for and for deltaoffsetImplementation 2 of the frequency shift resetting the phase sample by sample differs from implementation 3 of the frequency shift resetting the phase by OFDM symboloffsetImplementation 3 is similar to implementation 2, except that phase compensation is performed.
Fig. 12 is a diagram illustrating another use example of the present disclosure.
As described above, in order to prevent phase mismatch between OFDM symbols or facilitate phase compensation, the analog OSC uses a frequency corresponding to an integer multiple of 128 Δ f in OFDM symbol signal generation or OFDM symbol signal recovery. For example, when changing the parameter set of a frequency band supporting a plurality of parameter sets, for example, when the subcarrier spacing (SCS) of the frequency band is changed from 30kHz to 15kHz, and vice versa, frequencies that are integer multiples of 128 Δ f may be mismatched. For example, assume that when Δ f is 30kHz, f is closest to fTx128 Δ f corresponds to a frequency f of an integer multiple ofbase，1(wherein f is 30kHz for Δ fbase，1＝Nint，1× 128 Δ f) if SCS is on fTxIn the same frequency band, when the frequency changes from Δ f to 30kHz to Δ f to 15kHz, the frequency is closest to the upconversion frequency f when Δ f is 15kHzTxCorresponding to an integer multiple of 128 deltafbase，0(where f is 15kHz for Δ fbase，0＝Nint128 Δ f) may be compared with fbase，1Phase difference Δ fbase. Alternatively, the up/down conversion frequency may be varied by Δ f according to the change in SCSbase. In this case, the transmitter and receiver of the present disclosure may use the method according toThe digital OSC or IFFT/FFT resource mapping/demapping of implementation 2 of the present disclosure described above compensates for Δ fbaseRather than performing RF retuning.
Fig. 13A and 13B illustrate examples of transmitter structures and receiver structures according to the present disclosure. The transmission-side structure and the reception-side structure of the present disclosure are described based on the basic structures of fig. 13A and 13B.
Referring to fig. 13A, the transmitter generates symbols (hereinafter, referred to as information symbols) for an input bit sequence according to a signal generation technique defined in, for example, the standard. The transmitter performs appropriate resource mapping (i.e., subcarrier mapping) of the generated information symbols according to a frequency band in which transmission is to be performed on the input side of the IFFT, and performs IFFT of the frequency domain signal into a time domain signal on the resource-mapped information symbols. The transmitter inserts a Cyclic Prefix (CP) configured to mitigate/avoid interference between OFDM symbols into the IFFT output. For reference, fig. 5A to 11B illustrate that the IFFT/FFT includes a resource mapping/demapping function and a CP attaching/splitting function, which may be implemented separately from the FFT/IFFT, as shown in fig. 13. For signals generated by IFFT and CP attachment, the transmitter may perform filtering or windowing to meet spectral characteristics before performing up-conversion to a carrier frequency. However, depending on the characteristics of the RF device, filtering or windowing may not be a necessary function. In order to transmit the signal generated via the IFFT and CP attachment or the signal generated via the IFFT and CP attachment using a predetermined carrier frequency (and filtering/windowing), the transmitter performs up-conversion of the signal to the predetermined carrier frequency. Generally, up-conversion is performed using a digital-to-analog converter (DAC) for converting a digital signal into an analog signal, an oscillator and a PLL (phase locked loop) for generating a carrier frequency, a mixer for moving a baseband signal to a desired carrier frequency, and the like. Thereafter, the transmitter transmits the up-converted signal to the outside through an analog filter, an amplifier, and an antenna.
Since the signal input to the digital-to-analog converter in the transmitter is a digital signal and the signal output from the digital-to-analog converter is an analog signal, the transmitter module used for signal processing before the digital-to-analog converter may be a digital module and the transmitter module used for signal processing after the digital-to-analog converter may be an analog module.
The receiver performs an operation corresponding to the inverse process of the transmitter. In receiver operation, a signal transmitted by a transmitter is received by the receiver through an antenna, an amplifier, and an analog filter of the receiver. Referring to fig. 13B, the receiver performs down-conversion on the received signal. In general, down-conversion is performed using an analog-to-digital converter (ADC) for converting an analog signal into a digital signal, an oscillator and a PLL for generating a carrier frequency, a mixer for shifting a signal received through the carrier frequency to a band signal, and the like. The receiver may filter the signal transmitted through the baseband according to the spectral characteristics. Depending on the characteristics of the RF device, filtering may not be implemented. The receiver splits a Cyclic Prefix (CP) from a (filtered or unfiltered) baseband signal according to pre-measured timing information and converts the split CP signal into a frequency domain signal through an FFT for converting a time domain signal into a frequency domain signal. The FFT function includes a resource demapping function for deriving only the signal sent to the receiver from the entire frequency domain signal. The receiver recovers the signal transmitted by the transmitter from the resource-demapped signal through a symbol recovery process for compensating for a distorted portion on the channel, performs a decoding process with respect to a specific signal generation technique (for example, a signal generation technique defined according to a communication standard), and then obtains a final signal (bit sequence). Both the process of compensating for the distorted part on the channel and the decoding process correspond to a symbol recovery process.
Since the signal input to the analog-to-digital converter in the receiver is an analog signal and the signal output from the analog-to-digital converter is a digital signal, the receiver module for signal processing before the analog-to-digital converter may be an analog module and the receiver module for signal processing after the analog-to-digital converter may be a digital module.
Although not shown in fig. 13A and 13B, the transmitter and receiver may include digital oscillators configured to perform operations according to the present disclosure.
Fig. 14 is a block diagram illustrating an example of elements of the transmitting
The
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In some scenarios of the present disclosure, the functions, processes, and/or methods disclosed in the present disclosure may be implemented by a processing chip (also referred to as a processing device). The processing chip may be a system on chip (SoC). The processing chip may include the processor 11 and/or 21 and the
The processor 11 installed on the transmitting
The signal processing process of the
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In the present disclosure, a User Equipment (UE) (i.e., a terminal) operates as a transmitting
The processor 11 mounted on the transmitting
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Although the transmitting
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