阅读说明：本技术 用于无线遥测的极化分集接收方法 (Polarization diversity reception method for wireless telemetry ) 是由 陈炜炜 刘兆辉 迟东明 王燕玲 朱敏 于 2020-06-15 设计创作，主要内容包括：本发明的用于无线遥测的极化分集接收方法,左、右旋信号分别经带通滤波、数字AGC、数字正交下变频得到同相、正交两路基带信号；左旋两路基带信号和右旋两路基带信号送至差模环鉴相器进行鉴相,得到相位差信号,经环路滤波后反馈至左右两路数字正交下变频中的本地NCO；计算左旋信号和右旋信号的加权比,根据加权比对左旋同相基带信号和右旋同相基带信号进行最大比值合成、对左旋正交基带信号和右旋正交基带信号进行最大比值合成,得到合成后的同相基带信号和正交基带信号；该合成后的同相基带信号和正交基带信号经叉积鉴频、双向峰值检波、环路滤波后,得到频差信号,并反馈至左右两路数字正交下变频中的本地NCO。(The invention relates to a polarization diversity receiving method for wireless remote measurement.A left-handed signal and a right-handed signal are respectively subjected to band-pass filtering, digital AGC and digital quadrature down-conversion to obtain an in-phase baseband signal and an orthogonal baseband signal; the left-handed two paths of baseband signals and the right-handed two paths of baseband signals are sent to a differential mode loop phase discriminator to be subjected to phase discrimination to obtain phase difference signals, and the phase difference signals are subjected to loop filtering and then fed back to local NCO in the left-handed and right-handed digital orthogonal down-conversion; calculating the weighting ratio of the left-handed signal and the right-handed signal, performing maximum ratio synthesis on the left-handed in-phase baseband signal and the right-handed in-phase baseband signal according to the weighting ratio, and performing maximum ratio synthesis on the left-handed orthogonal baseband signal and the right-handed orthogonal baseband signal to obtain a synthesized in-phase baseband signal and a synthesized orthogonal baseband signal; the synthesized in-phase baseband signal and quadrature baseband signal are processed by cross product frequency discrimination, bidirectional peak detection and loop filtering to obtain a frequency difference signal, and the frequency difference signal is fed back to a local NCO in the left and right digital quadrature down-conversion circuits.)

1. The polarization diversity receiving method for wireless remote measurement is characterized in that the left-handed signal is subjected to band-pass filtering, digital AGC and digital quadrature down-conversion to obtain in-phase baseband signals and quadrature baseband signals; the right-handed signal is subjected to band-pass filtering, digital AGC and digital quadrature down-conversion to obtain in-phase and quadrature baseband signals; the left-handed two paths of baseband signals and the right-handed two paths of baseband signals are sent to a differential mode loop phase discriminator to be subjected to phase discrimination to obtain phase difference signals, and the phase difference signals are subjected to loop filtering and then fed back to local NCO in the left-handed digital orthogonal down-conversion and the right-handed digital orthogonal down-conversion; calculating the weighting ratio of the left-handed signal and the right-handed signal, performing maximum ratio synthesis on the left-handed in-phase baseband signal and the right-handed in-phase baseband signal according to the weighting ratio, and performing maximum ratio synthesis on the left-handed orthogonal baseband signal and the right-handed orthogonal baseband signal to obtain a synthesized in-phase baseband signal and a synthesized orthogonal baseband signal; and after the synthesized in-phase baseband signal and quadrature baseband signal are subjected to cross product frequency discrimination, bidirectional peak detection and loop filtering, a frequency difference signal is obtained and fed back to a local NCO in the left-handed digital quadrature down-conversion and the right-handed digital quadrature down-conversion.

2. The polarization diversity reception method for wireless telemetry of claim 1 wherein the local NCO in each of the left-hand digital quadrature downconversion and the right-hand digital quadrature downconversion is controlled by a local carrier signal, a phase difference signal and a frequency difference signal to eliminate phase and frequency differences between the left-hand and right-hand.

3. The polarization diversity reception method for wireless telemetry of claim 1 wherein the bandpass filtering is implemented using a configurable universal intermediate frequency bandpass filter that selects a bandwidth from a plurality of bandwidths to filter out-of-band noise based on a desired code rate.

4. The polarization diversity reception method for wireless telemetry of claim 1 wherein the digital AGC control employs a fixed step gain digital AGC and the AGC feedback time is determined by an AGC time constant.

5. The polarization diversity reception method for wireless telemetry according to claim 1, wherein the differential mode loop phase detector uses a cross-correlation based phase detection algorithm, and the differential mode loop phase detector sends the obtained phase difference signal to the local NCO in the left-hand digital orthogonal down-conversion and the right-hand digital orthogonal down-conversion in opposite polarities, so that the left and right signals are pushed to a same frequency and phase in opposite directions.

6. The polarization diversity reception method for wireless telemetry according to claim 1, wherein the left-hand signal-to-noise ratio is calculated using the left-hand in-phase baseband signal and the quadrature baseband signal output from the left-hand digital down-converter, and the right-hand signal-to-noise ratio is calculated using the right-hand in-phase baseband signal and the quadrature baseband signal output from the right-hand digital down-converter; calculating the weighting coefficients of the left-handed signal and the right-handed signal according to the signal-to-noise ratio of the left-handed signal and the signal-to-noise ratio of the right-handed signal, so as to obtain the weighting ratio of the left-handed signal and the right-handed signal; and performing maximum ratio synthesis on the left-handed in-phase baseband signal and the right-handed in-phase baseband signal by using weighting comparison to obtain a synthesized in-phase baseband signal I, and performing maximum ratio synthesis on the left-handed orthogonal baseband signal and the right-handed orthogonal baseband signal by using weighting comparison to obtain a synthesized orthogonal baseband signal Q.

7. The polarization diversity reception method for wireless telemetry of claim 5 wherein the signal-to-noise ratio estimation algorithm uses a second order fourth moment method.

Technical Field

The invention relates to wireless communication, in particular to a polarization diversity receiving method for wireless telemetry.

Background

With the rapid development of communication technology, a great deal of research has been conducted on the channel characteristics of wireless communication, and various techniques have been developed to overcome multipath fading, such as: diversity techniques, coding and interleaving techniques, adaptive equalization techniques, and the like.

Diversity techniques are applied to telemetry systems to improve the tracking and data reception performance of the telemetry system. Diversity is classified into polarization diversity, space diversity, and time diversity. Space diversity and time diversity have not been widely used due to the poor reliability of time segmentation when conditional constraints and tracking dynamic targets. Polarization diversity is achieved by optimally ratio-weighting two mutually orthogonal polarization signals (left-hand and right-hand or vertical and horizontal) such that their output signal-to-noise ratio is improved. Polarization mismatch can be improved by 2-3 dB by polarization diversity. The polarization diversity can overcome the influence of deep zero point existing in single polarization of the antenna, and can improve dozens of dB or more.

The combining mode of diversity reception is the key to realize diversity reception, and the maximum ratio combining mode controls the phase and amplitude of the received signals according to the principle that the SNR after combination is the maximum. The polarization diversity maximum ratio synthesis technology has very important significance in telemetering and receiving aircrafts with serious signal polarization attenuation, such as a fast flying target, a low elevation angle moving target, a rotary moving target and the like, not only can enable the system to obtain diversity gain, but also can ensure the uninterrupted reception of telemetering signals as long as a signal with one rotary direction is higher than a receiving threshold, and greatly improves the working reliability of the system.

In order to meet the requirements of high sensitivity, diversity synthesis and high signal-to-noise ratio, the traditional analog telemetry diversity receiver adopts the technologies of multi-stage analog device amplification, filtering, multi-loop circuit and the like, which inevitably causes the defects of complex circuit, easy drift of working points, large internal noise, low stability and the like. And the anti-interference performance of the analog signals is poorer than that of the digital signals.

Disclosure of Invention

The invention aims to provide a polarization diversity receiving method for wireless telemetry, which realizes digital telemetry diversity receiving.

In order to achieve the aim, the invention provides a polarization diversity receiving method for wireless telemetry, wherein a left-handed signal is subjected to band-pass filtering, digital AGC and digital quadrature down-conversion to obtain in-phase and quadrature baseband signals; the right-handed signal is subjected to band-pass filtering, digital AGC and digital quadrature down-conversion to obtain in-phase and quadrature baseband signals; the left-handed two paths of baseband signals and the right-handed two paths of baseband signals are sent to a differential mode loop phase discriminator to be subjected to phase discrimination to obtain phase difference signals, and the phase difference signals are subjected to loop filtering and then fed back to local NCO in the left-handed digital orthogonal down-conversion and the right-handed digital orthogonal down-conversion; calculating the weighting ratio of the left-handed signal and the right-handed signal, performing maximum ratio synthesis on the left-handed in-phase baseband signal and the right-handed in-phase baseband signal according to the weighting ratio, and performing maximum ratio synthesis on the left-handed orthogonal baseband signal and the right-handed orthogonal baseband signal to obtain a synthesized in-phase baseband signal and a synthesized orthogonal baseband signal; and after the synthesized in-phase baseband signal and quadrature baseband signal are subjected to cross product frequency discrimination, bidirectional peak detection and loop filtering, a frequency difference signal is obtained and fed back to a local NCO in the left-handed digital quadrature down-conversion and the right-handed digital quadrature down-conversion.

According to the polarization diversity receiving method for wireless telemetry, NCO in the left-hand digital orthogonal down-conversion and the right-hand digital orthogonal down-conversion are controlled through the local carrier signal, the phase difference signal and the frequency difference signal, so that the phase difference and the frequency difference between the left-hand side and the right-hand side are eliminated.

The polarization diversity receiving method for wireless telemetry is characterized in that the band-pass filtering is realized by adopting a configurable universal intermediate frequency band-pass filter, and the bandwidth is selected from multi-level bandwidths according to the required code rate to filter out-of-band noise.

The polarization diversity reception method for wireless telemetry described above, wherein the digital AGC control employs a fixed step gain digital AGC, and the AGC feedback time is determined by an AGC time constant.

The polarization diversity receiving method for wireless telemetry is characterized in that the differential mode loop phase discriminator adopts a phase discrimination algorithm based on cross correlation, and the differential mode loop phase discriminator sends the obtained phase difference signals to the left-hand digital orthogonal down-conversion and the right-hand digital orthogonal down-conversion in opposite polarities, so that the left-hand signals and the right-hand signals are pushed to the same frequency and phase in opposite directions.

The polarization diversity receiving method for wireless telemetry comprises the steps of calculating a signal-to-noise ratio of a left-handed signal by using a left-handed in-phase baseband signal and a quadrature baseband signal output by a left-handed digital down converter, and calculating a signal-to-noise ratio of a right-handed signal by using a right-handed in-phase baseband signal and a quadrature baseband signal output by a right-handed digital down converter; calculating the weighting coefficients of the left-handed signal and the right-handed signal according to the signal-to-noise ratio of the left-handed signal and the signal-to-noise ratio of the right-handed signal, so as to obtain the weighting ratio of the left-handed signal and the right-handed signal; and performing maximum ratio synthesis on the left-handed in-phase baseband signal and the right-handed in-phase baseband signal by using weighting comparison to obtain a synthesized in-phase baseband signal I, and performing maximum ratio synthesis on the left-handed orthogonal baseband signal and the right-handed orthogonal baseband signal by using weighting comparison to obtain a synthesized orthogonal baseband signal Q.

The polarization diversity receiving method for wireless telemetry adopts a second-order fourth-order moment method for a signal-to-noise ratio estimation algorithm.

Compared with the prior art, the invention has the beneficial technical effects that:

the polarization diversity receiving method for wireless remote measurement adopts a maximum ratio combining mode to realize that the maximum gain of two paths of diversity receiving is more than 2.5dB, and automatic gain control is added in the diversity receiving, so that the AGC control range of the input signal intensity of a receiver is improved;

the polarization diversity receiving method for wireless remote measurement is used for receiving wireless remote measurement signals, not only has stable performance, but also can easily change the modulation and demodulation mode of wireless channel transmission, adapt to different transmission code rates, adapt to remote measurement communication of different frequency bands and the like by embedding different digital processing software or changing related software settings, and has the flexibility characteristic of adapting to wireless multi-band multi-mode.

Drawings

The polarization diversity reception method for wireless telemetry according to the present invention is provided by the following embodiments and accompanying drawings.

Fig. 1 is a schematic diagram of a polarization diversity reception method for wireless telemetry according to a preferred embodiment of the present invention.

Fig. 2 is a block diagram of the digital AGC design in the preferred embodiment of the present invention.

FIG. 3 is a block diagram of a digital down conversion implementation in a preferred embodiment of the present invention.

FIG. 4 is a schematic diagram of a differential mode loop in a preferred embodiment of the present invention.

Fig. 5 is a block diagram of a cross-correlation phase detection implementation in a preferred embodiment of the invention.

Fig. 6 is a schematic diagram of a digital loop filter in accordance with a preferred embodiment of the present invention.

FIG. 7 is a diagram illustrating dual peak detection in accordance with a preferred embodiment of the present invention.

Detailed Description

The polarization diversity reception method for wireless telemetry according to the present invention will be described in further detail with reference to fig. 1 to 7.

The invention relates to a polarization diversity receiving method for wireless remote measurement.A symmetrical diversity phase-locked loop is used for eliminating the frequency difference and the phase difference of each branch signal at a merging point before a left-hand signal and a right-hand signal are synthesized according to the maximum ratio; after the frequency difference and the phase difference are eliminated, the weighting ratio is calculated according to the amplitude of the input signal, and finally, the maximum ratio synthesis is realized.

The left-handed signal is subjected to band-pass filtering, digital AGC and digital quadrature down-conversion to obtain in-phase baseband signals and quadrature baseband signals; the right-handed signal is subjected to band-pass filtering, digital AGC and digital quadrature down-conversion to obtain in-phase and quadrature baseband signals; the left-handed two paths of baseband signals and the right-handed two paths of baseband signals are sent to a differential mode loop phase discriminator to be subjected to phase discrimination to obtain phase difference signals, and the phase difference signals are subjected to loop filtering and then fed back to local NCO in the left-handed digital orthogonal down-conversion and the right-handed digital orthogonal down-conversion; calculating the weighting ratio of the left-handed signal and the right-handed signal, performing maximum ratio synthesis on the left-handed in-phase baseband signal and the right-handed in-phase baseband signal according to the weighting ratio, and performing maximum ratio synthesis on the left-handed orthogonal baseband signal and the right-handed orthogonal baseband signal to obtain a synthesized in-phase baseband signal and a synthesized orthogonal baseband signal; after the synthesized in-phase baseband signal and quadrature baseband signal are subjected to cross product frequency discrimination, bidirectional peak detection and loop filtering, a frequency difference signal is obtained and fed back to a local NCO in the left-handed digital quadrature down-conversion and the right-handed digital quadrature down-conversion; and the local NCO in the left-hand digital orthogonal down-conversion and the right-hand digital orthogonal down-conversion is controlled by a local carrier signal, a phase difference signal and a frequency difference signal so as to eliminate the phase difference and the frequency difference between the left hand and the right hand.

Fig. 1 is a schematic diagram of a polarization diversity reception method for wireless telemetry according to a preferred embodiment of the present invention.

As shown in fig. 1, the module structure in this embodiment includes: the device comprises a left-handed intermediate frequency band-pass filter, a left-handed digital AGC module, a left-handed digital down converter, a left-handed signal-to-noise ratio calculation unit, a right-handed intermediate frequency band-pass filter, a right-handed digital AGC module, a right-handed digital down converter, a right-handed signal-to-noise ratio calculation unit, a differential mode loop phase discriminator, a first loop filter, a common mode loop frequency difference removal module (comprising a cross product frequency discriminator, a double-peak detector and a second loop filter) and a weight coefficient calculation unit.

The left-handed signal firstly passes through a left-handed intermediate frequency band-pass filter to filter out-of-band frequency components, and then enters a left-handed digital AGC module to stabilize the signal amplitude within a certain range; the left-handed signal adjusted by AGC is converted by left-handed digital quadrature down-conversion to obtain in-phase baseband signals and quadrature baseband signals, namely left-handed in-phase baseband signals and left-handed quadrature baseband signals. The right-handed signal is firstly filtered out of band frequency components by a right-handed intermediate frequency band-pass filter, and then enters a right-handed digital AGC module to stabilize the signal amplitude within a certain range; the right-hand signal adjusted by AGC is converted by right-hand digital quadrature down-conversion to obtain in-phase and quadrature baseband signals, namely a right-hand in-phase baseband signal and a right-hand quadrature baseband signal.

In this embodiment, the intermediate frequency band pass filter is a configurable general intermediate frequency band pass filter, and completes band pass filtering of the input 70MHz intermediate frequency signal. Different tasks require different code rates, and thus occupy different bandwidths. The universal intermediate frequency band-pass filter can be configured to select a proper bandwidth from 14-gear bandwidths to filter out-of-band noise according to the required code rate, wherein the 14-gear bandwidths are 300kHz, 500kHz, 750kHz, 1.0MHz, 1.5MHz, 2.4MHz, 3.3MHz, 4MHz, 6MHz, 8MHz, 10MHz, 15MHz, 20MHz and 24 MHz. The configurable universal intermediate frequency band-pass filter ensures that intermediate frequency useful signals pass through and filters out-of-band useless signals.

Fig. 2 is a block diagram of the digital AGC design in accordance with the preferred embodiment of the present invention.

When diversity reception is utilized to process received signals, the digital AGC further adjusts the sampled signals to improve the quality of the received signals, so that the amplitudes of the left and right signals are consistent, and signals with consistent amplitudes are provided for differential-mode loop phase discrimination.

The digital AGC control of the present embodiment employs a fixed step gain digital AGC design, with the AGC feedback time determined by the AGC time constant. Referring to fig. 2, after the signal is subjected to the intermediate frequency band pass filtering, the signal is subjected to amplitude control by the gain control module, and the factors affecting the amplitude control, namely, the gain _ pulse signal and the direction signal, are derived from the gain feedback module, the gain _ pulse signal is used for indicating whether the gain adjustment is performed, and the direction signal is used for indicating whether the gain should be adjusted in the large direction or the small direction. The gain feedback is mainly based on the comparison between the modulus of the branch signal and the set threshold, so as to generate a gain _ pulse signal and a direction signal, and the agc time constant is used for controlling the time of each feedback. The module value of the branch signal can be obtained by squaring, adding and then calculating the root number of the signals of the in-phase and orthogonal paths obtained after the branch frequency down-conversion.

The main function of digital down-conversion (DDC) is to frequency convert the incoming intermediate frequency signal to zero intermediate frequency, resulting in an in-phase component and a quadrature component of the baseband signal, as shown in fig. 3. In this embodiment, the digital down converter includes a digital mixer, a local NCO, and a low pass filter. Two paths of orthogonal local oscillation signals (cos and sin components) generated by the local NCO and an input signal (output of the digital AGC module) are subjected to frequency mixing through a digital mixer, and the frequency-mixed signal is subjected to low-pass filter to filter out a frequency multiplication component and out-of-band noise.

The signal sources for controlling the local NCO are three: first, a local carrier signal; the four paths of baseband signals are subjected to cross-correlation phase discrimination to obtain the phase difference of two paths of signals, the four paths of baseband signals are subjected to loop filtering and divided into two paths, one path of baseband signals is in opposite phase, and the two paths of baseband signals are pulled to the same phase from opposite directions to obtain the same-frequency and same-phase signals; and thirdly, performing cross product frequency discrimination on a combined branch obtained after the maximum ratio is synthesized by using an outer ring common mode loop (AFC), performing peak detection on a cross product frequency discrimination result to obtain a frequency deviation error signal (namely a frequency difference signal), filtering the error signal by using a loop, and dividing the error signal into two paths to control the local NCO from the same direction. Therefore, two paths of signals with the same frequency and phase are obtained before the maximum ratio synthesis.

And the left-handed in-phase baseband signal, the left-handed orthogonal baseband signal, the right-handed in-phase baseband signal and the right-handed orthogonal baseband signal are sent to a differential mode loop phase discriminator to be subjected to phase discrimination to obtain a phase difference so as to control the local NCO. In this embodiment, the differential mode loop phase detector sends the obtained phase difference to the local NCO in the left and right channels with opposite polarities, so that the left and right signals are pushed to the same frequency and phase in opposite directions, and the principle is as shown in fig. 4.

In this embodiment, the differential mode loop phase detection adopts a phase detection algorithm based on cross-correlation, as shown in formula (1) and formula (2):

I'_L*I'_R+Q'_L*Q'_R＝A²cos[(ω_R-ω_L)t+θ_R-θ_L]+n_I(1)

Q'_L*I'_R-I'_L*Q'_R＝A²sin[(ω_R-ω_L)t+θ_R-θ_L]+n_R(2)

as shown in the figure, the signals are cross-correlated and then calculated by an arctan method to obtain a phase difference signal (ω)_R-ω_L)t+θ_R-θ_L。

The phase difference signal obtained by the differential mode loop phase discrimination is also sent to a first loop filter, and the first loop filter averages the phase difference signal containing ripples output by the differential mode loop phase discriminator, so that the phase difference signal is changed into low-pass filtering output containing few alternating current components and direct current signals. The first loop filter determines the transmission characteristics of the entire PLL circuit, and further determines characteristics (loop control) such as the stability, the capture bandwidth, and the capture speed of the circuit. The digital loop filter (first loop filter) structure is shown in fig. 6.

Calculating the signal-to-noise ratio of a left-handed signal by using a left-handed in-phase baseband signal and a quadrature baseband signal output by the left-handed digital down converter, and calculating the signal-to-noise ratio of a right-handed signal by using a right-handed in-phase baseband signal and a quadrature baseband signal output by the right-handed digital down converter; calculating the weighting coefficients of the left-handed signal and the right-handed signal according to the signal-to-noise ratio of the left-handed signal and the signal-to-noise ratio of the right-handed signal, so as to obtain the weighting ratio of the left-handed signal and the right-handed signal; and performing maximum ratio synthesis on the left-handed in-phase baseband signal and the right-handed in-phase baseband signal by using weighting comparison to obtain an in-phase baseband signal I, and performing maximum ratio synthesis on the left-handed orthogonal baseband signal and the right-handed orthogonal baseband signal by using weighting comparison to obtain an orthogonal baseband signal Q.

In this embodiment, the purpose of the common-mode ring frequency offset removal module is to remove frequency offset, and the common-mode ring frequency offset removal module includes a cross product discriminator, a dual peak detector, and a second loop filter. The in-phase baseband signal I and the orthogonal baseband signal Q synthesized by the maximum ratio are sent to a cross product frequency detector for frequency detection, a frequency deviation component is estimated from a frequency detection result, and then the frequency deviation component is sent to a local NCO in a left-handed digital down converter and a right-handed digital down converter after noise is filtered by a second loop filter, so that the output frequency of the local NCO is changed, and the frequency deviation component in the frequency detection result is eliminated in the digital down converter.

The common-mode loop frequency deviation removing module obtains a frequency deviation signal by adopting a double-peak detection method. The term "double peak" means a maximum value and a minimum value, and the principle of double peak detection is to detect the maximum value and the minimum value of a signal in a piece of signal data having a length of N, and to use the average value of the maximum value and the minimum value as an estimated value of the frequency offset component of the data in the piece of data at the current time. The AFC time constant is used to control the length of this period, the maximum and minimum values of the signal are detected, and the average value of the two is used as the estimated value of the frequency offset component in the length of the current time, and the functional block diagram is shown in fig. 7.

In this embodiment, the snr estimation is located before the maximum ratio synthesis. The maximum ratio synthesis determines the weight coefficient of each branch according to the signal-to-noise ratio calculated by each branch (I and Q).

The signal-to-noise ratio estimation algorithm is a second-order fourth-moment method. Let the second moment of the observed signal be:

M₂＝E[x(n)x^*(n)]＝A²+σ²(3)

the fourth moment is:

M₄＝E[x(n)x^*(n)]²＝A²+2σ⁴+4A²σ²(4)

in which A is the amplitude of the s (n) signal, σ²For white gaussian noise variance, the SNR can be expressed as SNR ═ a²/σ²。

Simultaneous resolution of (3) and (4)

In practice, the second and fourth moments are calculated from the time average of the received signal, with the estimated value being

The signal-to-noise ratio estimate is

In this embodiment, before the maximum ratio synthesis is performed, the maximum ratio synthesis is performed after the signal-to-noise ratio modulation weight coefficient measured according to the left-right-handed 2-path signal. The voltage of the 2 paths of signals received in diversity is set as S_AAnd S_BNoise voltage of N_AAnd N_BThe measured signal-to-noise ratio is SNR_AAnd SNR_B. When the 2-way signal-to-noise ratio values are not very different (less than 10dB), an appropriate weighting function C_AAnd C_BCan be defined by the following formula:

C_A+C_B＝1 (11)

the weighting coefficient can be calculated from equations (10) and (11).

The coherent noise of the signals is added in root mean square, so that the resultant output signal S₀Sum noise N₀Respectively as follows:

S₀＝C_AS_A+C_BS_B(12)

N₀＝[(C_AN_A)²+(C_BN_B)²]^1/2(13)

let A be C_A/C_BThen the output signal-to-noise ratio is:

when N is present_A＝N_BWhen N is equal to N, there are

It can be obtained from the formula (15),

when S is_A＝S_BWhen the temperature of the water is higher than the set temperature,(16)

S_A＞＞S_Bwhen S is present₀/N₀≈S_A/N (17)

S_A＜＜S_BWhen S is present₀/N₀≈S_B/N (18)

Therefore, when the signal-to-noise ratios of 2 paths of input signals are equal during maximum ratio synthesis, the maximum polarization synthesis gain can reach 3 dB; when the signal-to-noise ratio of one path is much larger than that of the other path, the output signal-to-noise ratio is close to that of the high signal-to-noise ratio branch, and the high signal-to-noise ratio branch can be directly selected for output.

Tests show that the polarization diversity receiving method for wireless telemetry can realize the following steps:

1) diversity gain >2 dB;

2) adapted input intermediate frequency signal range: -60-0 dBm;

3) AFC range: 1 MHz;

4) configurable AGC time constant: closing for 1ms, 10ms, 100ms and 1000 ms;

5) configurable AFC time constant: off, 100ms, 10ms, 3 ms.