阅读说明：本技术 系统发现与信令 (System discovery and signaling ) 是由 M·J·西蒙 K·A·谢尔比 M·厄恩肖 S·M·坎那帕 于 2016-03-09 设计创作，主要内容包括：本公开的实施例涉及系统发现与信令。本文描述了可扩展通信系统。该系统包括用于接收根索引值并基于根值来生成恒幅零自相关序列的第一模块。该系统还包括用于接收种子值并用于基于种子值来生成伪噪声序列的第二模块。该系统还包括用于通过伪噪声序列来调制恒幅零自相关序列并用于生成复序列的第三模块。系统还包括用于将复序列转换为时域序列的第四模块,其中第四模块向时域序列应用循环移位,以获得经移位的时域序列。(Embodiments of the present disclosure relate to system discovery and signaling. A scalable communication system is described herein. The system includes a first module for receiving a root index value and generating a constant amplitude zero auto-correlation sequence based on the root value. The system also includes a second module for receiving a seed value and for generating a pseudo-noise sequence based on the seed value. The system also includes a third module for modulating the constant amplitude zero auto-correlation sequence with a pseudo-noise sequence and for generating a complex sequence. The system also includes a fourth module for converting the complex sequence to a time-domain sequence, wherein the fourth module applies a cyclic shift to the time-domain sequence to obtain a shifted time-domain sequence.)

1. A method for generating a symbol of a plurality of symbols, the method comprising:

generating a complex-valued time-domain sampled main sequence from the frequency-domain sequence;

generating a first auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the first auxiliary sequence comprises a first plurality of samples of the main sequence;

generating a second auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the second auxiliary sequence comprises a second plurality of samples of the main sequence, and wherein the first plurality of samples overlaps the second plurality of samples;

forming the symbol as the first auxiliary sequence, followed by the second auxiliary sequence, followed by the main sequence; and

transmitting the symbol to one or more receiver devices, wherein the symbol facilitates initial synchronization at the one or more receiver devices.

2. The method of claim 1, wherein generating the first auxiliary sequence comprises:

selecting 504 samples before the last 16 samples of the main sequence as the first plurality of samples; and

multiplying the selected samples by a complex number to generate the first auxiliary sequence.

3. The method of claim 1, wherein generating the second auxiliary sequence comprises:

the last 520 samples of the main sequence are selected as the second plurality of samples.

4. The method of claim 1, wherein generating the master sequence comprises:

a cyclic shift is applied to a time domain sequence derived from the frequency domain sequence.

5. A transmitter for generating a symbol of a plurality of symbols, the transmitter comprising:

a memory storing instructions; and

a processor, when executing the instructions, configured to:

generating a complex-valued time-domain sampled main sequence from the frequency-domain sequence;

generating a first auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the first auxiliary sequence comprises a first plurality of samples of the main sequence;

generating a second auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the second auxiliary sequence comprises a second plurality of samples of the main sequence, and wherein the first plurality of samples overlaps the second plurality of samples;

forming the symbol as a first auxiliary sequence followed by the second auxiliary sequence followed by the main sequence; and

transmitting the symbol to one or more receiver devices, wherein the symbol facilitates initial synchronization at the one or more receiver devices.

6. A communication system, comprising:

a transmitter for generating a symbol of a plurality of symbols, the transmitter comprising:

a memory storing instructions; and

a processor, when executing the instructions, configured to:

generating a complex-valued time-domain sampled main sequence from the frequency-domain sequence;

generating a first auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the first auxiliary sequence comprises a first plurality of samples of the main sequence;

generating a second auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the second auxiliary sequence comprises a second plurality of samples of the main sequence, and wherein the first plurality of samples overlaps the second plurality of samples;

forming the symbol as a first auxiliary sequence followed by the second auxiliary sequence followed by the main sequence; and

transmitting the symbol to a receiver device, wherein the symbol facilitates initial synchronization at the receiver device.

7. A method of communication, comprising:

generating a first auxiliary sequence of complex-valued time-domain samples based on the first main sequence;

generating a second auxiliary sequence of complex-valued time-domain samples based on the first main sequence;

forming a first symbol as the second auxiliary sequence, followed by the first main sequence, followed by the first auxiliary sequence;

generating a third auxiliary sequence of complex-valued time-domain samples based on the second main sequence, wherein the third auxiliary sequence comprises a first plurality of samples of the second main sequence;

generating a fourth auxiliary sequence of complex-valued time-domain samples based on the second main sequence, wherein the fourth auxiliary sequence comprises a second plurality of samples of the second main sequence, and wherein the first plurality of samples overlaps the second plurality of samples;

forming a second symbol as a third auxiliary sequence, followed by the fourth auxiliary sequence, followed by the second main sequence;

forming a plurality of symbols based on the first symbol and the second symbol; and

transmitting the plurality of symbols to a receiver device configured to distinguish the first symbol from the second symbol for initial synchronization at the receiver device.

8. A communication system, comprising:

a memory configured to store program instructions; and

a processor, upon execution of the program instructions, configured to:

generating a Pseudo Noise (PN) sequence based on the seed value;

generating a Constant Amplitude Zero Autocorrelation (CAZAC) sequence based on the root index value;

mapping a product of the PN sequence and the CAZAC sequence to a plurality of subcarriers such that each of the plurality of subcarriers has a subcarrier value, wherein:

a subcarrier value of a DC subcarrier of the plurality of subcarriers is zero, an

The subcarrier values of the plurality of subcarriers have reflection symmetry with respect to the DC subcarrier; and

converting the subcarrier values for each of the plurality of subcarriers to a time domain sequence, wherein one or more receiver devices may perform initial synchronization using the time domain sequence.

9. A method, comprising:

generating a Pseudo Noise (PN) sequence based on the seed value;

generating a Constant Amplitude Zero Autocorrelation (CAZAC) sequence based on the root index value;

mapping a product of the PN sequence and the CAZAC sequence to a plurality of subcarriers such that each of the plurality of subcarriers has a subcarrier value, wherein:

a subcarrier value of a DC subcarrier of the plurality of subcarriers is zero, an

The subcarrier values of the plurality of subcarriers have reflection symmetry with respect to the DC subcarrier; and

converting the subcarrier values for each of the plurality of subcarriers to a time domain sequence, wherein one or more receiver devices may perform initial synchronization using the time domain sequence.

10. A transmitting device, comprising:

a memory storing program instructions; and

a processor, upon execution of the program instructions, configured to:

generating a Pseudo Noise (PN) sequence based on the seed value;

generating a Constant Amplitude Zero Autocorrelation (CAZAC) sequence based on the root index value;

mapping a product of the PN sequence and the CAZAC sequence to a plurality of subcarriers such that each of the plurality of subcarriers has a subcarrier value, wherein:

a subcarrier value of a DC subcarrier of the plurality of subcarriers is zero, an

The subcarrier values of the plurality of subcarriers have a reflection symmetry about the DC;

converting the subcarrier values to a time domain sequence; and

transmitting the time domain sequence to one or more receiver devices, wherein the one or more receiver devices can perform initial synchronization using the time domain sequence.

Technical Field

The present disclosure relates to the field of wireless communications, and more particularly, to mechanisms for enabling robust signal detection and service discovery in a broadcast network.

Background

The broadcast spectrum is divided into different frequencies and allocated among different broadcasters for various uses in different geographical areas. The frequencies of the spectrum are allocated according to the permissions granted to the broadcaster. Based on the distribution, a broadcaster may be limited to broadcasting a particular type of content, such as a television signal, on a certain frequency within a certain geographic radius. Broadcasts outside of the allocated spectrum may be a broadcaster violation.

If the broadcaster wishes to transmit another type of content within the geographic radius, the broadcaster may need to obtain additional spectrum licenses and, in turn, allocate additional frequencies within that frequency. Similarly, if a broadcaster wishes to send content within another geographic radius, the broadcaster may need to obtain additional spectrum licenses for that region. However, obtaining additional spectrum licenses can be difficult, time consuming, expensive, and impractical.

Furthermore, broadcasters may not always make full use of the entire portion of spectrum that has been licensed. This can cause inefficiencies in broadcast spectrum utilization.

Furthermore, the expected usage of the broadcast spectrum may be changing. For example, current broadcast television solutions are monolithic and designed for a predominantly single service. However, in addition to broadcasting television in the future, broadcasters may also expect to provide a variety of wireless-based types of content, including mobile broadcast and internet of things services. In particular, there are many scenarios where a large number of devices may all wish to receive identical data from a common source other than a broadcast television set. One such example is mobile communications services, where a large number of mobile communications devices in various geographic locations may all need to receive a common broadcast signal conveying the same content, such as a software update or emergency alert, for example. In such a scenario, it is significantly more efficient to broadcast or multicast data to such devices rather than individually signal the same data to each device. Thus, a hybrid solution may be required.

To more efficiently utilize the broadcast spectrum, different types of content may be time division multiplexed together within a single RF channel. Further, different sets of transmission content may need to be sent simultaneously in a time division multiplexed manner (TDM), Frequency Division Multiplexed (FDM), Layer Division Multiplexed (LDM), or a combination thereof, with different encoding and transmission parameters. The amount of content to be transmitted may vary over time and/or frequency.

Furthermore, content with different quality levels (e.g., high definition video, standard definition video, etc.) may need to be sent to different groups of devices with different propagation channel characteristics and different reception environments. In other scenarios, it may be desirable to transmit device-specific data to a particular device, and the parameters used to encode and transmit that data may depend on the location of the device and/or propagation channel conditions.

At the same time, the demand for high-speed wireless data is increasing, and it is desirable to use available wireless resources (such as some portion of the wireless spectrum) as efficiently as possible on a potentially time-varying basis.

Disclosure of Invention

An exemplary scalable communication system is described herein. The system includes a first module for receiving a root index value and generating a constant amplitude zero auto-correlation sequence based on the root value. The system also includes a second module for receiving a seed value and for generating a pseudo-noise sequence based on the seed value. The system also includes a third module for modulating the constant amplitude zero auto-correlation sequence with a pseudo-noise sequence and for generating a complex sequence. The system also includes a fourth module for converting the complex sequence to a time-domain sequence, wherein the fourth module applies a cyclic shift to the time-domain sequence to obtain a shifted time-domain sequence. In one example, the root index value comprises a non-prime number.

Example scalable communication methods are described herein. The method includes the steps of receiving a root index value and generating a constant amplitude zero auto-correlation sequence based on the root value. The method also includes the steps of receiving a seed value and generating a pseudo-noise sequence based on the seed value. The method further comprises the step of modulating the constant amplitude zero auto-correlation sequence by a pseudo-noise sequence and generating a complex sequence. The method further comprises the step of converting the complex sequence into a time domain sequence and applying a cyclic shift to the time domain sequence to obtain a shifted time domain sequence. In one example, the root index value comprises a non-prime number.

In some embodiments, a method for generating a symbol of a plurality of symbols, the method comprising: generating a complex-valued time-domain sampled main sequence from the frequency-domain sequence; generating a first auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the first auxiliary sequence comprises a first plurality of samples of the main sequence; generating a second auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the second auxiliary sequence comprises a second plurality of samples of the main sequence, and wherein the first plurality of samples overlaps the second plurality of samples; forming the symbol as the first auxiliary sequence, followed by the second auxiliary sequence, followed by the main sequence; and transmitting the symbol to one or more receiver devices, wherein the symbol facilitates initial synchronization at the one or more receiver devices.

In some embodiments, wherein generating the first auxiliary sequence comprises: selecting 504 samples before the last 16 samples of the main sequence as the first plurality of samples; and multiplying the selected samples by a complex number to generate the first auxiliary sequence.

In some embodiments, wherein generating the second auxiliary sequence comprises: the last 520 samples of the main sequence are selected as the second plurality of samples.

In some embodiments, wherein generating the master sequence comprises: a cyclic shift is applied to a time domain sequence derived from the frequency domain sequence.

In some embodiments, wherein the cyclic shift comprises a relative cyclic shift between the symbol and a previous symbol and an absolute cyclic shift of the previous symbol.

In some embodiments, wherein applying the cyclic shift comprises: adding the relative cyclic shift and the absolute cyclic shift before applying the cyclic shift.

In some embodiments, wherein the plurality of symbols have a fixed sampling rate and a fixed bandwidth.

In some embodiments, wherein the fixed sample rate is 6.144 Msamples/sec, and the fixed bandwidth is 4.5 MHz.

In some embodiments, a transmitter for generating a symbol of a plurality of symbols, the transmitter comprising: a memory storing instructions; and a processor, when executing the instructions, configured to: generating a complex-valued time-domain sampled main sequence from the frequency-domain sequence; generating a first auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the first auxiliary sequence comprises a first plurality of samples of the main sequence; generating a second auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the second auxiliary sequence comprises a second plurality of samples of the main sequence, and wherein the first plurality of samples overlaps the second plurality of samples; forming the symbol as a first auxiliary sequence followed by the second auxiliary sequence followed by the main sequence; and transmitting the symbol to one or more receiver devices, wherein the symbol facilitates initial synchronization at the one or more receiver devices.

In some embodiments, wherein to generate the first auxiliary sequence, the processor is further configured to: selecting 504 samples before the last 16 samples of the main sequence as the first plurality of samples; and multiplying the selected samples by a complex number to generate the first auxiliary sequence.

In some embodiments, wherein to generate the second auxiliary sequence, the processor is further configured to select a last 520 samples of the main sequence as the second plurality of samples.

In some embodiments, wherein to generate the master sequence, the processor is further configured to apply a cyclic shift to a time domain sequence derived from the frequency domain sequence.

In some embodiments, wherein the cyclic shift comprises a relative cyclic shift between the symbol and a previous symbol and an absolute cyclic shift of the previous symbol.

In some embodiments, wherein the processor is further configured to add the relative cyclic shift and the absolute cyclic shift before applying the cyclic shift.

In some embodiments, wherein the plurality of symbols have a fixed sampling rate and a fixed bandwidth.

In some embodiments, wherein the fixed sample rate is 6.144 Msamples/sec, and the fixed bandwidth is 4.5 MHz.

In some embodiments, a communication system comprises: a transmitter for generating a symbol of a plurality of symbols, the transmitter comprising: a memory storing instructions; and a processor, when executing the instructions, configured to: generating a complex-valued time-domain sampled main sequence from the frequency-domain sequence; generating a first auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the first auxiliary sequence comprises a first plurality of samples of the main sequence; generating a second auxiliary sequence of complex-valued time-domain samples based on the main sequence, wherein the second auxiliary sequence comprises a second plurality of samples of the main sequence, and wherein the first plurality of samples overlaps the second plurality of samples; forming the symbol as a first auxiliary sequence followed by the second auxiliary sequence followed by the main sequence; and transmitting the symbol to a receiver device, wherein the symbol facilitates initial synchronization at the receiver device.

In some embodiments, wherein to generate the first auxiliary sequence, the processor is further configured to: selecting 504 samples before the last 16 samples of the main sequence as the first plurality of samples; and multiplying the selected samples by a complex number to generate the first auxiliary sequence.

In some embodiments, wherein to generate the second auxiliary sequence, the processor is further configured to select a last 520 samples of the main sequence as the second plurality of samples.

In some embodiments, wherein to generate the master sequence, the processor is further configured to apply a cyclic shift to a time domain sequence derived from the frequency domain sequence.

In some embodiments, wherein the cyclic shift comprises a relative cyclic shift between the symbol and a previous symbol and an absolute cyclic shift of the previous symbol.

In some embodiments, wherein the processor is further configured to add the relative cyclic shift and the absolute cyclic shift before applying the cyclic shift.

In some embodiments, wherein the plurality of symbols have a fixed sampling rate and a fixed bandwidth.

In some embodiments, wherein the fixed sample rate is 6.144 Msamples/sec, and the fixed bandwidth is 4.5 MHz.

In some embodiments, the system further comprises: the receiver is configured to include a plurality of the transmitted symbols.

In some embodiments, a method of communication, comprising: generating a first auxiliary sequence of complex-valued time-domain samples based on the first main sequence; generating a second auxiliary sequence of complex-valued time-domain samples based on the first main sequence; forming a first symbol as the second auxiliary sequence, followed by the first main sequence, followed by the first auxiliary sequence; generating a third auxiliary sequence of complex-valued time-domain samples based on the second main sequence, wherein the third auxiliary sequence comprises a first plurality of samples of the second main sequence; generating a fourth auxiliary sequence of complex-valued time-domain samples based on the second main sequence, wherein the fourth auxiliary sequence comprises a second plurality of samples of the second main sequence, and wherein the first plurality of samples overlaps the second plurality of samples; forming a second symbol as a third auxiliary sequence, followed by the fourth auxiliary sequence, followed by the second main sequence; forming a plurality of symbols based on the first symbol and the second symbol; and transmitting the plurality of symbols to a receiver device, the receiver device configured to distinguish the first symbol from the second symbol for initial synchronization at the receiver device.

In some embodiments, wherein generating the third auxiliary sequence comprises: selecting 504 samples before the last 16 samples of the second main sequence as the first plurality of samples; and multiplying the selected samples by a complex number to generate the first auxiliary sequence.

In some embodiments, wherein generating the first auxiliary sequence comprises: selecting the last 504 samples of the first primary sequence; and multiplying the selected samples by a complex number.

In some embodiments, a communication system comprises: a memory configured to store program instructions; and a processor, upon execution of the program instructions, configured to: generating a Pseudo Noise (PN) sequence based on the seed value; generating a Constant Amplitude Zero Autocorrelation (CAZAC) sequence based on the root index value; mapping a product of the PN sequence and the CAZAC sequence to a plurality of subcarriers such that each of the plurality of subcarriers has a subcarrier value, wherein: a subcarrier value of a DC subcarrier of the plurality of subcarriers is zero, and the subcarrier value of the plurality of subcarriers has reflection symmetry with respect to the DC subcarrier; and converting the subcarrier values for each of the plurality of subcarriers to a time domain sequence, wherein one or more receiver devices may perform initial synchronization using the time domain sequence.

In some embodiments, wherein the time domain sequence is one of a plurality of symbols, and the processor, when executing the program instructions, is further configured to: mapping a product of the CAZAC sequence and a second PN sequence to the plurality of subcarriers across each of the plurality of symbols, wherein the second PN sequence is a continuously advancing PN sequence across the plurality of symbols.

In some embodiments, wherein the time domain sequence is one of a plurality of symbols, and the processor when executing the program instructions is further configured to invert the subcarrier value for each of the plurality of subcarriers of a final symbol of the plurality of symbols to indicate termination of the plurality of symbols.

In some embodiments, wherein the processor, when executing the program instructions, is further configured to apply a cyclic shift to the time domain sequence to obtain a shifted time domain sequence.

In some embodiments, wherein the time domain sequence is one symbol of a plurality of symbols, and the processor when executing the program instructions is further configured to generate the cyclic shift of the one symbol based on an absolute cyclic shift of a previous symbol and a relative cyclic shift of the one symbol, the relative cyclic shift being relative to the absolute cyclic shift of the previous symbol.

In some embodiments, wherein the plurality of symbols have a fixed sampling rate and a fixed bandwidth.

In some embodiments, wherein the fixed sample rate is 6.144 Msamples/sec, and the fixed bandwidth is 4.5 MHz.

In some embodiments, wherein to convert the subcarrier values to the time-domain sequence, the processor, when executing the program instructions, is configured to convert the subcarrier values to the time-domain sequence using a 2048-point Inverse Fast Fourier Transform (IFFT).

In some embodiments, a method, comprising: generating a Pseudo Noise (PN) sequence based on the seed value; generating a Constant Amplitude Zero Autocorrelation (CAZAC) sequence based on the root index value; mapping a product of the PN sequence and the CAZAC sequence to a plurality of subcarriers such that each of the plurality of subcarriers has a subcarrier value, wherein: a subcarrier value of a DC subcarrier of the plurality of subcarriers is zero, and the subcarrier value of the plurality of subcarriers has reflection symmetry with respect to the DC subcarrier; and converting the subcarrier values for each of the plurality of subcarriers to a time domain sequence, wherein one or more receiver devices may perform initial synchronization using the time domain sequence.

In some embodiments, wherein the time domain sequence is one of a plurality of symbols, and further comprising: mapping a product of the CAZAC sequence and a second PN sequence to the plurality of subcarriers across each of the plurality of symbols, wherein the second PN sequence is a continuously advancing PN sequence across the plurality of symbols.

In some embodiments, wherein the time domain sequence is one of a plurality of symbols, and the method further comprises: inverting the subcarrier value for each of the plurality of subcarriers of a final symbol of the plurality of symbols to indicate termination of the plurality of symbols.

In some embodiments, the method further comprises: applying a cyclic shift to the time domain sequence to obtain a shifted time domain sequence, wherein the cyclic shift represents communication information; and transmitting the shifted time domain sequence to the one or more receiver devices.

In some embodiments, wherein the time domain sequence is one of a plurality of symbols, and the method further comprises: generating the cyclic shift of the one symbol based on an absolute cyclic shift of a previous symbol and a relative cyclic shift of the one symbol, the relative cyclic shift being relative to the absolute cyclic shift of the previous symbol.

In some embodiments, wherein the plurality of symbols have a fixed sample rate of 6.144 Msamples/sec and a fixed bandwidth of 4.5 MHz.

In some embodiments, wherein converting the subcarrier values to a time domain sequence comprises: the subcarrier values are converted to the time domain sequence using 2048 Inverse Fast Fourier Transform (IFFT).

In some embodiments, a transmitting device comprises: a memory storing program instructions; and a processor, upon execution of the program instructions, configured to: generating a Pseudo Noise (PN) sequence based on the seed value; generating a Constant Amplitude Zero Autocorrelation (CAZAC) sequence based on the root index value; mapping a product of the PN sequence and the CAZAC sequence to a plurality of subcarriers such that each of the plurality of subcarriers has a subcarrier value, wherein: a subcarrier value of a DC subcarrier of the plurality of subcarriers is zero, and the subcarrier value of the plurality of subcarriers has a reflection symmetry with respect to the DC; converting the subcarrier values to a time domain sequence; and transmitting the time domain sequence to one or more receiver devices, wherein the one or more receiver devices can perform initial synchronization using the time domain sequence.

In some embodiments, wherein the time domain sequence is one of a plurality of symbols, and the processor when executing the program instructions is further configured to invert the subcarrier value for each of the plurality of subcarriers of a final symbol of the plurality of symbols to indicate termination of the plurality of symbols.

In some embodiments, wherein the time domain sequence is one of a plurality of symbols, and the processor, when executing the program instructions, is further configured to: generating a cyclic shift of the one symbol based on an absolute cyclic shift of a previous symbol and a relative cyclic shift of the one symbol, the relative cyclic shift being relative to the absolute cyclic shift of the previous symbol; and applying the cyclic shift to the time domain sequence to obtain a shifted time domain sequence

In some embodiments, wherein to apply the cyclic shift, the processor, when executing the program instructions, is further configured to: adding the relative cyclic shift and the absolute cyclic shift before applying the cyclic shift.

In some embodiments, wherein the shifted time domain sequence comprises control signaling configured to allow reception and decoding of a waveform.

Drawings

In the accompanying drawings, structures of exemplary embodiments of the claimed invention are illustrated in conjunction with the detailed description provided below. Similar elements are identified with the same reference numerals. It will be understood that elements shown as a single component may be substituted for multiple components and elements shown as multiple components may be substituted for a single component. The figures are not to scale and the proportions of certain elements may be exaggerated for illustrative purposes.

Fig. 1 illustrates an example broadcast network.

FIG. 2 illustrates an example system for initiating a boot (bootstrap) symbol.

FIG. 3 illustrates a complex I/Q constellation of ZC + PN.

4A-4B illustrate example framework control combinations, respectively.

Fig. 5 illustrates example field termination signaling.

Fig. 6 illustrates example signal waveforms illustrated in fig. 1.

FIG. 7 illustrates an example system for initiating a bootstrap symbol.

Fig. 8 illustrates an example PN sequence generator.

Fig. 9 is an exemplary diagram of a mapping to a frequency domain sequence of subcarriers.

10A-10B illustrate example time domain structures.

Fig. 11 illustrates an example for initiating a bootstrap symbol.

Detailed Description

Described herein is a robust and scalable signaling framework, and in particular, a pilot signal is designed to enable robust detection and service discovery, system synchronization, and receiver configuration. The boot provides two main functions: synchronization and signaling to discover a waveform transmitted via low level signaling to begin decoding a subsequent waveform. It is a robust waveform that provides scalability that can evolve over time. In particular, the pilot signal works for current broadcast systems, but also allows support for new services including mobile broadcast and internet of things services.

Robust signaling systems enable signals to be found in high noise, low "carrier to noise ratio" (CNR), and high doppler environments. It will be appreciated that this is possible, only the pilot signal may be robust, while the actual waveform after the pilot may not be so robust. Having a robust pilot signal allows the receiver to synchronize to achieve and maintain lock on signals picked up in less than ideal environments. When the noise condition deteriorates and the receiver can no longer identify the payload from the noise, it can still remain locked to the channel by steering. When the noise conditions improve, the receiver does not need to go through the entire reacquisition procedure, since it already knows where to find the channel.

With a scalable signaling system, many different waveforms can be signaled, one for each of the service types to be transmitted in the future. Thus, new waveforms that may need to be used, which do not exist today, may also be signaled by the bootstrap.

It is to be understood that the following abbreviations and abbreviations may be used herein:

BSR Baseband sampling Rate

BW bandwidth

CAZAC constant amplitude zero autocorrelation

DC direct current

EAS emergency alert system

FFT fast Fourier transform

Institute of IEEE (institute of Electrical and electronics Engineers)

IFFT inverse fast Fourier transform

kHz kilohertz

LDM layered multiplexing

LFSR linear feedback shift register

MHz

ms

PN pseudo noise

Us microsecond

ZC Zadoff-Chu

Fig. 1 illustrates an example broadcast network communication system 100, the example broadcast network communication system 100 including a plurality of content providers 102A, 102B, and 102C (hereinafter content providers 102) that provide various types of content 104A, 104B, and 104C (hereinafter content 104) via a broadcast network 106. It should be understood that although three content providers 102 are illustrated, the system 100 may include any suitable number of content providers 102. Further, the content provider 102 may be a provider of any suitable type of content, such as television broadcast signals, software updates, emergency alerts, and the like. It should also be understood that the content provider 102 may provide the content 104 to the gateway 108 via a wireless connection or a wired connection.

Content 104 is time-multiplexed into a single RF channel 110 at gateway 108. Broadcast receivers 112A, 112B, and 112C (hereinafter broadcast receiver 112) are configured to identify and receive broadcast signals 114 via RF channel 110. It should be understood that although three different types of broadcast receivers 112 (laptop 112A, mobile phone 112B, and television 112C) are illustrated, system 100 may include any suitable number and type of broadcast receivers 112.

The guide (not shown) indicates at a low level the type or form of signal 114 being transmitted during a particular time period so that broadcast receiver 112 can discover and identify signal 114, which in turn indicates how to receive the services available via that signal 114. Therefore, bootstrapping relies on being an integral part of each transmitted frame to allow synchronization/detection and system configuration. As will be described, the bootstrap design includes a flexible signaling method for transmitting frame configuration and content control information to the broadcast receiver 112. Signal design describes the mechanism by which signal parameters are modulated on a physical medium. The signaling protocol describes the particular encoding used to convey the parameter selection that governs the configuration of the transmit frame. This enables reliable service discovery while providing scalability for adapting to evolving signaling requirements from a common frame structure. In particular, the pilot design enables generic signal discovery independent of the channel bandwidth.

Steering also enables reliable detection in the presence of various channel impairments such as time dispersion and multipath fading, doppler frequency shift, and carrier frequency offset. Furthermore, detecting that multiple service contexts are accessible based on patterns during signal discovery enables a wide range of flexibility in system configuration. Bootstrapping also facilitates extensibility to accommodate the continued evolution in service capabilities based on a layered signaling structure. Thus, new signal types that have not yet been contemplated may be provided by the content provider 102 and identified within the transmitted signal 114 through the use of the bootstrap signal. Furthermore, regardless of the level of scalability provided, a reusable bit field interpreted based on the detected service mode/type enables bit efficient signaling. In one example, the pilot is configured as a robust signal and is detectable even at low signal levels. Thus, the separate signaling bits in the pilot can be quite expensive in terms of the physical resources they occupy for transmission. Thus, steering may be intended to signal only the minimum amount of information needed for system discovery and initial decoding of subsequent signals.

General guide overview

Described herein is booting, independent of implementation examples to be described later. As will be further described, ATSC 3.0 is one example implementation of a bootstrap capability, and places certain constraints on generic bootstrap capabilities. An understanding of these general concepts in a guided configuration will help those skilled in the art to see the wide application of this technology in future communication systems of various bandwidths and bands in the RF spectrum.

FIG. 2 illustrates an example system 200 for generating guidance. The pilot signal 202 generated by the system 200 is comprised of (N) OFDM symbols labeled (0-N). By design, the frequency occupancy or bandwidth is less than the trailing pilot signal 204 or waveform. The post pilot signal 204 represents the service transmitted by the pilot signal and consumed by the receiver. The post-pilot signal 204 may be any waveform that enables future flexibility and scalability as will be discussed.

Described herein are pilot signals. The Baseband Sampling Rate (BSR) is represented by:

equation (2)

(N +16) x (m) or time domain, T_s＝1/F_s

Wherein F_sIs the frequency sampling, N is the operating variable scaled over the selected bandwidth, and M is the factor (MHz) used to select the bandwidth.

The OFDM subcarrier spacing (Hz) is defined as:

equation (3)

ΔF＝F_s/FFT_(Size)

With FTTs sized to some power of 2 (e.g., 1024, 2048, 4096, 8192.).

In one example of the U.S. 6MHz broadcast television bandwidth (ATSC 3.0) design process, the equation M is chosen to be 0.384 because of the existing relationship to LTE (based on WCDMA). Other relationships may be selected. Thus, in this example:

equation (4)

F_s＝(0+16)X(.384MHz)＝6.144MHz；

F_s＝6.144MHz,FFT_(Size)2048; and

ΔF＝6.144MHz/2048＝3000Hz

then the length N of the Zadoff Chu sequence is selected_(ZC)(based on prime numbers) to perform FFT_(Size)To support the selected bandwidth. Therefore, the temperature of the molten metal is controlled,

equation (5)

Bandwidth＝ΔF X(N_(ZC)+1)；

Wherein N is_ZCIs mapped to the center of the FFT (1500 subcarriers including DC) and zero padding is used on the remaining subcarriers. In the ATSC 3.0 example, N is selected_(ZC)1499. Therefore, the temperature of the molten metal is controlled,

Bandwidth＝3000(Hz)x(1499+1)＝4.5MHz

thus, in the described example ATSC 3.0 implementation, the design consumes 4.5MHz bandwidth and has Δ F of 3000Hz, which will give adequate doppler performance (MPH) for broadcast bands in a mobile environment.

It will be appreciated that other choices of parameters in the above general equation may enable a wider bandwidth or frequency band (doppler), etc. Specifically, although the value (N) is designated 0 in ATSC 3.0, the entire range of (0-127) is available for N. In the illustrated example, N is constrained to be 0 to achieve 6 MHz. It should be understood, however, that bandwidths greater than 50MHz may be supported by substituting N127. This is illustrative of the scalability of the boot.

Referring again to FIG. 2, the system also includes a Zadoff-Chu module or sequence generator 206 and a Pseudo Noise (PN) module or sequence generator 208. Zadoff-chu (zc) sequences are complex-valued mathematical sequences that, when applied to radio signals, result in a pair of interesting attributes, one of which is a constant amplitude signal. It can be defined as:

equation (6)

Fig. 3 shows a complex I/Q constellation 300 of ZC + PN, where each I/Q value resides on a unit circle 302 and is depicted as a phase around the unit circle 302, where the amplitude is constant.

It will be appreciated that another theoretical property of (ZC) is that different cyclically shifted versions of the root sequence may be imposed on the signal and may result in ideal zero auto-correlation. The already generated Zadoff-Chu sequence that has not been shifted is considered to be the "root sequence". Referring again to fig. 2, the symbol #0, which is mainly used for synchronization and versioning, is not shifted. However, it should be understood that theoretical zero autocorrelation by using (ZC) alone cannot be achieved by a large range of cyclic shifts.

As a result of this basic design requirement, the need for a large number of cyclic shifts with theoretically ideal autocorrelation is foreseen, which is unnatural for individual (ZCs). Then, through simulations and experiments, it was discovered and developed that by introducing pseudo-noise (PN) sequences in addition to ZC, all cyclic shifts can be able to achieve near theoretically ideal autocorrelation.

FIG. 4B shows simulation results of only a single ZC and the resulting non-ideal autocorrelation, while FIG. 4A is simulation results of a ZC + PN and shows the resulting approximate ideal autocorrelation. In particular, the PN sequence phase rotation maintains a single composite subcarrier of the desired constant amplitude zero autocorrelation waveform ("CAZAC") properties of the original ZC sequence, as shown in fig. 3. The added phase rotation is intended to provide greater signal separation between cyclic shifts of the same root sequence that suppresses the spurious autocorrelation response using ZC sequences according to the sequence without adding PN sequence modulation, as shown in figure 4B. Thus, as can be appreciated, the discovery of (ZC + PN) greatly improves the signaling robustness and capacity (number per symbol) conveyed by the cyclic shift mechanism.

Referring again to fig. 2, the first symbol #0 is the root without cyclic shift, while symbols 1-N carry signaling via a cyclic shift mechanism. Furthermore, it can be seen that mapping and zero padding are applied to symbol #0 by mapping module 210. The symbols (1-N) have a PN added to ZC, which results in reflection symmetry as shown, and which will be discussed by way of example later.

The signal is then sent to the IFFT module 212 and converted from the frequency domain to the time domain. The signal is then processed in the time domain. The signal leaving the IFFT is called "a" which then has prefix and suffix portions called "B" and "C" derived from "a". Symbol #0 has the time sequence "CAB" while all other symbols have the time sequence "BCA". It should be appreciated that the purpose is to increase robustness and distinguish symbol #0 for synchronization and versioning.

The length of the pilot symbol is defined by:

equation (7)

T_Symbol＝[C+A+B]x T_s

In one example (ATSC 3.0), the symbol length is 500 μ β.

To enable the ability to extend the number of symbols, a mechanism to direct the inversion of the (ZC) on the last symbol in the sequence is used, as illustrated in fig. 5. Specifically, the signaling field terminates by a 180 ° phase reversal relative to the previous symbol period within the last symbol period. Thus, instead of needing to specify in advance how long the signal will be in order for the receiver to be able to identify the end of the signal, the receiver can instead look for an inverted symbol in the signal that will indicate the end of the signal. This allows the boot to be flexible and scalable, since it is not necessary to know in advance how long the signal will be. Thus, instead of pre-defining the length of the pilot and wasting extra space or not reserving enough space (in which case it may not be possible to fully transmit the intended information), the length of the pilot is flexible as it can be discovered. Furthermore, the inverted signal can be detected relatively easily, and thus does not require a lot of additional resources to implement.

It should be appreciated that the receiver may gracefully ignore its unappreciated master version (root). This ensures scalability without destroying legacy receivers in the future. Indeed, one such signaling method is provided by ATSC 3.0 for later discussion and is illustrated by table 2 herein.

Fig. 6 illustrates an example signal waveform 114 illustrated in fig. 1. The signal waveform 114 includes a pilot 602 followed by a post-pilot waveform 604 or the rest of the waveform 602. Boot 602 provides a generic entry point into signal waveform 114. Which employs a fixed configuration (e.g., sampling rate, signal bandwidth, subcarrier spacing, time domain structure) known to all broadcast receivers 112.

It should be appreciated that the flexible or variable sampling defined in the pilot provides flexibility not previously available. In particular, rather than designing a solution for a particular service with a fixed or defined sampling rate as a function of bandwidth, a flexible sampling rate enables scaling of a variety of different bandwidths in order to accommodate different services with different requirements and constraints. Thus, the same system used for synchronization and discovery can be used for a wide range of bandwidths and can serve a large frequency band, as different parts of the frequency band can be better suited for different types of services.

The guide 602 may be composed of a plurality of symbols. For example, the pilot 202 may start with a synchronization symbol 606 located at the beginning of each waveform to enable service discovery, coarse synchronization, frequency offset estimation, and initial channel estimation. The remainder 608 of the pilot 602 may contain sufficient control signaling to allow reception and decoding of the remainder of the signal waveform 114 to begin.

The boot 602 is configured to exhibit flexibility, scalability, and extensibility. For example, boot 602 may implement versioning for increased flexibility. In particular, the boot 602 design may enable a major version number (corresponding to a particular service type or mode) and a minor version (within a particular major version). In one example, versioning may be signaled (as will be described) via appropriate selection of Zadoff-Chu root (primary version) and pseudo-noise sequence seed (secondary version) for generating the base coding sequence for pilot symbol content. Decoding of the signaling field within the bootstrap 602 may be performed with respect to the detected service version, enabling hierarchical signaling, where each allocated bit field is reusable and configured based on the indicated service version. The syntax and semantics of the signaling fields within the guide 612 may be specified, for example, within the standards referenced by the major and minor versions.

To further demonstrate scalability and scalability, the number of bits signaled per pilot 602 symbol may be defined (up to a maximum) for a particular primary/secondary version. The maximum number of bits per symbol is defined by:

equation (1)

Depending on the desired cyclic shift tolerance, which in turn depends on the desired channel deployment scenario and environment. Additional new signaling bits can be added to existing symbols in a backward compatible manner if available, without the need to change the service version.

Thus, the pilot 602 signal duration is scalable throughout the symbol period, with each new symbol carrying up to N_bpsAn additional signaling bit. The pilot 602 signal capacity may be dynamically increased until the field termination is reached.

Fig. 7 illustrates an example system 700 for initiating a boot 602 symbol. As described above, the value for each pilot 602 symbol originates in the frequency domain with a Zadoff-chu (zc) sequence 704 modulated with a pseudo-noise (PN) cover sequence 702 with a sequence generator 708. The ZC root 704 and the PN seed 702 determine the primary and secondary versions of a service, respectively. The resulting complex sequence is applied subcarrier-by-subcarrier at an inverse fast fourier transform ("IFFT") input 706. The system 700 also includes a subcarrier mapping module 710 for mapping the output of the sequence generator 708 to the IFFT input 706. The PN sequence 702 introduces a phase rotation into individual composite subcarriers that preserve the desired Constant Amplitude Zero Autocorrelation (CAZAC) properties of the original ZC sequence 704. The PN sequence 702 further suppresses spurious emissions in the autocorrelation response, thereby providing additional signal separation between cyclic shifts of the same root sequence.

It should be further appreciated that modulating the ZC sequence with, in particular, a pseudo-noise sequence gives different characteristics to the waveforms that make it easy to find. In particular, modulation with a PN sequence results in a near-ideal correlation with less uncertainty. After testing many combinations of algorithms and sequences, such combinations were discovered by simulation. In particular, modulating a ZC sequence with a PN sequence produces the unexpected result of producing a signal correlation that is susceptible to correlation with no spurious signals created during association. This results in a signal that is easily discoverable, meaning that the receiver may correlate with the signal even in a high noise setting.

Guide-implementation (ATSC 3.0 example)

Described herein are example implementations of example boot 602. It should be understood that although the examples described herein may refer to particular implementations of guidance, it is contemplated that guidance 602 will have broader application beyond the examples illustrated below.

Guide norm-size

In one example, the boot 602 structure is intended to remain fixed, even if the version number and/or other information signaled by the boot 602 evolves. In one example, the pilot 602 uses a fixed sampling rate of 6.144 Msamples/sec and a fixed bandwidth of 4.5MHz, regardless of the channel bandwidth used for the rest of the waveform 604. The time length of each sample is also fixed by the sample rate. Therefore, the temperature of the molten metal is controlled,

equation (2A)

f_s＝6.144Ms/sec

T_s＝1/f_s

BW_Bootstrap＝4.5MHz

The FFT size of 2048 results in a subcarrier spacing of 3 kHz.

Equation (3)

N_FFT＝2048

f_Δ＝f_s/N_FFT＝3kHz

In this example, each boot 602 symbol has a duration of 333.33 μ s. When processing is done in the time domain (discussed later using (CAB or BCA)), T_symbolThe exact length of (3) is 500. mu.s. The entire duration of boot 602 depends on being designated as N_sThe number of pilot 602 symbols. A fixed number of pilot 602 symbols should not be assumed.

Equation (4)

T_symbol＝500μs

It should be appreciated that a bandwidth of 4.5MHz may be selected based on current industry consensus, which in this example also covers broadcasts of 5MHz and less than 6MHz as the lowest bandwidth commonly used. Thus, the baseband sampling rate may be calculated using the following equation:

equation (5)

(N +16) X0.384 MHz 6.144MS/sec (N0 guide)

The 2048FFT length is chosen to have good gain, resulting in a Δ f of 3kHz giving good doppler performance. It should be understood that similar systems may be implemented for other portions of the frequency band. For example, variations of the same formula in which the formula for that particular bandwidth and the value of N are to be optimized may be used for other bandwidths, such as 20 MHz.

It should be appreciated that by basing the BSR formula on a 0.384MHz factor in relation to LTE systems (and in relation to WCDMA), new systems may be able to exclude oscillators for other implementations. In addition, all 3GPP LTE baseband sampling rates for all current bandwidths described in today's standards can also be calculated from the formula by inserting the value (N). Thus, employing this formula allows future versions of devices that contain some sort of changes to LTE changes to still work. However, it should be understood that the BSR formula may similarly be based on other suitable factors.

It should also be understood that while the examples described herein utilize a selected FFT size of 2048, other suitable FFT sizes may similarly be used. The receiver must first synchronize and identify the incoming signal so that it can begin decoding its information. However, longer signaling sequences (such as 2048FFT size) have higher gain and are therefore easier to find because the amount of information the receiver can use to find or correlate is larger.

In existing cellular communications, gain may not be a factor, since communications occur in a unicast nature, and Primary Synchronization Signals (PSS) are frequently inserted for random access by multiple users. Furthermore, the broadcaster may not be concerned with the gains in the past, as the broadcast may generally be intended for static receivers located high. However, higher gains may become more important when broadcasting to mobile devices or in locations that receive poor reception. However, the mobile device may not have the optimal antenna shape to rely on for gain and may not be ideally located for optimal reception, and therefore may rely on mathematical gain.

Thus, a longer signal length (such as the example FFT 2048) provides a longer sequence to correlate and thus results in more robust reception. For example, with longer signals, the guide may be found at a subterranean location below the noise floor. In addition, longer signal lengths also enable more dedicated sequences. For example, each transmitter may be assigned a dedicated sequence and then the receiver may independently search for the sequence. This information may be used by, for example, a Global Positioning System (GPS) system to calculate the position of the receiver using TDOA techniques, which are not discussed herein.

It should be appreciated that signal lengths of 2048 have been identified herein in order to optimize performance, although other suitable signal lengths may be selected. In particular, selecting different signal lengths may result in a trade-off between different parameters including the amount of gain that may affect performance.

Pilot canonical-frequency domain sequence

In one example, the Zadoff-Chu (ZC) sequence has a length N_ZC1499, where this is the maximum prime number that results in a channel bandwidth of no more than 4.5MHz with subcarrier spacing of f Δ — 3 kHz. ZC sequences are parameterized by the root q, corresponding to the major version number:

equation (5)

Wherein

q∈{1，2，...，N_zc-1}

And

k＝0,1,2,...，N_zc-1.

modulating ZC sequences using pseudo-noise sequences has allowed relaxation of the constraints on the ZC root. While previous signaling methods with ZC (e.g., LTE primary synchronization sequence) were limited to selecting the primary root to ensure good autocorrelation properties, in this system, PN modulation allows good autocorrelation even when non-primary roots are selected for ZC. Having a non-primary root for ZC allows for the creation of more waveforms, allowing the system to signal more types of services, i.e. to create a more scalable system.

Fig. 8 illustrates an example PN sequence generator 708. The PN sequence generator 808 is derived from a Linear Feedback Shift Register (LFSR)802 of length (order) l-16. The operation of LFSR802 is governed by a generator polynomial 804, the generator polynomial 804 specifying taps in the LFSR feedback path, the generator polynomial 804 followed by a mask 806, the mask 806 specifying elements that contribute to a sequence output 808. The designation of generator polynomial 804 and the initial state of the registers represents a seed corresponding to a minor version number. That is, the seed is defined as f (g, r)_tnit)。

The PN sequence generator register 802 is reinitialized from the seed with an initial state prior to the generation of the first symbol in the new boot 602. The PN sequence generator 708 continues the sequence from one symbol to the next within the pilot 602 and does not re-initialize for consecutive symbols within the same pilot 602.

The output of the PN sequence generator 708 is defined as p (k), which will have a value of 0 or 1. After the PN sequence generator 708 has been initialized with the appropriate seed value and before any clocking of the shift register 802, p (0) will be equal to the output at the PN sequence generator. Each time the shift register 802 is clocked to the right position, a new output p (k) will then be generated. Thus, in one example, the generator polynomial 804 for the PN sequence generator 708 should be defined as:

equation (6)

g＝{g1,...,g0}＝{1,1,1,0,0,0,0,0,0,0,0,0,0,0,0,1,1}

Wherein

p(x)＝x¹⁶+x¹⁵+x¹⁴+x

Fig. 9 is an example illustration of a mapping 900 to a frequency domain sequence of subcarriers. ZC sequence values (i.e. z) mapped to DC sub-carriers_q((N_zc-1)/2)) are zeroed so that the DC subcarrier is null. The subcarrier index is illustrated with the center DC subcarrier having an index of 0.

The product of the ZC and PN sequences has reflection symmetry about the DC subcarrier. ZC sequences have natural reflection symmetry about the DC subcarrier. Reflection symmetry about the PN sequence of the DC subcarrier is introduced by mirror reflecting the PN sequence values assigned to subcarriers below the DC subcarrier to subcarriers above the DC subcarrier. For example, as shown, the PN sequence values at subcarriers-1 and +1 are identical, as are the PN sequence values at subcarriers-2 and + 2. Therefore, the product of the ZC sequence and the PN sequence also has reflection symmetry with respect to the DC subcarrier.

It should be appreciated that the symmetry described herein enables a more robust signal to be found more easily. In particular, symmetry serves as an additional aid to discovery (i.e., additional gain). This is an additional feature of the signal that the receiver can look for, making it easier to find. Thus, allowing guidance to be identified even below the noise floor is one of the elements.

As illustrated by the mapping 900, the nth symbol of the pilot (0 ≦ n)<N_s) The subcarrier values of (a) may be expressed as:

equation (7)

Wherein

N_H＝(N_ZC-1)/2

And is

c(k)＝1-2x p(k)

Wherein c (k) has the value +1 or-1. It should be understood that the ZC sequence is the same for each symbol, while the PN sequence progresses with each symbol.

In one example, the final symbol in the pilot is indicated by a phase reversal (i.e., a rotation of 180 °) of the subcarrier value for that particular symbol. The boot termination signaling enables scalability by allowing the number of symbols in the boot to be increased for additional signaling capacity in a backward compatible manner without requiring major or minor version numbers to be changed. Phase inversion simply involves multiplying each subcarrier value by e^jπ＝-1：

Equation (8)

This phase reversal enables the receiver to correctly determine the endpoint of the boot. For example, the receiver may determine an endpoint for a secondary version of the bootstrap that is later than the secondary version for which the receiver was designed and that has been extended by one or more bootstrap symbols. Thus, the receiver does not need to assume a fixed number of pilot symbols. In addition, the receiver may ignore the signaling bit content of pilot symbols that the receiver is not provided with for decoding, but still detect the presence of the pilot symbols themselves.

Once mapped, the frequency domain sequence is via N_FFTThe 2048 point IFFT is converted to the time domain:

equation (9)

Guiding specification-symbol signaling

By using A_n(t) cyclic shifts in the time domain of the time domain sequence signal information via pilot symbols. The sequence has N_FFT2048, and therefore 2048 different cyclic shifts (from 0 to 2047, including 0 and 2047) are possible. With 2048 possible cyclic shifts, up to log can be signaled₂(2048)＝11log₂(2048) 11 bits. It should be understood that not all of these bits will actually be used. In particular, the amount of the solvent to be used,

indicates for the nth pilot symbol (1 ≦ n)<N_s) Of the signaling bits, and

representing the values of these bits.

The number of valid signaling bits in the received pilot symbols may be greater than the signaling bits expected by the receiverThe number of the cells. To facilitate future signaling extensions while maintaining backward compatibility, a receiver should not assume that the number of valid signaling bits in a received pilot symbol is not greater than the signaling bits expected by the receiver

The number of the cells. For example, one or more specific bootstrap symbols may be added when a new minor version is defined in the same major version

To take advantage of previously unused signaling bits while still maintaining backward compatibility. Thus, a receiver provided with signaling bits for decoding a particular major/minor version may ignore any bits that may be used in later minor versions in the same major versionWhat new additional signaling bits.

It should be appreciated that in the examples described herein, the distance between correlation peaks between the leads of symbols in the time domain is the distance at which the signaling information is encoded. Specifically, the symbol #0 is a reference point (absolute displacement), and its distance from a subsequent peak (relative to the first peak) carries information. For example, the meaning of the distance may be determined from a defined look-up table. Thus, the receiver does not attempt to decode the bits, but rather attempts to identify the correlation peaks. Once the receiver finds a peak, it waits for the next peak and the time between these peaks holds signaling information. This creates a more robust system because the time difference between peaks is more easily found in high noise conditions, even though using 256 cyclic shifts, for example, to represent 8 bits of binary information may be relatively expensive. However, the actual signaling of the payload after the bootstrapping may still include a modulation scheme with actual bits carrying the information.

In one example, for the nth pilot symbol (1)<n<N_s) Is expressed as a cyclic shift of the previous pilot symbol with respect to the cyclic shift of the previous pilot symbol

Is calculated from the signaling bit value of the nth pilot symbol using a gray code method.

Represented in binary form as a set of bits

Each bit of (a) is calculated as follows:

equation (10)

The addition of the signalling bits followed by the modulo operation effectively performs a logical exclusive-or operation on the signalling bits in question.

This equation ensures relative cyclic shifts

Is calculated to provide maximum tolerance to any error at the receiver when estimating the relative cyclic shift for the received pilot symbols. If the number of valid signaling bits of a particular pilot symbol

Added in a future minor version in the same major version, the equation also ensures that the relative cyclic shift of the future minor version guide symbol will be calculated in a manner that still allows a receiver provided for the previous minor version to correctly decode the signaling bit values provided for decoding, and thus will maintain backward compatibility.

It should be understood that, in general, signaling bits

Will be greater than

Is given if i<k。

In one example, a first pilot symbol is used for initial time synchronization and signals a major version number and a minor version number via ZC root and PN seed parameters. The symbol does not signal any additional information and therefore always has a cyclic shift of 0.

Computing a differentially encoded absolute cyclic shift M applied to an nth pilot symbol by summing the absolute cyclic shift of the pilot symbol n-1 and the relative cyclic shift of the pilot symbol n and modulo the length of the time domain sequence_n(0≤M_n<N_FFT)：

Equation (11)

An absolute cyclic shift is then applied to obtain a shifted time domain sequence from the output of the IFFT operation:

equation (12)

Guiding a canonical-time domain structure

Each pilot symbol consists of three parts: A. b and C, wherein each of these parts consists of a sequence of complex-valued time-domain samples. Part A is derived as an IFFT with a frequency domain structure to which an appropriate cyclic shift is applied, while B and C consist of samples taken from A, where f_ΔFrequency shift sum (equal to subcarrier spacing)^-jπIs introduced into the frequency domain sequence of samples used to calculate the part B. Portions A, B and C each include N_A＝N_FFT＝2048、N_B504, and N_C520 samples. Each pilot symbol thus contains N for an equivalent time length of 500 mus_A+N_B+N_C3072 samples.

In one example, the time domain structure includes two variants: CAB and BCA. The initial symbol that provides the pilot for synchronous detection (i.e., pilot symbol 0) employs a C-a-B variation. The remaining pilot symbols (i.e., pilot symbol n, where 1 ≦ n<N_s) Conforming to the B-C-a change up to and including a bootstrap symbol indicating termination of the field.

It will be appreciated that repeating a portion of the pilot allows for improved initial synchronization and discovery, since the receiver knows that the repetition is expected in a particular order, and here makes it easier to find and lock on to the signal even under high noise conditions.

Fig. 10A illustrates an example CAB structure 1010. In this example, part C1012 is represented by the last N of part A1014_B504 samples, whichMiddle + f_ΔFrequency shift sum e of^-jπIs applied to the starting frequency domain sequence S for the calculation of the part a1014_n(k) In that respect The samples of portion B1016 may be taken as the last N of the computed cyclically shifted time-domain sequence_BNegation of samples where the input frequency domain sequence is equal to S shifted in frequency by one higher frequency subcarrier position_n(k) (i.e., S)_n(k)＝S_n((k-1+N_FFT)mod N_FFT) In which S is_n(k) An input frequency domain sequence of frequency and phase shifted samples for generating part B1016). Alternatively, by multiplying the appropriately extracted samples from part A1014 by e^j2πfΔtThe frequency and phase shift used to generate the part B1016 samples may be introduced in the time domain (as shown by the following equation):

equation (13)

Fig. 10B illustrates an example BCA structure 1020. In this example, portion C1012 is again represented by the last N of A1014_C520 samples, but B1016 consists of the first N of C1012_B504 samples, where-f_ΔIs applied to the starting frequency domain sequence S for the calculation of the part a1014_n(k) In that respect In a similar manner as described with respect to the example CAB structure 1010, the samples of the portion B1016 may be used as the last N of the calculated cyclic shifted time domain sequence_BOne sample, where the input frequency domain sequence is equal to S shifted in frequency by one lower subcarrier position_n(k) (i.e., S)_n(k)＝S_n((k-1)mod N_FFT) In which S is_n(k) Is the input frequency domain sequence of frequency shifted samples used to generate part B1016). The frequency shift used to generate the portion B1016 samples may alternatively be in the time domain by multiplying the appropriate samples from portion A1014 by e-^j2πfΔtIs introduced, wherein a constant time offset of-520T is included to account for the correct extraction of the appropriate sample portion a1014, as shown in the following equation:

equation (14)

It should be appreciated that the samples of the portion B1016 may be obtained from slightly different portions of the portion a1014 of each of the CAB structure 1010 and the BCA structure 1020.

Pilot signal structure

Example pilot signal structures are described herein. The signaling set or structure includes a list of configuration parameter values, control information fields, and the assignment of these values and fields to particular signaling bits. It should be understood that the pilot signal structure may take other suitable forms than the examples described herein.

The example pilot signal structure described herein may apply when the major version number is equal to 0. The corresponding root (q) of the ZC sequence is 137. The cardinality of the symbols in the pilot (including the initial synchronization symbols) should be N_s4. It should be understood that N_s4 denotes the minimum number of symbols that can be transmitted. Thus, to enable the transmission of additional signaling bits, N_s4 denotes the minimum number of symbols to be transmitted within the pilot signal (but not necessarily the maximum).

In one example, the generator polynomial for the pseudo-noise sequence generator is defined as:

equation (15)

g＝{g_l,...,g₀}＝{1,1,1,0,0,0,0,0,0,0,0,0,0,0,0,1,1}＝[16 15 14 10]

p(x)＝x¹⁶+x¹⁵+x¹⁴+x+1

And the initial register state of the pseudo-noise sequence generator is defined as:

equation (16)

r_init＝{η_-1,...,r₀}＝{0,0,...,0,1}

In one example, the initial register state of the PN sequence generator of the leading minor version of a selected one of the given major versions is set to a value from a list of predefined values in order to signal the corresponding minor version being used. Table 1 illustrates example initial register states for respective sub-versions of a PN sequence generator.

Table 1-initial register states of PN sequence generators

The pilot signal structure may include additional signaling fields following the major and minor version signals. For example, the signal structure may include a wake-up bit. This may be, for example, an emergency alert wakeup bit. This is a 1-bit field that is either on (1) or off (0).

The signal structure may also include a minimum time interval to the next frame of the same major and minor version fields. This is defined as the time period measured from the start of the leading of frame a to the earliest possible occurrence of the leading of frame B. Pilot B is guaranteed to be within a time window starting at the minimum time interval value of the signal transmission and ending with the next higher minimum time interval value that may have been signaled. The time window is not terminated if the signal conveys the highest possible minimum time interval value. An example signal mapping formula may be defined as:

equation (17)

Thus, an example signaled value of X ═ 10 would indicate that boot B is located somewhere in a time window starting at 700 milliseconds from the beginning of boot a and ending at 800 milliseconds from the beginning of boot a.

As the minimum time interval value of the signaling increases, the number is signaled via a sliding scale with increasing granularity. X denotes a signaled 5-bit value and T denotes a minimum time interval (in milliseconds) to the next frame matching the same version number as the current frame. Table 2 shows example values.

Table 2-example of minimum time interval to next frame

The signal structure may also include a system bandwidth field. This field signals the system bandwidth for the late leader portion of the current frame. Values include 00-6 MHz, 01-7 MHz, 10-8 MHz, and 11-greater than 8 MHz. It should be appreciated that the "greater than 8 MHz" option facilitates future operations using system bandwidths greater than 8 MHz. A receiver not provided for processing system bandwidths greater than 8MHz may ignore frames with this field equal to 11.

In one example, table 3 illustrates that the pilot signaling field is mapped to specific signaling bits and pilot symbols. The most significant bit to the least significant bit of each signaling field is mapped to the designated signaling bits in a given order from left to right. It should be understood that,the ith signaling bit referring to the nth pilot symbol, and pilot symbol 0 does not carry any particular signaling bit.

Table 3-example guide signaling bitmap

Fig. 11 illustrates an example scalable communication method. In step 1102, a first module receives a root index value and generates a constant amplitude zero auto-correlation sequence based on the root value. In step 1104, the second module receives a seed value and generates a pseudo-noise sequence based on the seed value. In step 1106, the third module modulates the constant amplitude zero auto-correlation sequence with a pseudo-noise sequence and generates a complex sequence. In step 1108, the fourth module converts the complex sequence to a time domain sequence and applies a cyclic shift to the domain sequence to obtain a shifted time domain sequence.

Any of the various embodiments described herein may be implemented in any of various forms, such as a computer-implemented method, as a computer-readable storage medium, as a computer system, and so on. The system may be implemented by one or more custom hardware devices, such as an Application Specific Integrated Circuit (ASIC), by one or more programmable hardware elements, such as Field Programmable Gate Arrays (FPGAs), by one or more processors executing stored program instructions, or by any combination of the preceding.

In some embodiments, a non-transitory computer-readable storage medium may be configured such that it stores program instructions and/or data, wherein the program instructions, if executed by a computer system, cause the computer system to perform a method, e.g., any one of the method embodiments described herein, or any combination of the method embodiments described herein, or any subset of any one of the method embodiments described herein, or any combination of these subsets.

In some embodiments, a computer system may be configured to include a processor (or a collection of processors) and a storage medium, wherein the storage medium stores program instructions, wherein the processor is configured to read and execute the program instructions from the storage medium, wherein the program instructions are executable to implement any of the various method embodiments described herein (or any combination of the method embodiments described herein, or any subset of any of the method embodiments described herein, or any combination of these subsets). The computer system may be implemented in any of various forms. For example, the computer system may be a personal computer (in any of its various implementations), a workstation, a computer on a card, a special purpose computer in a case, a server computer, a client computer, a handheld device, a mobile device, a wearable computer, a sensing device, a television, a video capture device, a computer embedded within a living being, and so forth. The computer system may include one or more display devices. Any of the various computational results disclosed herein may be displayed via a display device, or otherwise displayed as output via a user interface device.

To the extent that the term "includes" or "including" is used in either the detailed description or the claims, it is intended to be inclusive in a manner similar to the term "comprising" as that term is interpreted when employed as a transitional word in a claim. Further, to the extent that the term "or" (e.g., a or B) is employed, it is intended to mean "a or B or both". When applicants intend to indicate "only a or B but not both," then the term "only a or B but not both" will be employed. Thus, use of the term "or" herein is the inclusive, and not the exclusive use. See Bryan A. Garner, A Dictionary of Modem LegalUsage 624(2d. Ed. 1995). Furthermore, to the extent that the term "in" or "into" is used in either the specification or the claims, it is intended to be used to additionally mean "at …" or "to …". Furthermore, to the extent that the term "connected" is used in either the specification or the claims, it is intended to mean not only "directly connected," but also "indirectly connected," such as by way of another component or components.

While the present application has been illustrated by the description of embodiments thereof, and while the embodiments have been described in considerable detail, it is not the intention of the applicants to restrict or in any way limit the scope of the appended claims to such detail. Additional advantages and modifications will readily appear to those skilled in the art. The application, in its broader aspects, is therefore not limited to the specific details, representative apparatus and method, and illustrative examples shown and described. Accordingly, departures may be made from such details without departing from the spirit or scope of the applicant's general inventive concept.