阅读说明：本技术 马达控制系统和助力转向系统 (Motor control system and power steering system ) 是由 远藤修司 原田昆寿 上田智哉 绵引正伦 森智也 馆胁得次 于 2019-02-08 设计创作，主要内容包括：一个方式的马达控制系统具有：逆变器,其使马达驱动；以及控制运算部,其运算表示从上述逆变器向上述马达提供的电流的电流指令值,上述控制运算部具有：电压控制运算部,其根据上述电流指令值与上述实际电流检测值之间的电流偏差来运算表示从上述逆变器对上述马达施加的电压的电压指令值；以及补偿运算部,其针对经由上述电压控制运算部的信号流的上游侧和下游侧中的至少一方的信号值,补偿上述马达的k次成分和1/k次成分中的至少一方,上述补偿运算部根据上述目标电流指令值和表示上述马达旋转的角速度的实际角速度值并且也考虑提前角控制来运算补偿值。(One embodiment of a motor control system includes: an inverter that drives the motor; and a control calculation unit that calculates a current command value indicating a current supplied from the inverter to the motor, the control calculation unit including: a voltage control calculation unit that calculates a voltage command value indicating a voltage applied from the inverter to the motor based on a current deviation between the current command value and the actual current detection value; and a compensation calculation unit that compensates at least one of a k-th order component and a 1/k-th order component of the motor for a signal value of at least one of an upstream side and a downstream side of a signal flow passing through the voltage control calculation unit, wherein the compensation calculation unit calculates a compensation value based on the target current command value and an actual angular velocity value indicating an angular velocity of rotation of the motor, and also in consideration of advance angle control.)

1. A motor control system for driving a motor having a number of phases n of 3 or more,

the motor control system includes:

an inverter that drives the motor; and

a control calculation unit that calculates a current command value indicating a current supplied from the inverter to the motor, based on a target current command value supplied as a control target of the motor from outside,

the control calculation unit includes:

a voltage control calculation unit that calculates a voltage command value indicating a voltage applied from the inverter to the motor, based on a current deviation between the current command value and the actual current detection value; and

a compensation calculation unit that compensates at least one of a k-th order component and a 1/k-th order component of the motor rotation for a signal value of at least one of an upstream side and a downstream side of a signal flow passing through the voltage control calculation unit,

the compensation calculation portion calculates a compensation value from the target current command value and an actual angular velocity value representing an angular velocity of rotation of the motor, and also in consideration of advance control.

2. The motor control system of claim 1,

the compensation calculation unit compensates for noise generated by coupling of the motor and at least a part of a steering mechanism.

3. The motor control system according to claim 1 or 2, wherein,

the compensation calculation unit compensates for noise in a resonance band of a coupling that couples the motor and the steering mechanism.

4. The motor control system according to claim 2 or 3,

the compensation calculation unit sets the advance angle control to a control condition suitable for reducing the noise, as compared to a control condition suitable for reducing torque ripple of the motor.

5. The motor control system according to any one of claims 1 to 4,

the compensation calculation unit calculates a compensation value gamma satisfying the following expression,

γ＝Asin(Bθ+C)

A＝Lookuptable_A(Iq_target，ω)

C＝Lookuptable_C(Iq_target，ω)

wherein Iq _ target is the target current command value.

6. The motor control system according to any one of claims 1 to 5,

the control calculation unit feeds back an actual current detection value indicating a current supplied from the inverter to the motor to the current command value to control the inverter.

7. The motor control system according to any one of claims 1 to 6,

the compensation calculation unit calculates the compensation value as a current value and adds the current value to a signal value on the upstream side of the voltage control calculation unit.

8. The motor control system according to any one of claims 1 to 6,

the compensation calculation unit calculates the compensation value as a voltage value and adds the voltage value to a signal value on the downstream side of the voltage control calculation unit.

9. A power steering system, having:

the motor control system of any one of claims 1 to 8;

a motor controlled by the motor control system; and

and a power steering mechanism driven by the motor.

10. A motor control method for controlling the driving of a motor having a phase number n of 3 or more,

the motor control method calculates a current command value indicating a current supplied from an inverter for driving the motor to the motor, based on a target current command value supplied as a control target of the motor from outside, and the calculation of the current command value includes:

a voltage control calculation step of calculating a voltage command value indicating a voltage applied from the inverter to the motor, based on a current deviation between the current command value and the actual current detection value; and

a compensation calculation step of compensating at least one of a k-th order component and a 1/k-th order component of the motor rotation for a signal value of at least one of an upstream side and a downstream side of a signal flow passing through the voltage control calculation step,

in the compensation calculating step, a compensation value is calculated from the target current command value and an actual angular velocity value indicating an angular velocity of the motor rotation, taking into account advance control.

11. The motor control method according to claim 10,

the compensation calculation step compensates for noise generated by coupling between the motor and at least a part of a steering mechanism.

12. The motor control method according to claim 10 or 11, wherein,

the compensation arithmetic process compensates for noise in a resonance band of a coupling the motor and the steering mechanism.

13. The motor control method according to claim 11 or 12, wherein,

in the compensation calculation step, the advance control is set to a control condition suitable for reducing the noise, as compared with a control condition suitable for reducing the torque ripple of the motor.

14. The motor control method according to any one of claims 10 to 13,

the compensation calculation unit calculates a compensation value gamma satisfying the following expression,

γ＝Asin(Bθ+C)

A＝Lookuptable_A(Iq_target，ω)

C＝Lookuptable_C(Iq_target，ω)，

wherein Iq _ target is the target current command value.

Technical Field

The present disclosure relates to a motor control system and a power steering system.

Background

Conventionally, as a motor control technique, a method is known in which a control device feedback-controls a motor using a command value. For example, a control device is known that feeds back a current command value that is in an opposite phase to a torque ripple and adds the current command value to a basic command value. In such a configuration, a method is known in which the control device superimposes a current command value of a harmonic component of a current value on a basic command value to compensate for torque ripple (for example, patent document 1).

Disclosure of Invention

Problems to be solved by the invention

However, even when the operating sound is reduced by the motor alone, when the motor is coupled to the steering mechanism, noise may be generated due to resonance or the like. Therefore, an object of the present invention is to realize low operating noise both when the motor is a single body and when the motor is coupled to a steering mechanism.

Means for solving the problems

A motor control system according to an aspect of the present invention is a motor control system for driving a motor having a number of phases n of 3 or more, the motor control system including: an inverter that drives the motor; and a control calculation unit that calculates a current command value indicating a current supplied from the inverter to the motor, based on a target current command value supplied as a control target of the motor from outside, the control calculation unit including: a voltage control calculation unit that calculates a voltage command value indicating a voltage applied from the inverter to the motor, based on a current deviation between the current command value and the actual current detection value; and a compensation calculation unit that compensates at least one of k-th order components and 1/k-th order components of the motor for a signal value of at least one of an upstream side and a downstream side of a signal flow passing through the voltage control calculation unit, wherein the compensation calculation unit calculates a compensation value from the target current command value and an actual angular velocity value indicating an angular velocity at which the motor rotates, and also in consideration of advance angle control.

Further, a power steering system according to an aspect of the present invention includes: the motor control system; a motor controlled by the motor control system; and a power steering mechanism driven by the motor.

Effects of the invention

According to the exemplary embodiments of the present invention, a low operating sound can be achieved both in the case where the motor is a single body and in the case where the motor is coupled to the steering mechanism.

Drawings

Fig. 1 is a schematic diagram of a motor control system of a first embodiment.

Fig. 2 is a schematic diagram of a control arithmetic unit according to the first embodiment.

Fig. 3 is a gain characteristic diagram for the target q-axis current Iq _ target.

Fig. 4 is a phase plot for a target q-axis current Iq _ target.

Fig. 5 is a schematic diagram showing a flow of the operation processing based on the 2D map.

Fig. 6 is a diagram showing a simulation result of torque fluctuation in the first embodiment.

Fig. 7 is a schematic diagram of a motor control system of the second embodiment.

Fig. 8 is a schematic diagram of a control calculation unit according to the second embodiment.

Fig. 9a is a plan view of the first motor of the present embodiment.

Fig. 9b is a plan view of the second motor of the present embodiment.

Fig. 10 is a schematic view of the electric power steering apparatus of the present embodiment.

Fig. 11 is a conceptual diagram of a motor unit having a traction motor.

Fig. 12 is a side schematic view of the motor unit.

Fig. 13 is a schematic diagram of a motor control system of the third embodiment.

Fig. 14 is a schematic diagram of a control arithmetic unit according to the third embodiment.

Fig. 15 is a diagram showing a lookup table stored in the storage section.

Fig. 16 is a diagram showing a process of adjusting the compensation value γ and recording.

Fig. 17 is a diagram schematically showing a state of noise measurement.

Fig. 18 is a graph showing an example of noise data obtained by noise measurement.

Fig. 19 is a diagram illustrating the monitored noise level.

Fig. 20 is a graph showing the effect of compensating for torque fluctuations.

Fig. 21 is a graph showing the effect of compensating for noise.

Fig. 22 is a schematic diagram of a motor control system of the fourth embodiment.

Fig. 23 is a schematic diagram of a control arithmetic unit according to the fourth embodiment.

Detailed Description

Hereinafter, embodiments of a controller, a motor control system having the controller, and an electric power steering system having the motor control system according to the present disclosure will be described in detail with reference to the accompanying drawings. However, in order to avoid unnecessarily long descriptions below, it is easy for those skilled in the art to understand that the detailed descriptions above may be omitted. For example, detailed descriptions of known matters and repetitive descriptions of substantially the same structure may be omitted.

< first embodiment >

A motor control system according to a first embodiment is described, in which a torque ripple compensation calculation unit that compensates for torque ripple is provided as a compensation calculation unit, and the output of the torque ripple compensation calculation unit is a "current value". The motor control system according to the first embodiment is a control system that controls a 3-phase brushless motor, for example. Hereinafter, for convenience, a case where the d-axis current Id and the q-axis current Iq are positive with each other, that is, a case where the rotation is in one direction will be described. In the motor control system of the present embodiment, torque ripple can be mainly reduced.

Generally, the q-axis current Iq has a larger influence than the d-axis current Id with respect to the generation of torque in the 3-phase motor. Therefore, to reduce the torque ripple, it is preferable to mainly control the q-axis current Iq to apply the present control system. In addition, even in the case of a control system that reduces Back Electromotive Force (BEMF), feedback control can be performed by the same configuration as the present invention. That is, in the control method of the present invention, only the q-axis current may be used as the command value, or both the q-axis current Iq and the d-axis current Id may be used as the command value. In this specification, the description of the control method related to the d-axis current Id is omitted.

Fig. 1 is a schematic diagram of a motor control system according to a first embodiment, and fig. 2 is a schematic diagram of a control arithmetic unit according to the first embodiment. As shown in fig. 1, the motor control system 5 includes a motor rotation angle sensor 51, an inverter 52, and a control calculation unit 53. The control arithmetic unit 53 functions as a so-called current controller. As shown in fig. 2, the control arithmetic unit 53 includes a torque ripple compensation arithmetic unit 531, a current limit arithmetic unit 532, a voltage control arithmetic unit 533, an induced voltage compensation arithmetic unit 534, a 2-axis/3-phase conversion unit 535, a dead time compensation arithmetic unit 536, and a PWM control arithmetic unit 537.

The motor control system 5 controls the motor 1 via the inverter 52. The motor 1 includes a rotor 3, a stator 2, and a motor rotation angle sensor 51. The motor rotation angle sensor 51 detects the rotation angle of the rotor 3 of the motor 1. The detected rotation angle of the rotor is expressed in arbitrary angular units, and is appropriately converted from a mechanical angle to a motor electrical angle θ or from the motor electrical angle θ to a mechanical angle. The relationship between the mechanical angle and the motor electrical angle θ is expressed by a relational expression of the motor electrical angle θ ═ mechanical angle × (number of magnetic poles ÷ 2). Further, instead of the rotation angle sensor, an angular velocity sensor, for example, may be provided as a sensor for detecting the rotation of the motor.

The motor control system 5 of the present embodiment performs control of feeding back a current value (actual q-axis current value IQR) flowing through the inverter 52. Although not shown, the motor control system 5 may also perform known arithmetic processing such as field weakening control. The motor control system 5 can suppress torque variation of the motor 1 by performing field weakening control.

The target q-axis current Iq _ target is input to the motor control system 5 from the outside. The increase or decrease of the motor output is externally instructed by the increase or decrease of the target q-axis current Iq _ target. The present motor control system 5 current-limits the input target q-axis current Iq _ target. The current limit is processed by the current limit calculation unit 532. The current limit calculation unit 532 receives an input of the target q-axis current Iq _ target and performs adaptive control, thereby limiting the target q-axis current Iq _ target (output value) to a predetermined current value or less.

When the target q-axis current Iq _ target is not limited to exceed the predetermined current value, the motor applied voltage may be saturated as a result of the processing described later. When the motor applied voltage is saturated in this way, there is no room for adding the compensation current for suppressing the motor torque variation to the target q-axis current Iq _ target. As a result, the following problems occur: the torque fluctuation is increased sharply, and the working sound is generated. In order to avoid this problem, it is effective to limit the target q-axis current Iq _ target by the current limit calculation unit 532 so as to make a margin for the compensation current. The saturation of the motor applied voltage occurs depending on both the motor current and the motor rotation angular velocity. Therefore, the current limit calculation unit 532 of the present embodiment limits the motor current (target q-axis current Iq _ target) using a function having the motor rotation angular velocity as a parameter. By such current limitation, a margin for compensating for torque ripple at ordinary times (when the voltage is not saturated) is secured. Therefore, quiet and smooth rotation of the motor is achieved.

More specifically, the adaptive control of the current limit calculation unit 532 reduces the range using a function having a motor rotation angular velocity as a parameter. This function is a continuous function for the input target q-axis current Iq _ target. That is, the current limit calculation unit 532 does not perform discontinuous limitation such as peak off of the current, but performs continuous range reduction in which the current is limited more greatly as the input current value is larger. The function used for the range reduction in the current limit calculation unit 532 may be a function that exhibits linear reduction or a function that exhibits nonlinear (and continuous) reduction.

The reduction range of the range reduction is such that the reduction range of current value i is reduced so as to satisfy the following inequality.

Vsat＞(Ls+R)i+keω···(1)

Here, Vsat is a saturation voltage, Ls is an inductance of the motor, R is a resistance of the motor, and ke ω is an induced voltage accompanying the rotation of the motor.

In the adaptive control of the current limit calculation unit 532, the limit value of the current based on the range reduction is a limit value corresponding to the battery voltage Vbat when the driving is performed by the battery power supply. The battery power is used in the case where the supply amount of the alternator is insufficient. Since the battery power supply has an internal resistance, the internal resistance changes with the deterioration of the battery power supply, and the effective output voltage changes. Therefore, adaptive control is performed in accordance with the battery voltage Vbat.

The motor control system 5 performs torque ripple compensation control using the target q-axis current Iq _ target and the angular velocity ω of the rotor. The torque ripple compensation control is processed by the torque ripple compensation arithmetic unit 531. The torque ripple compensation calculation unit 531 performs calculation processing using the target q-axis current Iq _ target before correction and the angular velocity ω as parameters. The angular velocity ω is calculated from the rotation angle of the rotor 3 detected by the motor rotation angle sensor 51. More specifically, the torque ripple compensation operation unit 531 includes a phase compensation unit 5311. The phase compensation unit 5311 performs the torque ripple compensation control. In the present embodiment, the torque ripple compensation calculation unit 531 and the phase compensation unit 5311 have the same function.

Typically, torque fluctuations are affected by fluctuations in current. Therefore, by performing correction such as superimposing a current command value (compensation current) for suppressing torque ripple on the target q-axis current Iq _ target indicating the current to be supplied to the motor 1 in advance, it is possible to suppress torque ripple generated in the motor 1 (i.e., to perform torque ripple compensation).

The torque ripple compensation operation unit 531 of the present embodiment has a so-called lookup table. The look-up table is referred to with an angular velocity ω and a target q-axis current Iq _ target as inputs, and outputs a gain α and a phase β as reference values corresponding to the inputs. The torque ripple compensation calculation unit 531 calculates α sin6(θ + β) expressed by the following expression (2) and expression (3) using the gain α and the phase β, which are the output reference values, as parameters. This calculation result is superimposed on the target q-axis current Iq _ target output from the current limit calculation unit 532 as shown in fig. 2 and equation (2) described later, to calculate a corrected target q-axis current Iq _ correct as a new current command value.

Next, the correlation between the angular velocity ω, the target q-axis current Iq _ target, the gain α, and the phase β will be described. Fig. 3 is a gain characteristic diagram for the target q-axis current Iq _ target. Fig. 4 is a phase plot for a target q-axis current. The gain characteristic diagram of fig. 3 and the phase diagram of fig. 4 show the primary hysteresis characteristics, respectively. However, the gain α and the phase β may be obtained from characteristics in consideration of the quadratic response and the hysteresis thereafter.

In the phase diagram in fig. 4, the initial value is normalized to the target q-axis current Iq _ target. In fig. 3, the horizontal axis represents the angular velocity ω, and the vertical axis represents the value of the gain α (ω). In fig. 4, the horizontal axis represents the angular velocity ω and the vertical axis represents the phase β (ω). Here, the current (compensation value) for compensating for the torque ripple is a sine wave and is expressed by an approximation using a sixth harmonic component that is dominant among vibration components of the torque ripple. At this time, the target q-axis current value Iq _ corrected is expressed by the following equation (2) with the target q-axis current Iqt _ target before correction and the motor electrical angle θ (θ ═ ω t) as variables. In addition, t is a variable representing time.

Iq_correct＝Iq_target+αsin6(θ+β)···(2)

The gain α (ω) and the phase β (ω) are expressed by the following expression (3) using a look-up table. At this time, the gain α (ω) and the phase β (ω) are calculated by performing arithmetic processing using a 2D map (the absolute value of the target q-axis current Iq _ target before correction is U1, and the absolute value of the angular velocity ω is U2). Fig. 5 is a relational diagram of the arithmetic processing at this time. The angular velocity ω is calculated from the motor electrical angle θ acquired by the motor rotation angle sensor 51.

αsin6(θ+β)

α＝Lookuptable_α(Iq_target，ω)···(3)

β＝Lookuptable_β(Iq_target，ω)

As shown in fig. 5, the target q-axis current Iq _ target and the angular velocity ω are input to the torque ripple compensation operation unit 531, and the absolute value U1 of the target q-axis current Iq _ target and the absolute value U2 of the angular velocity ω are calculated. The look-up table (2D map) returns the values of the corresponding gain a and phase β according to the absolute values U1, U2.

Preferably, for each product including the motor control system 5 and the motor 1, the transmission characteristics are measured by, for example, individual measurement before product shipment, and a lookup table (2D map) shown in fig. 5 is individually created from the measured transmission characteristics. However, as for the lookup table (2D map), for example, the transmission characteristics may be measured as a representative value or an average value for the same type of motor 1 and the motor control system 5, and the lookup table created based on the transmission characteristics may be commonly used for the same type of motor 1. Alternatively, a common lookup table may be employed for the plurality of types of motors 1 and motor control systems 5 known to have transmission characteristics similar to each other.

The torque ripple compensation operation portion 531 calculates the compensation value α sin6(θ + β) in equation (2) and equation (3) using the value returned from the lookup table, and outputs the value. The compensation value α sin6(θ + β) output from the torque ripple compensation arithmetic unit 531 is superimposed on the target q-axis current Iq _ target before correction output from the current limit arithmetic unit 532.

The motor control system 5 of the present embodiment adds the compensation value of the q-axis current value output from the torque ripple compensation calculation unit 531 to the q-axis current value processed by the current limit calculation unit 532. The compensation value α sin6(θ + β) is a value for eliminating a component of the q-axis current due to torque ripple. That is, the compensation value α sin6(θ + β) is calculated from the angular velocity component and the reverse phase component of the sixth harmonic component (the order component of the torque ripple) of the command value.

In other words, in the present embodiment, the torque ripple compensation operation unit 531 calculates values for adjusting the gain and the phase (advance angle) based on a relational expression in which the inverse characteristic of the current controller is divided into the gain and the phase, and superimposes a compensation value based on these values on the command value. By compensating from the viewpoint of both the advance angle (i.e., the responsiveness of the current controller) and the torque ripple (i.e., the amplitude of the torque ripple), the torque ripple is suppressed. As described above, the torque ripple compensation operation unit 531 generates a compensation value for compensating the command value of the q-axis current for the responsiveness of the current controller (hereinafter referred to as "advance compensation") based on the advance control and a torque ripple compensation based on the addition of the reverse phase components of the torque ripple. Since the order component of the torque ripple is a high-frequency component such as the sixth harmonic component of the command value, the compensation of the responsiveness by the advance control functions efficiently. The value β in the advance control processed by the torque ripple compensation calculation unit 531 is a value that compensates for the responsiveness of the current controller, and therefore is usually a value different from 180 °.

Further, the motor control system 5 subtracts the actual q-axis current value IQR flowing through the inverter from the q-axis current value to which the compensation value is added as described above, and calculates a current deviation IQ _ err of the q-axis current. That is, the motor control system 5 of the present embodiment performs the following feedback control: the output of the motor and the like are controlled by performing PI control and the like using the current deviation IQ _ err of the q-axis current.

In the above description, the compensation value α sin6(θ + β) is added to the target q-axis current Iq _ target after the current limitation, but the compensation value α sin6(θ + β) may be added to the target q-axis current Iq _ target before the current limitation and then the current limitation is performed, or the compensation value α sin6(θ + β) may be added to the current deviation Iq _ err between the target q-axis current Iq _ target and the actual q-axis current value IQR.

As described above, the motor control system 5 of the first embodiment performs control for compensating for the responsiveness of the current controller in advance. That is, the motor control system 5 performs torque ripple compensation and advance angle compensation by feedback control. The advance angle compensation is performed based on a parameter calculated using the angular velocity component and the reverse phase component of the order component of the torque ripple in the target current value. In the present embodiment, the gain for adjusting (the amplitude of) the torque ripple is calculated based on the relational expression in which the inverse characteristic of the current controller is divided into the gain and the phase, the phase for adjusting the advance angle is calculated, and the compensation value using these values is derived.

By using the compensation value, torque ripple compensation for compensating for torque ripple generated in the motor 1 from the aspect of amplitude and advance angle compensation for compensating for torque ripple from the aspect of phase are performed. This can reduce the sensitivity of the motor control system 5 to quantization noise and sensor noise associated with the high-pass filter operation, and as a result, reduce torque ripple and also prevent deterioration of operating sound. In addition, the robustness of the motor control is improved.

In addition, by performing torque ripple compensation using an inverted phase component of the order component of torque ripple in the current value, the responsiveness of the current controller is improved in a low speed region of the motor. In addition, by calculating the parameters using not only the reverse phase component but also the angular velocity component of the current value, the responsiveness of the current controller is improved in a high-speed region of the motor. Therefore, the responsiveness of the current controller is improved even in a wide range from low speed to high speed.

As a method of compensating for torque ripple using the above-described reverse phase component of the current value, a method of adding a compensation value to a motor current command value and a method of adding a compensation value to a motor applied voltage command value are known, and in the present embodiment, a compensation value is added to a motor current command value. Thus, the torque fluctuation can be stably corrected regardless of the characteristic fluctuation of the motor.

After the current deviation IQ _ err of the q-axis current is obtained as described above, the motor control system 5 performs voltage control for calculating the motor applied voltage command value from the current deviation IQ _ err of the q-axis current. The voltage control is performed by the voltage control operation unit 533. In the present embodiment, PI control is used as the voltage control. The voltage control is not limited to PI control, and other control methods such as PID control may be employed. In the voltage control operation unit 533, the Q-axis PI control unit 5331 calculates a Q-axis voltage command value VQ1 from the current deviation IQ _ err of the Q-axis current, and adds the Q-axis voltage command value VQ1 to the non-interference element COR _ Q output from the non-interference processing unit 5332 to calculate a Q-axis voltage command value VQ 2. The non-interference element COR _ Q is a current element applied to avoid interference between a d-axis current (voltage) and a Q-axis current (voltage), for example.

Then, the motor control system 5 performs induction voltage compensation on the q-axis voltage command value VQ 2. The induced voltage compensation operation unit 534 performs induced voltage compensation. When the motor is driven, the motor is controlled in consideration of the influence of the induced voltage of the motor in addition to the current flowing through the motor. The induced voltage compensation calculation unit 534 performs advance angle control based on the reciprocal of the induced voltage (BEMF) generated by the motor, and compensates the induced voltage (BEMF).

That is, the induced voltage compensation calculation unit 534 obtains the reciprocal of the induced voltage (BEMF) generated by the motor, and calculates a compensation value for performing compensation of the advance angle of the adjustment voltage (or current) (advance angle compensation) based on the reciprocal. In the present embodiment, the induced voltage compensation arithmetic unit 534 calculates a q-axis voltage command value VQ3 by adding a compensation value for induced voltage compensation to the q-axis voltage command value VQ 2. In addition, if a compensation value based on the inverse of the induced voltage model is used, the compensation value may be subtracted from the q-axis voltage command value VQ2 instead of being added to the compensation value. The compensation value may be added to the voltage value of each phase converted from 2-axis/3-phase.

Further, the motor control system 5 performs 2-axis/3-phase conversion on the q-axis voltage command value VQ 3. The 2-axis/3-phase conversion is performed by the 2-axis/3-phase conversion calculation unit 535 according to the motor electrical angle θ. The 2-axis/3-phase conversion arithmetic unit 535 calculates the corresponding q-axis voltage and d-axis voltage from the q-axis voltage command value VQ3, and converts the q-axis voltage and d-axis voltage into 3-phase voltage command values for U, V, W phases.

Then, the motor control system performs dead time compensation based on the voltage command value for each phase output from the 2-axis/3-phase conversion arithmetic unit 535. The dead time compensation is performed by the dead time compensation operation unit 536. First, in the dead time compensation calculation unit 536, the midpoint modulation unit 5363 performs a calculation based on midpoint modulation in which a harmonic (for example, a third harmonic) that is n times a fundamental wave of a voltage is superimposed. n is a positive integer. By performing the midpoint modulation, the waveform of the voltage approaches a trapezoidal waveform from a sinusoidal waveform. This increases the effective voltage rate of the inverter 52.

Next, the dead time compensation operation unit 536 compensates for the dead time. The processing for the current deviation IQ _ err described above is performed up to the midpoint modulation section 5363, and the voltage component for reducing the current deviation IQ _ err is calculated. On the other hand, the target q-axis current IQ _ target is input to the target IQ 2-axis/3-phase conversion unit 5362, and the voltage command value corresponding to the target q-axis current IQ _ target is subjected to 2-axis/3-phase conversion. That is, the target IQ 2-axis/3-phase conversion section 5362 calculates a q-axis voltage and a d-axis voltage corresponding to the target q-axis current IQ _ target, and converts them into U, V, W-phase 3-phase voltage command values for the respective phases.

In the 2-axis/3-phase conversion by the target IQ 2-axis/3-phase conversion section 5362, the motor electrical angle is used for the calculation, as in the 2-axis/3-phase conversion by the 2-axis/3-phase conversion calculation section 535. However, in the motor control system 5 of the present embodiment, the motor electrical angle θ 2 obtained by phase-compensating the motor electrical angle θ detected by the sensor is used as the motor electrical angle input to the target IQ 2 axis/3 phase conversion unit 5362. The phase compensation is performed by the correction phase compensation section 5361, and the phase offset of the voltage caused by the motor rotation is compensated by the phase compensation.

Finally, the motor control system performs PWM control based on the voltage command value output from the dead time compensation arithmetic unit 536. The PWM control command value is calculated by a PWM control calculation unit 537. The PWM control calculation unit 537 controls the voltage of the inverter 52 based on the calculated command value. By this PWM control, a current corresponding to the current command value flows to the motor 1. In addition, as described above, the actual q-axis current value IQR flowing in the inverter 52 is fed back.

In the present system, the respective processes such as the voltage control, the induced voltage compensation, the 2-axis/3-phase conversion, the dead time compensation, and the PWM control are not limited to the above-described examples, and known techniques may be applied. In addition, in the present system, these compensation and control may not be performed as necessary. In the following description, the coupling of these elements (i.e., the above-described voltage control, induced voltage compensation, 2-axis/3-phase conversion, dead time compensation, PWM control, and other processes) is referred to as a controller element c(s). Note that only the coupling of the main block that performs feedback control such as PI control may be regarded as the controller element c(s). The coupling between the motor and the inverter is referred to as a device element p(s).

With respect to the first embodiment described above, the results obtained by the simulation are shown in fig. 6. Fig. 6 is a graph showing a variation of twenty-four times the component of torque (six times the component of electrical angle) with respect to the rotational speed of the motor. In this simulation, the range of rotation speed is 0[ min ]^-1]To 3000[ min ]^-1]The results of a total of 4 combined torque fluctuations, in which both the dead time on/off and the torque variation correction on/off are combined with each other, are obtained. As can be seen from fig. 6, when both the dead time compensation and the torque fluctuation correction are "on", the fluctuation of the motor torque (i.e., the torque ripple) becomes small. Therefore, it is understood that the torque ripple is reduced and the low operating sound is achieved by the first embodiment.

< second embodiment >

Next, a second embodiment of the present invention will be described, in which the output of the torque ripple compensation arithmetic unit is a "voltage value". The motor control system of the second embodiment is a control system of a 3-phase brushless motor. Note that, although the same contents as those of the first embodiment may be omitted from the description below, the same method or a different method may be employed. In the present embodiment, the torque ripple compensation calculation unit 531 and the phase compensation unit 5311 have the same functions as in the first embodiment.

Fig. 7 is a schematic diagram of a motor control system according to a second embodiment, and fig. 8 is a schematic diagram of a control arithmetic unit according to the second embodiment. As shown in fig. 7, the motor control system 5 includes a motor rotation angle sensor 51, an inverter 52, and a control calculation unit 53. As shown in fig. 8, the control arithmetic unit 53 includes a torque ripple compensation arithmetic unit 531, a current limit arithmetic unit 532, a voltage control arithmetic unit 533, an induced voltage compensation arithmetic unit 534, a 2-axis/3-phase conversion unit 535, a dead time compensation arithmetic unit 536, and a PWM control arithmetic unit 537. The motor control system performs feedback control for feeding back the current value of the inverter 52. Although not shown, the motor control system 5 may also perform known arithmetic processing such as field weakening control. The motor control system 5 can suppress torque variation of the motor 1 by performing field weakening control.

The target q-axis current Iq _ target is input to the motor control system 5 from the outside. The increase or decrease of the motor output is externally instructed by the increase or decrease of the target q-axis current Iq _ target. The present motor control system 5 performs current limiting processing on the target q-axis current Iq _ target. Then, the present motor control system performs feedback control of subtracting the fed-back actual q-axis current value IQR from the current-limited q-axis current.

Then, the present motor control system 5 performs voltage control on the current deviation IQ _ err obtained by the feedback control. The voltage control arithmetic unit 533 calculates a voltage command value VQ1 from the current deviation IQ _ err, and further adds a non-interference element COR _ Q that suppresses interference between the d-axis and the Q-axis to the voltage command value VQ 1. Then, the induced voltage compensation operation unit 534 adds the compensation value for induced voltage compensation to the q-axis voltage command value VQ 2.

Further, the present motor control system 5 calculates a correction voltage value (torque ripple compensation value) for suppressing torque ripple in the torque ripple compensation calculation unit 531 based on the target q-axis current Iq _ target and the angular velocity ω. In the second embodiment, unlike the first embodiment, the motor control system 5 adds the voltage correction value to the output VQ3 (i.e., the sum of the VQ2 and the induced voltage compensation value) of the induced voltage compensation arithmetic unit 534. Thus, the motor control system 5 can suppress torque ripple in the motor 1 by adding a compensation value for suppressing torque ripple to the voltage command value for the inverter 52.

As described above, the motor control system 5 of the second embodiment performs the torque ripple compensation that compensates the responsiveness of the current controller in advance. That is, the motor control system 5 performs torque ripple compensation and advance compensation using feedback control. As in the first embodiment, the advance compensation is performed based on a parameter calculated using the angular velocity component and the reverse phase component of the order component of the torque ripple in the target current value. In the present embodiment, compensation is performed in which the gain is set to a value corresponding to the target current value and the phase is set to the current advance angle adjustment value, based on a relational expression in which the inverse characteristic of the current controller is divided into elements of the gain and the phase. By such compensation, the sensitivity of the motor control system to quantization noise and sensor noise associated with the high-pass filter operation can be reduced, and as a result, torque ripple generated in the motor can be reduced and deterioration of operating sound can be prevented. Moreover, the robustness of the control in the motor control system is also improved.

In the second embodiment, as in the first embodiment, torque ripple compensation is performed in which the compensation value is calculated from the reverse phase component of the order component of the torque ripple in the command current value, whereby the current controller responsiveness is improved in the low speed region of the motor. In the second embodiment as well, as in the first embodiment, the responsiveness of the current controller is improved in the high speed region of the motor by calculating the parameter from not only the anti-phase component of the current value but also the angular velocity ω. Therefore, the responsiveness of the current controller is improved even in a wide range from low speed to high speed.

Here, the first embodiment differs from the second embodiment in the following respects: the output from the torque ripple compensation operation portion 531 changes from a current value to a voltage value; and in response to this, the addition position in the control flow is changed. Accordingly, the output based on the torque fluctuation compensation is determined only by the electric characteristics of the motor, and thus, there is an advantage that the torque fluctuation can be easily adjusted. Further, by adding the compensation value for the torque ripple to the voltage value, there is also an advantage that the calculation processing is faster than the case of adding the voltage value to the current value.

The current control, the induced voltage compensation, the 2-axis/3-phase conversion, the dead time compensation, and the PWM control in the second embodiment are the same as those in the first embodiment, and therefore, the description thereof is omitted. In addition, in the second embodiment, these compensation and control may also apply known techniques. In the second embodiment, these compensation and control may not be performed, if necessary. The coupling of these elements may be regarded as the controller element c(s), or the coupling of only the main block for performing feedback control may be regarded as the controller element c(s). The coupling between the motor and the inverter is referred to as a device element p(s).

< other embodiments >

Next, other embodiments will be explained. The contents described in the other embodiments can be applied to any of the first and second embodiments.

Here, a motor that can be controlled by the above-described embodiment will be described in outline. Fig. 9a is a plan view of the first motor of the present embodiment, and fig. 9b is a plan view of the second motor of the present embodiment. The motor 1 shown in fig. 9a and 9b has a stator 2 and a rotor 3. As shown in fig. 9a and 9b, the motor 1 is of an inner rotor configuration. In addition, as the motor 1, an outer rotor structure may be adopted in addition to the inner rotor structure. The first motor 1 shown in fig. 9a is an IPM (interior permanent Magnet) motor, and the second motor 1 shown in fig. 9b is an SPM (Surface permanent Magnet) motor.

The stator 2 has a cylindrical outer shape extending in the axial direction. The stator 2 is disposed radially outward of the rotor 3 with a predetermined gap from the rotor 3. The stator 2 includes a stator core 21, an insulator 22, and a coil 23. The stator core 21 is a cylindrical member extending in the axial direction. The stator core 21 is formed by laminating a plurality of magnetic steel plates in the axial direction. The stator core 21 has a core back 21a and teeth (not shown). The core back 21a is a circular ring-shaped portion. The teeth extend radially inward from the inner circumferential surface of the core back 21 a. The teeth are arranged at predetermined intervals in the circumferential direction. In addition, the gap between adjacent teeth is referred to as a slot S. In the motor 1 shown in fig. 9a and 9b, for example, 12 grooves S are provided.

The rotor 3 has a cylindrical outer shape extending in the axial direction. The rotor 3 is disposed radially inward of the stator 2 with a predetermined gap from the stator 2. The rotor 3 includes a shaft 31, a rotor core 40, and a magnet 32. The rotor 3 rotates about a shaft 31 extending in the vertical direction (the direction perpendicular to the paper surface in fig. 9a and 9 b). The rotor core 40 is a cylindrical member extending in the axial direction. The shaft 31 is inserted into a hole 41d located at a radially central portion of the rotor core 40. The rotor core 40 is formed by laminating a plurality of magnetic steel plates in the axial direction. The magnet 32 is disposed inside the rotor core 40 in the first motor 1 shown in fig. 9a, and is attached to the surface of the rotor core 40 in the second motor 1 shown in fig. 9 b. A plurality of magnets 32 are arranged at predetermined intervals in the circumferential direction. In the motor 1 shown in fig. 9a and 9b, for example, 8 magnets 32 are provided. That is, in the motor 1 shown in fig. 9a and 9b, the number of poles P is 8.

The magnetic characteristics of the motor are different depending on the number of poles P and the number of slots S. Here, the factors that generate the operating sound include radial force and torque ripple. In the case of the motor of 8P12S in which the number of poles P is 8 and the number of slots S is 12, the radial force, which is the radial component of the electromagnetic force generated between the rotor and the stator, cancels each other, and thus torque ripple becomes a cause of a main operating sound. That is, the operating sound of the motor of 8P12S is efficiently reduced by compensating only the torque ripple by the above-described motor control system. Accordingly, the motor control system of the present invention is particularly useful in motors of 8P 12S.

The motor control system of the present invention is particularly useful in SPM motors because the cancellation of radial forces is particularly effective in SPM motors. More specifically, in the SPM motor, reluctance torque is not generated, and only magnet torque acts. Therefore, by adopting the present invention, only the magnet torque is compensated, thereby achieving vibration reduction. Conversely, the cancellation of the radial force is not limited to the effect produced in the SPM motor and the motor of 8P12S, but is also produced in the IPM motor or, for example, the 10P12S motor, and therefore the motor control system of the present invention is also useful in the IPM motor or, for example, the 10P12S motor.

Next, an outline of the electric power steering apparatus will be explained. As shown in fig. 10, in the present embodiment, a column-type electric power steering apparatus is exemplified. The electric power steering apparatus 9 is mounted on a steering mechanism of a wheel of an automobile. The electric power steering device 9 is a column type power steering device that directly reduces a steering force by the power of the motor 1. The electric power steering apparatus 9 includes a motor 1, a steering shaft 914, and an axle 913.

The steering shaft 914 transmits an input from a steering wheel 911 to an axle 913 having wheels 912. The power of the motor 1 is transmitted to the steering shaft 914 via a coupling 915 having a gear or the like. The motor 1 employed in the column-type electric power steering apparatus 9 is disposed inside an engine room (not shown). In addition, the electric power steering apparatus 9 shown in fig. 10 is of a column type as an example, but the power steering apparatus of the present invention may be of a rack type.

Here, in an application requiring low torque ripple and low operating sound like the electric power steering device 9, the following effects are provided: by controlling the motor 1 by the motor control system 5 described above, a compromise between low torque ripple and low operating sound is achieved. The reason for this is that torque ripple having a frequency exceeding the response of current control is compensated for by compensating the response of the current controller without using a high-pass filter for amplifying noise, thereby producing an effect of torque ripple compensation. Therefore, the present invention is particularly useful in a power steering apparatus.

The present invention is also useful for applications other than power steering devices. The present invention is useful for motors requiring a reduction in operating noise, such as a traction motor (a motor for running), a motor for a compressor, and a motor for an oil pump.

Hereinafter, a motor unit having a traction motor will be described.

In the following description, unless otherwise specified, a direction parallel to the motor axis J2 of the motor 102 is simply referred to as "axial direction", a radial direction with the motor axis J2 as a center is simply referred to as "radial direction", and a circumferential direction with the motor axis J2 as a center, that is, a direction around the motor axis J2 is simply referred to as "circumferential direction". However, the "parallel direction" also includes a substantially parallel direction. Fig. 11 is a conceptual diagram of the motor unit 100 having the traction motor, and fig. 12 is a side schematic diagram of the motor unit 100.

The motor unit 100 is mounted on a vehicle having a motor as a power source, such as a Hybrid Electric Vehicle (HEV), a plug-in hybrid electric vehicle (PHV), or an Electric Vehicle (EV), and is used as a power source. The motor unit 100 of the present embodiment includes a motor (main motor) 102, a gear portion 103, a housing 106, and a motor control system 5.

As shown in fig. 11, the motor 102 includes a rotor 120 that rotates about a motor axis J2 extending in the horizontal direction, and a stator 130 located radially outward of the rotor 120. A housing space 180 for housing the motor 102 and the gear portion 103 is provided inside the housing 106. The housing space 180 is divided into a motor chamber 181 housing the motor 102 and a gear chamber 182 housing the gear portion 103.

The motor 102 is housed in a motor chamber 181 of the housing 106. The motor 102 has a rotor 120 and a stator 130 located radially outward of the rotor 120. The motor 102 is an inner rotor type motor, and includes a stator 130 and a rotor 120 rotatably disposed inside the stator 130.

The rotor 120 is rotated by supplying electric power from a battery, not shown, to the stator 130 via the motor control system 5. The rotor 120 includes a shaft (motor shaft) 121, a rotor core 124, and a rotor magnet (not shown). The rotor 120 (i.e., the shaft 121, the rotor core 124, and the rotor magnet) rotates about a motor axis J2 extending in the horizontal direction. The torque of the rotor 120 is transmitted to the gear portion 103.

The shaft 121 extends with a motor axis J2 extending in the horizontal direction and the width direction of the vehicle as the center. The shaft 121 rotates about a motor axis J2.

The shaft 121 extends across the motor chamber 181 and the gear chamber 182 of the housing 106. One end portion of the shaft 121 protrudes to the gear chamber 182 side. A first gear 141 is fixed to an end of the shaft 121 protruding into the gear chamber 182.

The rotor core 124 is formed by laminating silicon steel plates (magnetic steel plates). The rotor core 124 is a cylindrical body extending in the axial direction. A plurality of rotor magnets are fixed to the rotor core 124.

The stator 130 surrounds the rotor 120 from the radially outer side. In fig. 11, a stator 130 has a stator core 132 and a coil 131. The stator 130 is held by the housing 106. Although not shown, the stator core 132 has a plurality of magnetic pole teeth radially inward from the inner circumferential surface of the annular yoke. A coil wire (not shown) is wound between the magnetic pole teeth to form a coil 31.

The gear portion 103 is housed in a gear chamber 182 of the housing 106. The gear portion 103 is connected to the shaft 121 on one axial side of the motor axis J2. The gear portion 103 has a reduction gear 104 and a differential gear 105. The torque output from the motor 102 is transmitted to the differential device 105 via the reduction gear 104.

The reduction gear 104 is connected to the rotor 120 of the motor 102. The reduction gear 104 has the following functions: the rotation speed of the motor 102 is reduced, and the torque output from the motor 102 is increased according to the reduction ratio. The reduction gear 104 transmits the torque output from the motor 102 to the differential 105.

The reduction gear 104 has a first gear (intermediate drive gear) 141, a second gear (intermediate gear) 142, a third gear (final drive gear) 143, and an intermediate shaft 145. The torque output from the motor 102 is transmitted to a ring gear (gear) 151 of the differential device 105 via a shaft 121 of the motor 102, a first gear 141, a second gear 142, an intermediate shaft 145, and a third gear 143.

The differential device 105 is connected to the motor 102 via the reduction gear 104. The differential device 105 is a device for transmitting the torque output from the motor 102 to the wheels of the vehicle. The differential device 105 has the following functions: the speed difference of the left and right wheels is absorbed when the vehicle turns, and the torque is transmitted to the axles 155 of the left and right wheels.

The motor control system 5 is electrically connected to the motor 102. The motor control system 5 supplies electric power to the motor 102 through an inverter. The motor control system 5 controls the current supplied to the motor 2. By compensating for the torque fluctuation with the motor control system 5, the operating sound of the motor 102 is reduced.

< reduction of operating sound in coupled system >

For example, when the motor 1 is incorporated in the electric power steering apparatus 9 shown in fig. 10, there is a possibility that acoustic vibration (noise) is generated due to resonance or the like caused by coupling of the motor 1 to other elements, and a sufficiently low operating noise cannot be realized only by the above-described reduction of the torque ripple. Hereinafter, an embodiment for reducing such sound vibration (noise) will be described. Fig. 13 is a schematic diagram of a motor control system according to a third embodiment, and fig. 14 is a schematic diagram of a control arithmetic unit according to the third embodiment.

The motor control system 5 according to the third embodiment includes a sound vibration compensation calculating unit 538 instead of the torque fluctuation compensation calculating unit 531 according to the first embodiment. Note that, although the example in which the sound vibration compensation calculating unit 538 is provided instead of the torque fluctuation compensation calculating unit 531 has been described here for the sake of simplicity of description, the motor control system 5 may include both the torque fluctuation compensation calculating unit 531 and the sound vibration compensation calculating unit 538.

As shown in fig. 14, the acoustic vibration compensation calculation unit 538 includes a storage unit 5382 in which a lookup table is stored and a reference unit 5381 which refers to the lookup table to obtain a compensation value. The lookup table stored in the storage section 5382 is a two-dimensional lookup table that uses the target q-axis current value Iq _ target and the rotation speed ω of the motor to obtain a reference value. The compensation value γ obtained by the acoustic vibration compensation calculating unit 538 can be obtained by the following equation (4).

γ＝Asin(Bθ+C)

A＝Lookuptable_A(Iq_target，ω)···(4)

C＝Lookuptable_C(Iq_target，ω)

That is, the gain a and the phase C are acquired as reference values in the lookup table stored in the storage section 5382. The number B of times of the electrical angle θ of the motor is a fixed value given to the lookup table, and is a number selected from k times and 1/k times (k is an integer). In other words, the lookup table is a recording table in which the compensation value γ of the order B is recorded. As a recording method of the compensation value γ, a recording method of an approximate expression based on the compensation value γ may be used in addition to the lookup table.

As the lookup table, a plurality of lookup tables different in the number of times B may be stored. In this case, the acoustic vibration compensation arithmetic unit 538 calculates each compensation value γ from the gain a and the phase C obtained from each lookup table using the above equation (4), and outputs a value obtained by adding the compensation values γ to each other as a compensation value. That is, the acoustic vibration compensation calculating unit 538 compensates for at least one of the k-th order component and the 1/k-th order component in the rotation of the motor. Fig. 15 is a diagram showing the lookup table stored in the storage section 5382. The storage unit 5382 stores a first table T1 obtained with the gain a as a reference value and a second table T2 obtained with the phase C as a reference value.

In each of the lookup tables T1 and T2, a change in the motor rotation speed ω corresponds to a row change, and a change in the target current value corresponds to a column change. That is, the higher the motor rotation speed ω is, the lower the row is referred to, and the larger the target current value is, the right-hand column is referred to. In the example shown in fig. 15, a lookup table of m rows and n columns is shown, but in general, the motor rotation speed ω or the target current value has a value equivalent to between rows or between columns. Therefore, the reference value can be obtained by, for example, linear interpolation from the values described in the look-up tables T1 and T2.

The reference unit 5381 shown in fig. 14 calculates the compensation value γ by substituting the gain a and the phase C obtained by referring to the lookup table into γ ═ Asin (B θ + C). As in the first embodiment, the compensation value γ is superimposed (added) on the target q-axis current Iq _ target before correction, which is output from the current limit calculation unit 532. By adding the compensation value γ, the motor control system 5 according to the third embodiment reduces acoustic vibration (noise) caused by resonance or the like in the coupling system. Hereinafter, a procedure of adjusting the compensation value γ for reducing the acoustic vibration (noise) due to resonance or the like and recording the compensation value γ in the form of a lookup table will be described. Fig. 16 is a diagram showing a process of adjusting the compensation value γ and recording.

First, in step S101, as a coupling system including a motor and a drive body coupled to and driven by the motor, noise is measured for the coupling system in which the motor is coupled to at least a part of the steering mechanism. Fig. 17 is a diagram schematically showing a state of noise measurement in step S101 in fig. 16.

Here, a state is shown in which the coupling system in which the motor 1 and the steering shaft 914 are coupled by the coupling 915 is provided in the soundproof room 6, and the coupling system is driven in the soundproof room 6 to measure the noise. Such measurement is preferably used for a column type power steering mechanism shown in fig. 10. In the case of a rack-type power steering mechanism, the compensation value γ can be obtained from the measurement result of the column-type power steering mechanism. Fig. 18 is a graph showing an example of noise data obtained by the noise measurement in step S101 of fig. 16.

The horizontal axis of the graph of fig. 18 represents the frequencies obtained by decomposing noise into frequency components, and the vertical axis represents the rotation speed of the motor. The magnitude of (the component of) noise is represented by the density of dots in the graph, and the darker the color of the dots, the greater the noise.

In the graph, regions R1, R2, and R3 in which points having high noise are connected in an oblique line are shown. These regions R1, R2, R3 are regions where the frequency of noise is proportional to the rotational speed of the motor, one region corresponding to one of the number of times of the order component of the motor rotation.

In addition, the graph also shows a region R4 in which points with high noise are concentrated in a band shape in the vicinity of a specific motor rotation speed. This region R4 corresponds to the resonance band of the coupling the motor and the steering mechanism.

When such noise data is obtained in step S101 of fig. 16, next, in step S102, the above-described number B is determined. That is, the number B of components contributing to noise compensation among the order components of the motor rotation is determined. The components contributing to noise compensation are, for example, components largely contained in noise.

As a method of determining the number of times, two methods are explained here. In the first method, the order components of the motor rotation are compared with each other to determine the number B of components having large noise. That is, the regions R1, R2, and R3 of each order component shown in fig. 18 are compared with each other to determine the number of times of the region having large noise.

In the second method, a frequency band in which noise is large is focused on a frequency band in which the rotational speed of the motor is large, and the frequency B of a component in which noise is large is determined in the frequency band. That is, focusing on the band-shaped region R4 shown in fig. 18, the number of times of the order component with large noise is specified in the band-shaped region R4.

In the case of the noise data shown in fig. 18, when either of the above-described two methods is used, 24 times (times in electrical angle) as the number of times corresponding to the region R1 shown in fig. 18 are determined. This determined number of times 24 corresponds to a common multiple (in particular, here, the smallest common multiple) of the number of poles (8) and the number of slots (12) of the motor 1 illustrated in fig. 9a, 9 b.

After the number of times B is determined in step S102, next, in step S103, adjustment of gain and phase is performed. That is, the drive of the motor 1 is controlled using the component of the determined number of times B (for example, 6 times in the mechanical angle in the case of 24 times in the electrical angle) as a compensation value, and the noise is reduced by adjusting the component value of the number of times B. More specifically, as the output value of the acoustic vibration compensation calculating unit 538 shown in fig. 13 and 14, the compensation value γ, Asin (B θ + C), which is based on the gain a and the phase C that are arbitrarily adjusted, is used. Here, the value of the order component B of the mechanical angle is used in the calculation of the compensation value γ. Then, under the control of the control arithmetic unit 53, the motor 1 is driven at the determined rotation speed at which the noise is large, and the gain a and the phase C are adjusted to values at which the noise level is reduced while monitoring the level of the noise. That is, as the component value of the degree B, the gain and the phase are adjusted. Fig. 19 is a diagram illustrating the monitored noise level. In fig. 19, the horizontal axis represents the frequency of noise, and the vertical axis represents the level of noise.

The noise waveform illustrated in fig. 19 corresponds to the noise waveform of a constant rotation speed in the region R4 in fig. 18, and the noise level indicated by the density of dots in fig. 18 is indicated by the height in the vertical axis direction in fig. 19. Each peak generated in the noise waveform of fig. 19 corresponds to each order component, and the position of the broken line shown in the vicinity of the right end in fig. 19 corresponds to the above-described determination number B (for example, 24 times in electrical angle). If the gain a and the phase C that determine the degree B are appropriately adjusted, not only the peak value of the degree B (for example, 24 in electrical angle) but also the noise waveform illustrated in fig. 19 as a whole is reduced.

That is, by appropriately adjusting the gain a and the phase C for the fixed number of times B, the noise of the entire region R4 corresponding to the resonance band of the coupling the motor and the steering mechanism is reduced (compensated).

In step S103 of fig. 16, the adjustment of the gain a and the phase C for reducing noise in this way is performed at a plurality of motor rotation speeds, respectively, resulting in a series of gains a and a series of phases C. Then, in step S104, the series of gains a and the series of phases C are recorded as 1 column of the lookup tables T1 and T2 shown in fig. 15. That is, the component value of the degree B is recorded as a table map.

In step S103, the above-described procedure is further repeated for each of the plurality of target q-axis currents Iq _ target, and in step S104, the gain a and the phase C are recorded in the respective columns of the look-up tables T1 and T2 shown in fig. 15.

After the gain a and the phase C are recorded in the look-up tables T1 and T2 in step S104, the look-up tables T1 and T2 are recorded (stored) in the memory 5382 of the sound vibration compensation arithmetic unit 538 in the motor control system 5 in step S105. The motor control system 5 is realized by, for example, a microcomputer, and the look-up tables T1 and T2 are recorded (stored) in a memory element of the microcomputer.

The acoustic vibration compensation calculating unit 538 calculates the compensation value γ by the above equation (4) using the look-up tables T1 and T2 recorded (stored) in this manner, thereby reducing noise generated in a coupling system that couples the motor and other elements.

As described above, a plurality of types of lookup tables with different times B may be stored as the lookup tables T1 and T2. The compensation value γ in this case may be, for example, a compensation value obtained by adding up the respective compensation values calculated from the reference values of the respective look-up tables as described above. Alternatively, the compensation value γ may be calculated by determining the number of times B by the method described in step S102 of fig. 16 and using the look-up tables T1 and T2 of the determined number of times B, for example. The noise data used for determining the number B is, for example, noise data measured in a vehicle room during steering driving. Fig. 20 and 21 are graphs showing the effect of compensation based on the compensation value γ.

The horizontal axis of fig. 20 and 21 represents the rotation speed of the motor, the vertical axis of fig. 20 represents torque ripple, and the vertical axis of fig. 21 represents noise. In fig. 20 and 21, the state without compensation is indicated by a broken line, and the state with compensation is indicated by a solid line.

As a result of the compensation based on the compensation value γ, as shown in fig. 20, the torque ripple is reduced in the entire region of the motor rotation speed. However, the peak around about 900 rpm is not so reduced. On the other hand, as shown in fig. 21, the noise is greatly reduced in the entire region of the motor rotation speed. It can therefore be seen that compensation based on the compensation value γ is particularly effective in noise reduction of the coupled system.

Such an action is considered to be a result of performing the advance control based on the phase C under the control condition suitable for reducing the noise, as compared with the control condition suitable for reducing the torque ripple of the motor. In other words, the compensation based on the compensation value γ uses the torque fluctuation of the motor to eliminate the resonance of the coupling system and the like.

Fig. 22 is a schematic diagram of a motor control system according to the fourth embodiment, and fig. 23 is a schematic diagram of a control arithmetic unit according to the fourth embodiment.

The motor control system 5 according to the fourth embodiment includes a sound vibration compensation calculating unit 538 instead of the torque fluctuation compensation calculating unit 531 according to the second embodiment.

As shown in fig. 23, the acoustic vibration compensation calculation unit 538 includes a storage unit 5382 in which a lookup table is stored and a reference unit 5381 which refers to the lookup table to obtain a compensation value. The acoustic vibration compensation calculation unit 538 calculates the gain a and the phase C using the target q-axis current value Iq _ target and the motor rotation speed ω, and calculates the compensation value γ Asin (B θ + C). Note that the compensation value γ calculated by the acoustic vibration compensation arithmetic unit 538 of the fourth embodiment is a compensation value added to the q-axis voltage command value VQ 4.

The lookup table stored in the storage unit 5382 of the fourth embodiment is also created and recorded by the same procedure as that shown in the flowchart of fig. 16. By the compensation based on the compensation value γ, the acoustic vibration (noise) generated by the coupling system in which the motor 1 is coupled to other elements is reduced as in the third embodiment.

As an example of the coupling system, although the coupling system for coupling the motor 1 and the elements of the steering mechanism is shown in the above description, for example, a motor unit having a traction motor shown in fig. 11 and 12 may be considered as an object of noise compensation based on the compensation value γ.

While the embodiment and the modification of the present invention have been described above, the configurations and combinations thereof in the embodiment and the modification are merely examples, and addition, omission, replacement, and other modifications of the configurations can be made within the scope not departing from the gist of the present invention. The present invention is not limited to the embodiments.

Industrial applicability

Embodiments of the present invention can be widely applied to various apparatuses having various motors, such as a dust collector, a dryer, a ceiling fan, a washing machine, a refrigerator, and a power steering apparatus.