阅读说明：本技术 一种电机直接转矩控制方法 (Direct torque control method for motor ) 是由 李云 诸自强 任远 马雅青 朱世武 詹姆斯·格林 李子健 于 2019-03-08 设计创作，主要内容包括：本发明提供了一种电机直接转矩控制方法,该方法通过对转矩误差进行比例积分调节来补偿定子磁链矢量角的方法来得到双三相永磁同步电机系统中的转矩相关的电压空间矢量给定值,通过比例积分控制器和二倍频谐振控制器来减小双三相永磁同步电机系统中电机和逆变器的固有不对称导致的谐波电压,进而有效抑制了该谐波电压产生的谐波电流。通过六倍频谐振控制器来减小双三相永磁同步电机系统中逆变器非性线因素导致的谐波电压,进而有效抑制了该谐波电压产生的谐波电流,并且在参考电压空间矢量的计算过程中没有引入额外的电机参数,有效增强了对电机参数的鲁棒性。(The invention provides a direct torque control method of a motor, which obtains a voltage space vector set value related to torque in a double three-phase permanent magnet synchronous motor system by a method of compensating a stator flux linkage vector angle by carrying out proportional integral regulation on a torque error, reduces harmonic voltage caused by inherent asymmetry of a motor and an inverter in the double three-phase permanent magnet synchronous motor system by a proportional integral controller and a double frequency resonance controller, and further effectively inhibits harmonic current generated by the harmonic voltage. Harmonic voltage caused by non-linear factors of an inverter in a double three-phase permanent magnet synchronous motor system is reduced through a six-frequency-multiplication resonance controller, harmonic current generated by the harmonic voltage is effectively restrained, extra motor parameters are not introduced in the calculation process of a reference voltage space vector, and robustness of the motor parameters is effectively enhanced.)

1. A method of direct torque control of an electric machine, comprising the steps of:

s10, acquiring six-phase stator voltage signals and six-phase stator current signals of the double three-phase motor, and performing coordinate transformation on the acquired signals to obtain α - β sub-plane stator voltage components, stator current components and z₁-z₂A stator current component of the harmonic sub-plane; acquiring the rotor speed and the rotor position of the motor;

s20: estimating a stator flux linkage, a stator flux linkage position, and an electromagnetic torque from the stator voltage component and the stator current component of the α - β sub-plane obtained in step S10;

s30: determining a given value of the electromagnetic torque by performing proportional integral calculation on the difference between the given rotor speed and the rotor speed obtained in step S10, and further determining a given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque; estimating a given value of a stator flux linkage amplitude corresponding to the rotor speed according to the direct-current bus voltage of the inverter and the rotor speed obtained in the step S10; taking the smaller value of the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque and the given value of the stator flux linkage amplitude corresponding to the rotor rotation speed as the reference value of the stator flux linkage amplitude;

s40: determining an angle increment of the stator flux linkage by performing a proportional integral calculation on a difference between a given value of the electromagnetic torque and the calculated value of the electromagnetic torque obtained in step S20, determining a reference value of the stator flux linkage according to the angle increment of the stator flux linkage, the reference value of the amplitude of the stator flux linkage determined in step S30, and the calculated value of the position of the stator flux linkage obtained in step S20, and determining a given value of a voltage space vector of the α - β sub-plane according to the reference value of the stator flux linkage, the calculated value of the stator flux linkage obtained in step S20, and a stator current component of the α - β sub-plane obtained in step S10;

s50: z obtained based on step S10₁-z₂Stator current component of harmonic sub-plane, obtaining z by harmonic current controller₁-z₂A given value of the voltage space vector of the harmonic sub-plane;

s60, using space vector pulse width modulation method to obtain the given value of the voltage space vector of α - β sub-plane obtained in step S40 and the given value of z obtained in step S50₁-z₂The given value of the voltage space vector of the harmonic sub-plane is modulated to generate a switching signal for controlling a switching tube of the inverter, so that the direct torque control of the motor is realized;

the harmonic current controller is configured to reduce harmonic voltages caused by inherent asymmetry of the motor and the inverter and harmonic voltages caused by nonlinear linear factors of the inverter, and further suppress harmonic currents generated by the harmonic voltages.

2. The motor direct torque control method according to claim 1, wherein the step S50 further includes the steps of:

s51, based on the rotor position obtained in step S10, for z obtained in step S10₁-z₂Stator current components of the harmonic sub-planes are subjected to synchronous rotation coordinate transformation to obtain z₁-z₂Stator current components under a harmonic sub-plane synchronous rotation coordinate system;

s52, based on z₁-z₂The condition that the reference value of the stator current component is zero in the harmonic sub-plane synchronous rotation coordinate system is based on z obtained in step S51₁-z₂Obtaining the stator current component in the harmonic sub-plane synchronous rotation coordinate system by using a harmonic current controller₁-z₂Given values of voltage space vectors under a sub-plane synchronous rotation coordinate system;

s53, for z obtained in step S52₁-z₂Carrying out synchronous rotation coordinate inverse transformation on a given value of a voltage space vector under a sub-plane synchronous rotation coordinate system to obtain z₁-z₂A given value of the voltage space vector of the harmonic sub-plane;

the harmonic current controller comprises a proportional integral resonance regulator and a six-frequency-doubling resonance regulator, and the resonance regulator of the proportional integral resonance regulator is a frequency-doubling resonance regulator.

3. The motor direct torque control method according to claim 1, wherein the step S50 further includes the steps of:

s51, based on the rotor position obtained in step S10, for z obtained in step S10₁-z₂The stator current component of the harmonic sub-plane is subjected to forward rotation coordinate transformation and reverse rotation coordinate transformation to obtain z₁-z₂Stator current components under a harmonic sub-plane forward rotation coordinate system and stator current components under a harmonic sub-plane reverse rotation coordinate system;

s52, based on z₁-z₂The condition that the reference value of the stator current component in the harmonic sub-plane forward rotation coordinate system is zero is based on z obtained in step S51₁-z₂Harmonic sub-planeObtaining the component of stator current in forward rotating coordinate system by using a proportional-integral regulator and a six-fold resonance regulator in a harmonic current controller₁-z₂A given value of a voltage space vector under a harmonic sub-plane forward rotation coordinate system;

s53, based on z₁-z₂On condition that the reference value of the stator current component in the harmonic sub-plane reverse rotation coordinate system is zero, z is obtained from step S51₁-z₂Obtaining a stator current component in a harmonic sub-plane reverse rotation coordinate system by using another proportional-integral regulator in the harmonic current controller₁-z₂A given value of a voltage space vector under a harmonic sub-plane reverse rotation coordinate system;

s54, for z obtained in step S52₁-z₂Performing reverse transformation of forward rotation coordinate on the given value of the voltage space vector in the sub-plane forward rotation coordinate system and performing reverse transformation on the z obtained in the step S53₁-z₂Carrying out reverse rotation coordinate inverse transformation on the given value of the voltage space vector under the sub-plane reverse rotation coordinate system, and adding the inverse transformation results of the given value and the voltage space vector to obtain z₁-z₂A given value of the voltage space vector of the harmonic sub-plane.

4. The direct torque control method of an electric motor according to claim 1, wherein the step S50 is specifically:

based on z₁-z₂The condition that the reference value of the harmonic sub-plane stator current component is zero is based on z obtained in step S10₁-z₂Stator current component of harmonic sub-plane, obtaining z by harmonic current controller₁-z₂A given value of the voltage space vector of the sub-plane;

wherein the harmonic current controller comprises a proportional and fundamental resonance adjuster, a quintuple frequency resonance adjuster and a heptatuple frequency resonance adjuster.

5. The motor direct torque control method according to any one of claims 1 to 4, characterized in that:

in step S30, after the given value of the electromagnetic torque is obtained, the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque is determined by looking up the flux linkage-torque table obtained according to the maximum torque current ratio algorithm.

6. The direct torque control method of the motor according to any one of claims 1 to 4, wherein in the step S20, the stator flux linkage ψ is calculated from the stator voltage component and the stator current component of α - β sub-planes obtained in step S10 by the following equation_sαβStator flux linkage position theta_sAnd electromagnetic torque T_e：

T_e＝3P|ψ_sαi_sβ-ψ_sβi_sα)

Wherein u is_sα，u_sβThe components of the stator voltage, i, on the α axis and the β axis, respectively, in the α - β sub-plane_sα，i_sβThe components of the stator current in the α and β axes in the α - β sub-planes, ψ_sα，ψ_sβThe components of the stator flux linkage in the α and β axes in the α - β sub-planes, R_sIs the stator resistance of the motor, and P is the number of pole pairs of the motor.

7. The direct torque control method for motor according to any one of claims 1 to 4, wherein in the step S30, the given value of the stator flux linkage amplitude corresponding to the rotor speed is estimated from the DC bus voltage of the inverter and the rotor speed obtained in the step S10 by:

|ψ_s2 ^*|＝U_max/ω_r

wherein, | ψ_s2 ^*I is the calculated value of stator flux linkage amplitude, omega_rAs the rotational speed of the rotor, V_dcIs the DC bus voltage of the inverter, U_maxThe maximum amplitude of the phase voltage allowed to be provided by the inverter is η a positive coefficient.

8. The direct torque control method of an electric motor according to any one of claims 1 to 4, wherein in the step S40, the reference value of the stator flux linkage is determined from the angle increment of the stator flux linkage, the reference value of the amplitude of the stator flux linkage determined in the step S30, and the calculated value of the position of the stator flux linkage obtained in the step S20 by:

wherein the content of the first and second substances,the components of the reference value of the stator flux linkage on the α axis and the β axis in the α - β sub-plane respectively,in increments of the angle of the stator flux linkage,is a reference value of stator flux linkage amplitude, theta_sIs the stator flux linkage position.

9. The direct torque control method of an electric motor according to claim 8, wherein in the step S40, the given value of the voltage space vector of the α - β sub-plane is determined from the reference value of the stator flux linkage and the calculated value of the stator flux linkage obtained in the step S20 and the stator current component of the α - β sub-plane obtained in the step S10 by the following formula:

wherein the content of the first and second substances,the components, i, of a given value of the voltage space vector of the α - β sub-planes on the α axis and the β axis in the α - β sub-planes, respectively_sα，i_sβThe components of the stator current in the α and β axes in the α - β sub-planes, R_sIs the stator resistance, T, of the motor_sIs the sampling period of the system.

10. The motor direct torque control method according to claim 2, wherein in said step S50,

by the formula to z₁-z₂Stator current component i of the harmonic sub-plane_sz1，i_sz2Performing synchronous rotation coordinate transformation to obtain z₁-z₂Stator current component i under harmonic sub-plane synchronous rotation coordinate system_szd，i_szq：

By the formula to z₁-z₂The given value of the voltage space vector under the harmonic sub-plane synchronous rotation coordinate system is subjected to synchronous rotation coordinate inverse transformation to obtain z₁-z₂Given value of the voltage space vector of the harmonic sub-plane:

wherein v is_szd ^*，v_szq ^*Are each z₁-z₂Component of a given value of a voltage space vector, v, in a harmonic sub-plane synchronous rotation coordinate system_sz1 ^*，v_sz2 ^*Are each z₁-z₂Voltage space vector of harmonic sub-planeComponent of a given value of, theta_rIs the rotor position.

Technical Field

The invention belongs to the technical field of motor control, and particularly relates to a motor direct torque control method applicable to a double three-phase permanent magnet motor system.

Background

The multi-phase motor alternating current transmission system has the advantages of low torque pulsation, good fault tolerance performance, more control freedom degree and the like. Particularly five-phase and six-phase motor systems. The double three-phase permanent magnet motor system is a six-phase motor system and comprises two sets of stator windings with the difference of 30-degree electrical angles, and the two sets of stator windings can be respectively supplied with power by adopting independent three-phase inverters. Therefore, the double three-phase permanent magnet motor system has the advantages of both the traditional permanent magnet motor and the multi-phase motor, and is widely researched and applied in recent years.

The voltage, current and flux linkage vectors of the double three-phase permanent magnet motor can be projected into three mutually orthogonal two-dimensional sub-planes through six-dimensional decoupling coordinate transformation, wherein all electromechanical energy conversion components are projected to α - β sub-planes, and non-electromechanical energy conversion components are projected to z₁-z₂Sub-plane and o₁-o₂A sub-plane. For two sets of double three-phase permanent magnet motor systems with independent winding neutral points, o₁-o₂The voltage vector of the sub-plane is a zero vector and therefore can be disregarded for control.

In a dual three-phase permanent magnet motor system, although only the voltage and current components of the α - β sub-planes are relevant to electromechanical energy conversion (i.e., torque formation), the z-direction is not the same₁-z₂The voltage and current components of the sub-planes are effectively controlled, even for small harmonic voltages in z₁-z₂The sub-planes form larger harmonic currents, which easily causes the heating of the motor to be increased and the system efficiency to be reduced. This feature of the dual three-phase permanent magnet motor system determines that two problems must be considered simultaneously when controlling the dual three-phase motor system: effective control of torque and effective suppression of harmonic currents.

Harmonic current at z from harmonic voltage₁-z₂Acting on the sub-plane. The formation of harmonic voltages is mainly due to three factors:

1) a method for generating an inappropriate inverter switching signal.

2) Asymmetry of the motor and the inverter themselves, such as phase-to-phase asymmetry of each set of windings of the motor or asymmetry between two sets of windings.

3) The nonlinear factor of the inverter is mainly dead zone action, wherein the dead zone can be regarded as a square wave voltage signal, and the signal is applied to z after decomposition₁-z₂The main components on the sub-planes are the 5 th and 7 th voltage harmonics.

Therefore, effective suppression of the harmonics of the dual three-phase permanent magnet motor system needs to be performed while considering the above three factors.

The method is simple in structure and easy to implement, but large torque ripple can be caused by adopting the hysteresis comparator and the switch table in the traditional direct torque control to generate the inverter switch signals, and meanwhile, only the voltage component of the α - β sub-plane is controlled, z is the voltage component, and z is the voltage component of the α - β sub-plane₁-z₂The voltage component of the sub-plane can generate larger harmonic current, so that the motor loss is increased, and the system efficiency is reduced.

The deadbeat direct torque control based on Sine Pulse Width Modulation (SPWM) of the double three-phase permanent magnet synchronous motor is to directly obtain a voltage reference vector from a flux linkage error and a torque error according to a motor model equation and output a switch tube state control signal of an inverter through an SPWM technology. Although the method adopts a Pulse Width Modulation (PWM) strategy, the inherent asymmetry of the motor and the inverter is not considered, and the nonlinear factor of the inverter is not considered, so that more obvious current harmonics exist. At the same time, inaccuracies in the motor parameters may result in inaccuracies in the torque control, due to the strong dependence of the selected control strategy on the motor parameters.

In view of the above, a new direct torque control method for a motor is needed to solve the above technical problems.

Disclosure of Invention

In view of the above technical problems, the present invention provides a new method for controlling direct torque of a motor, so as to solve the technical problem that the prior art cannot effectively control the torque of a dual three-phase permanent magnet synchronous motor system and effectively suppress harmonic current.

The technical scheme of the invention is realized by the following modes:

a method of direct torque control of an electric machine, comprising the steps of:

s10, acquiring six-phase stator voltage signals and six-phase stator current signals of the double three-phase motor, and performing coordinate transformation on the acquired signals to obtain α - β sub-plane stator voltage components, stator current components and z₁-z₂A stator current component of the harmonic sub-plane; acquiring the rotor speed and the rotor position of the motor;

s20: estimating a stator flux linkage, a stator flux linkage position, and an electromagnetic torque from the stator voltage component and the stator current component of the α - β sub-plane obtained in step S10;

s30: determining a given value of the electromagnetic torque by performing proportional integral calculation on the difference between the given rotor speed and the rotor speed obtained in step S10, and further determining a given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque; estimating a given value of a stator flux linkage amplitude corresponding to the rotor speed according to the direct-current bus voltage of the inverter and the rotor speed obtained in the step S10; taking the smaller value of the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque and the given value of the stator flux linkage amplitude corresponding to the rotor rotation speed as the reference value of the stator flux linkage amplitude;

s40: determining an angle increment of the stator flux linkage by performing a proportional integral calculation on a difference between a given value of the electromagnetic torque and the calculated value of the electromagnetic torque obtained in step S20, determining a reference value of the stator flux linkage according to the angle increment of the stator flux linkage, the reference value of the amplitude of the stator flux linkage determined in step S30, and the calculated value of the position of the stator flux linkage obtained in step S20, and determining a given value of a voltage space vector of the α - β sub-plane according to the reference value of the stator flux linkage, the calculated value of the stator flux linkage obtained in step S20, and a stator current component of the α - β sub-plane obtained in step S10;

s50: z obtained based on step S10₁-z₂Stator current component of harmonic sub-plane, obtaining z by harmonic current controller₁-z₂A given value of the voltage space vector of the harmonic sub-plane;

s60, using space vector pulse width modulation technique to obtain the given value of the voltage space vector of α - β sub-plane obtained in step S40 and the given value of z obtained in step S50₁-z₂The given value of the voltage space vector of the harmonic sub-plane is modulated to generate a switching signal for controlling a switching tube of the inverter, so that the direct torque control of the motor is realized;

the harmonic current controller is configured to reduce harmonic voltages caused by inherent asymmetry of the motor and the inverter and harmonic voltages caused by nonlinear linear factors of the inverter, and further suppress harmonic currents generated by the harmonic voltages.

According to the first embodiment of the present invention, the step S50 further includes the steps of:

s51, based on the rotor position obtained in step S10, for z obtained in step S10₁-z₂Stator current components of the harmonic sub-planes are subjected to synchronous rotation coordinate transformation to obtain z₁-z₂Stator current components under a harmonic sub-plane synchronous rotation coordinate system;

s52, based on z₁-z₂The condition that the reference value of the stator current component is zero in the harmonic sub-plane synchronous rotation coordinate system is based on z obtained in step S51₁-z₂Obtaining the stator current component in the harmonic sub-plane synchronous rotation coordinate system by using a harmonic current controller₁-z₂Given values of voltage space vectors under a sub-plane synchronous rotation coordinate system;

s53, for z obtained in step S52₁-z₂Carrying out synchronous rotation coordinate inverse transformation on a given value of a voltage space vector under a sub-plane synchronous rotation coordinate system to obtain z₁-z₂A given value of the voltage space vector of the harmonic sub-plane;

the harmonic current controller comprises a proportional integral resonance regulator and a six-frequency-doubling resonance regulator, and the resonance regulator of the proportional integral resonance regulator is a frequency-doubling resonance regulator.

According to the second embodiment of the present invention, the step S50 further includes the steps of:

s51, based on the rotor position obtained in step S10, for z obtained in step S10₁-z₂The stator current component of the harmonic sub-plane is subjected to forward rotation coordinate transformation and reverse rotation coordinate transformation to obtain z₁-z₂Stator current components under a harmonic sub-plane forward rotation coordinate system and stator current components under a harmonic sub-plane reverse rotation coordinate system;

s52, based on z₁-z₂The condition that the reference value of the stator current component in the harmonic sub-plane forward rotation coordinate system is zero is based on z obtained in step S51₁-z₂The stator current component under the harmonic sub-plane forward rotation coordinate system is obtained by a proportional-integral regulator and a six-time frequency resonance regulator in the harmonic current controller₁-z₂A given value of a voltage space vector under a harmonic sub-plane forward rotation coordinate system;

s53, based on z₁-z₂On condition that the reference value of the stator current component in the harmonic sub-plane reverse rotation coordinate system is zero, z is obtained from step S51₁-z₂Obtaining a stator current component in a harmonic sub-plane reverse rotation coordinate system by using another proportional-integral regulator in the harmonic current controller₁-z₂A given value of a voltage space vector under a harmonic sub-plane reverse rotation coordinate system;

s54, for z obtained in step S52₁-z₂Performing reverse transformation of forward rotation coordinate on the given value of the voltage space vector in the sub-plane forward rotation coordinate system and performing reverse transformation on the z obtained in the step S53₁-z₂Carrying out reverse rotation coordinate inverse transformation on the given value of the voltage space vector under the sub-plane reverse rotation coordinate system, and adding the inverse transformation results of the given value and the voltage space vector to obtain z₁-z₂A given value of the voltage space vector of the harmonic sub-plane.

According to the third embodiment of the present invention, the step S50 specifically includes:

based on z₁-z₂The condition that the reference value of the harmonic sub-plane stator current component is zero is based on z obtained in step S10₁-z₂Stator current component of harmonic sub-plane, obtaining z by harmonic current controller₁-z₂A given value of the voltage space vector of the sub-plane;

wherein the harmonic current controller comprises a proportional and fundamental resonance adjuster, a quintuple frequency resonance adjuster and a heptatuple frequency resonance adjuster.

According to an embodiment of the present invention, after obtaining the given value of the electromagnetic torque, the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque is determined by looking up the flux linkage-torque table obtained according to the maximum torque current ratio algorithm in step S30.

According to an embodiment of the present invention, in the step S20, the stator flux linkage ψ is calculated from the stator voltage component and the stator current component of α - β sub-planes obtained in step S10 by the following equation_sαβStator flux linkage position theta_sAnd electromagnetic torque T_e：

T_e＝3P(ψ_sαi_sβ-ψ_sβi_sα)

Wherein u is_sα，u_sβThe components of the stator voltage, i, on the α axis and the β axis, respectively, in the α - β sub-plane_sα，i_sβThe components of the stator current in the α and β axes in the α - β sub-planes, ψ_sα，ψ_sβThe components of the stator flux linkage in the α and β axes in the α - β sub-planes, R_sIs the stator resistance of the motor, and P is the number of pole pairs of the motor.

According to the embodiment of the present invention, in the step S30, the given value of the stator flux linkage amplitude corresponding to the rotor speed is estimated from the dc bus voltage of the inverter and the rotor speed obtained in the step S10 by the following formula:

|ψ_s2 ^*|＝U_max/ω_r

wherein, | ψ_s2 ^*I is the calculated value of stator flux linkage amplitude, omega_rAs the rotational speed of the rotor, V_dcIs the DC bus voltage of the inverter, U_maxThe maximum amplitude of the phase voltage allowed to be supplied to the inverter, η, is a positive coefficient less than 1 and close to 1.

According to the embodiment of the present invention, in the step S40, the reference value of the stator flux linkage is determined according to the angle increment of the stator flux linkage, the reference value of the amplitude of the stator flux linkage determined in the step S30, and the calculated value of the position of the stator flux linkage obtained in the step S20 by the following formula:

wherein the content of the first and second substances,the components of the reference value of the stator flux linkage on the α axis and the β axis in the α - β sub-plane respectively,is the angle increment of the stator flux linkage,Is a reference value of stator flux linkage amplitude and theta_sIs the stator flux linkage position.

According to an embodiment of the present invention, in the step S40, the given value of the voltage space vector of the α - β sub-plane is determined according to the reference value of the stator flux linkage and the calculated value of the stator flux linkage obtained in the step S20 and the stator current component of the α - β sub-plane obtained in the step S10 by the following formula:

wherein the content of the first and second substances,

the components, i, of a given value of the voltage space vector of the α - β sub-planes on the α axis and the β axis in the α - β sub-planes, respectively_sα，i_sβThe components of the stator current in the α and β axes in the α - β sub-planes, R_sIs the stator resistance, T, of the motor_sIs the sampling period of the system.

According to an embodiment of the present invention, in the step S50,

by the formula to z₁-z₂Stator current component i of the harmonic sub-plane_sz1，i_sz2Performing synchronous rotation coordinate transformation to obtain z₁-z₂Stator current component i under harmonic sub-plane synchronous rotation coordinate system_szd，i_szq：

By the formula to z₁-z₂The given value of the voltage space vector under the harmonic sub-plane synchronous rotation coordinate system is subjected to synchronous rotation coordinate inverse transformation to obtain z₁-z₂Given value of the voltage space vector of the harmonic sub-plane:

wherein v is_szd ^*，v_szq ^*Are each z₁-z₂Component of a given value of a voltage space vector, v, in a harmonic sub-plane synchronous rotation coordinate system_sz1 ^*，v_sz2 ^*Are each z₁-z₂Component of a given value of the voltage space vector of the harmonic sub-plane, theta_rIs the rotor position.

Compared with the prior art, the direct torque control method of the motor provided by the invention has the following advantages or beneficial effects:

1) the large torque ripple in the traditional direct torque control is effectively reduced.

The method for compensating the stator flux linkage vector angle by carrying out proportional integral adjustment on the torque error is utilized to obtain the given value of the voltage space vector related to the torque, and the switching control of the inverter is realized by combining the Space Vector Pulse Width Modulation (SVPWM) technology, so that the large torque pulsation in the traditional direct torque control is effectively reduced.

2) Effective suppression of harmonic currents

Harmonic voltage caused by inherent asymmetry of a motor and an inverter in a double three-phase permanent magnet synchronous motor system is reduced by utilizing a proportional-integral regulator, a double-frequency resonance regulator and the like, and harmonic current generated by the harmonic voltage can be effectively inhibited.

Harmonic voltage caused by non-linear factors (mainly dead zone influence) of the inverter in a double three-phase permanent magnet synchronous motor system is reduced by utilizing the six-frequency multiplication resonance regulator, and further harmonic current generated by the harmonic voltage can be effectively inhibited.

In summary, by simultaneously considering three main factors (improper inverter switching signal generation method, asymmetry of the motor and the inverter, inverter nonlinearity factor) forming harmonic current, the harmonic current is effectively suppressed.

3) The dependence of a control strategy on motor parameters is effectively reduced.

The voltage space vector related to the torque is obtained by a method of compensating a stator flux linkage vector angle by carrying out Proportional Integral (PI) adjustment on the torque error, and no additional motor parameter is needed except for all stator resistors in a flux linkage observer needed by direct torque control. As no additional motor parameters are introduced in the calculation process of the reference voltage space vector, the robustness of the motor parameters is effectively enhanced.

Additional features and advantages of the invention will be set forth in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. The objectives and other advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.

Drawings

The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention and not to limit the invention.

Fig. 1 is a schematic structural diagram of a double three-phase permanent magnet synchronous motor according to an embodiment of the invention;

fig. 2 is an equivalent circuit diagram of a dual three-phase permanent magnet synchronous motor driving system according to an embodiment of the present invention;

FIG. 3 is a schematic diagram of the direct torque control system of the motor according to the first embodiment of the present invention;

FIG. 4 is a flow chart of an operation for implementing a corresponding motor direct torque control method using the motor direct torque control system of FIG. 3;

FIG. 5 is a schematic diagram of a harmonic current controller in a direct torque control system for an electric machine according to a first embodiment of the present invention;

FIG. 6 is a schematic diagram of a harmonic current controller in a direct torque control system for an electric machine according to a second embodiment of the present invention;

fig. 7 is a design diagram of a harmonic current controller in a direct torque control system of a motor according to a third embodiment of the present invention.

Detailed Description

The following detailed description of the embodiments of the present invention will be provided with reference to the drawings and examples, so that how to apply the technical means to solve the technical problems and achieve the technical effects can be fully understood and implemented. It should be noted that, as long as there is no conflict, the embodiments and the features of the embodiments of the present invention may be combined with each other, and the technical solutions formed are within the scope of the present invention.

In the following description, for purposes of explanation, numerous specific details are set forth in order to provide a thorough understanding of the embodiments of the invention. It will be apparent, however, to one skilled in the art that the present invention may be practiced without some of these specific details or with other methods described herein.

It should be noted that the method provided by the present invention is not only applicable to a dual three-phase permanent magnet synchronous motor system, but also applicable to a dual three-phase asynchronous motor system (i.e. a dual three-phase induction motor system). The method provided by the invention is not only suitable for a double three-phase motor system (a permanent magnet synchronous motor or an asynchronous motor) but also suitable for a five-phase motor system (a permanent magnet synchronous motor or an asynchronous motor). The method provided by the invention is not only suitable for the electric running state of a double three-phase or five-phase motor system, but also suitable for the power generation running state of the double three-phase or five-phase motor system. The motor torque control method of the present invention will be described below by taking only a double three-phase permanent magnet synchronous motor as an example.

Fig. 1 is a schematic structural diagram of a dual three-phase permanent magnet synchronous motor in the prior art. It can be seen from the distribution of the stator windings that the stator windings are composed of two sets of conventional three-phase windings ABC and XYZ, each set of windings is connected in a Y shape, the corresponding internal windings are different from each other by 120 degrees in space, and the included angle between the corresponding phases of the two sets of three-phase windings is 30 degrees. Therefore, in terms of the design of a hardware circuit, the double three-phase permanent magnet synchronous motor is a six-phase system, in order to enable the stator flux linkage and the permanent magnet flux linkage to interact to generate constant electromagnetic torque, the phase difference of current of phase windings in each set of Y-shaped windings is 120 degrees, and the phase difference of corresponding phase current between the Y-shaped windings is 30 degrees.

As shown in fig. 2, in the driving system of the dual three-phase permanent magnet synchronous motor, the central points n and n' of the two sets of Y-shaped windings are independent from each other, the dual three-phase permanent magnet synchronous motor is powered by a voltage source inverter, the driving main circuit is formed by connecting two sets of three-phase system driving circuits in parallel with a common direct current bus, wherein 6 mutually independent currents flow in 6-phase stator windings.