High-speed switching motor drive control method

文档序号：1892883 发布日期：2021-11-26 浏览：13次中文

阅读说明：本技术 高速开关电机驱动控制方法 (High-speed switching motor drive control method ) 是由朱博徐攀腾严海健谷裕李建勋宋述波郑星星李倩杨学广于 2021-07-14 设计创作，主要内容包括：本申请涉及一种高速开关电机驱动控制方法,该方法包括：获取电机的运行参数；将运行参数作为反电势滑模观测器的输入,通过反电势滑模观测器确定电机的第一反电势；滤除第一反电势中的目标次谐波,得到第二反电势；根据第二反电势、运行参数和预设的目标参数,生成电机转矩控制信号,电机转矩控制信号用于控制电机输出稳定的转矩。采用本方法能够对反电势滑模观测器确定的第一反电势进行滤波处理,滤除目标次谐波,提高反电势的精确度。通过第二反电势、运行参数和预设的目标参数对电机进行直接转矩控制时,能够有效抑制非理想反电动势谐波和电机低速运行时电流换相所引起的转矩脉动,使得电机输出稳定的转矩。(The application relates to a high-speed switching motor drive control method, which comprises the following steps: acquiring operation parameters of a motor; taking the operation parameters as the input of a back-emf sliding-mode observer, and determining a first back-emf of the motor through the back-emf sliding-mode observer; filtering a target sub-harmonic in the first counter electromotive force to obtain a second counter electromotive force; and generating a motor torque control signal according to the second counter electromotive force, the operation parameter and a preset target parameter, wherein the motor torque control signal is used for controlling the motor to output stable torque. By adopting the method, the first counter electromotive force determined by the counter electromotive force sliding mode observer can be filtered, the target subharmonic can be filtered, and the accuracy of the counter electromotive force can be improved. When the direct torque control is carried out on the motor through the second counter electromotive force, the operation parameters and the preset target parameters, the torque pulsation caused by non-ideal counter electromotive force harmonic waves and current phase commutation when the motor operates at a low speed can be effectively inhibited, so that the motor outputs stable torque.)

1. A method of controlling a motor, the method comprising:

acquiring operation parameters of a motor;

taking the operating parameters as input of a back-emf sliding-mode observer, and determining a first back-emf of the motor through the back-emf sliding-mode observer;

filtering out a target sub-harmonic in the first counter electromotive force to obtain a second counter electromotive force;

and generating a motor torque control signal according to the second counter electromotive force, the operation parameter and a preset target parameter, wherein the motor torque control signal is used for controlling the motor to output stable torque.

2. The motor control method of claim 1, wherein the operating parameters include stator phase voltages and stator phase currents;

determining a first back emf of the electric machine by the back-emf sliding-mode observer using the operating parameter as an input to the back-emf sliding-mode observer, comprising:

converting the stator phase current into coordinate axis measuring current under a two-phase static coordinate system and converting the stator phase voltage into coordinate axis measuring voltage under the two-phase static coordinate system through a preset first transformation function;

taking the coordinate axis measurement current and the coordinate axis measurement voltage as the input of the back electromotive force sliding mode observer to obtain the back electromotive force of the motor under a two-phase static coordinate system;

determining a back emf in the two-phase stationary coordinate system as the first back emf.

3. The motor control method of claim 1, wherein filtering out a target sub-harmonic in the first back emf to obtain a second back emf comprises:

converting the first counter electromotive force into a counter electromotive force under a rotating coordinate system through a preset second transformation function;

filtering out second harmonic and sixth harmonic in the counter electromotive force under the rotating coordinate system to obtain the counter electromotive force after filtering treatment under the rotating coordinate system;

and converting the counter electromotive force after the filtering processing in the rotating coordinate system into the counter electromotive force after the filtering processing in the two-phase static coordinate system through the inverse transformation of the second transformation function, and determining the counter electromotive force after the filtering processing in the two-phase static coordinate system as the second counter electromotive force.

4. A method of controlling a motor according to any of claims 1-3, wherein the operating parameters include stator phase voltage, stator phase current and rotor angular velocity; the target parameters comprise a target torque and a target flux linkage amplitude;

generating a motor torque control signal according to the second back emf, the operating parameter and a preset target parameter, including:

determining a torque measurement from the second back emf and the rotor angular velocity; determining a stator flux linkage amplitude according to the stator phase voltage and the stator phase current;

determining a torque error value according to the torque measurement value and the target torque, and determining a flux error value according to the stator flux linkage amplitude and the target flux linkage amplitude;

and generating the motor torque control signal according to the torque error value and the flux linkage error value.

5. The motor control method of claim 4, wherein said determining a torque measurement based on said second back emf and said rotor angular velocity comprises:

converting the second counter electromotive force into a counter electromotive force measured value of the motor under a three-phase static coordinate system;

determining the torque measurement from the back emf measurement and the rotor angular velocity.

6. The method of claim 4, wherein determining a stator flux linkage magnitude from the stator phase voltage and the stator phase current comprises:

converting the stator phase current into coordinate axis measuring current under a two-phase static coordinate system, and converting the stator phase voltage into coordinate axis measuring voltage under the two-phase static coordinate system;

and determining the stator flux linkage amplitude according to the coordinate axis measured current and the coordinate axis measured voltage.

7. The method of claim 4, wherein generating the motor torque control signal based on the torque error value and the flux linkage error value comprises:

determining a first control parameter by taking the torque error value as the input of a torque hysteresis regulator, and determining a second control parameter by taking the flux error value as the input of a flux hysteresis regulator;

generating a corresponding switching signal according to the first control parameter and the second control parameter; the switching signal is used for instructing the inverter to output the motor torque control signal.

8. A method of controlling a motor according to any of claims 1-3, wherein the operating parameters include stator phase voltage, stator phase current, rotor angular velocity and rotor field position; the target parameter comprises a target torque;

generating a motor torque control signal according to the second back emf, the operating parameter and a preset target parameter, including:

determining a torque measurement from the second back emf and the rotor angular velocity;

determining a torque error value based on the torque measurement and the target torque;

and generating the motor torque control signal according to the torque error value and the rotor magnetic field position.

9. The method of claim 8, wherein generating the motor torque control signal based on the torque error value and the rotor magnetic field position comprises:

determining a third control parameter using the torque error value as an input to the torque hysteresis regulator;

generating a corresponding switching signal according to the third control parameter and the position of the rotor magnetic field; the switching signal is used for instructing the inverter to output the motor torque control signal.

10. A computer arrangement comprising a memory and a processor, the memory storing a computer program, characterized in that the processor realizes the steps of the motor control method according to any of claims 1 to 9 when executing the computer program.

Technical Field

The application relates to the technical field of motor control, in particular to a high-speed switching motor drive control method.

Background

In the power industry, motors are used as main devices for electric energy production, transmission and application, and are widely applied to the aspects of agriculture, industrial and mining enterprises, national defense, traffic and transportation industry, scientific culture, daily life and the like.

Taking the brushless dc motor as an example, the brushless dc motor has the advantages of simple structure, reliable operation and convenient maintenance of the ac motor, and also has the advantages of high operating efficiency, no excitation loss, good speed regulation performance and the like of the dc motor. In order to enable the brushless direct current motor to be applied to the field of precise driving, the brushless direct current motor is required to provide smaller torque ripple, however, the torque ripple caused by non-ideal back electromotive force harmonic waves and the torque ripple caused by phase commutation seriously affect the performance of the brushless direct current motor, and the loss of the brushless direct current motor is increased.

Therefore, a method for suppressing the generation of large torque ripple of the brushless dc motor is required.

Disclosure of Invention

In view of the above, it is necessary to provide a high-speed switching motor drive control method capable of suppressing torque ripple.

In one aspect, a motor control method is provided, and is characterized in that the method includes:

acquiring operation parameters of a motor;

taking the operation parameters as the input of a back-emf sliding-mode observer, and determining a first back-emf of the motor through the back-emf sliding-mode observer;

filtering a target sub-harmonic in the first counter electromotive force to obtain a second counter electromotive force;

In one embodiment, the operating parameters include stator phase voltages and stator phase currents;

determining a first back emf of the electric machine through the back emf sliding mode observer using the operating parameters as input to the back emf sliding mode observer, comprising:

taking the coordinate axis measured current and the coordinate axis measured voltage as the input of the back electromotive force sliding mode observer to obtain the back electromotive force of the motor under a two-phase static coordinate system;

the back emf in the two-phase stationary frame is determined as the first back emf.

In one embodiment, filtering out the target sub-harmonic in the first back-emf to obtain the second back-emf comprises:

converting the first counter electromotive force into a counter electromotive force under a rotating coordinate system through a preset second transformation function;

filtering second harmonic and sixth harmonic in the counter electromotive force under the rotating coordinate system to obtain the counter electromotive force after filtering treatment under the rotating coordinate system;

and converting the counter electromotive force after the filtering processing under the rotating coordinate system into the counter electromotive force after the filtering processing under the two-phase static coordinate system through the inverse transformation of the second transformation function, and determining the counter electromotive force after the filtering processing under the two-phase static coordinate system to be the second counter electromotive force.

In one embodiment, the operating parameters include stator phase voltage, stator phase current, and rotor angular velocity; the target parameters comprise a target torque and a target flux linkage amplitude;

generating a motor torque control signal according to the second back emf, the operating parameter and a preset target parameter, including:

determining a torque measurement value according to the second counter potential and the rotor angular speed; determining a stator flux linkage amplitude according to the stator phase voltage and the stator phase current;

determining a torque error value according to the torque measured value and the target torque, and determining a flux linkage error value according to the stator flux linkage amplitude and the target flux linkage amplitude;

and generating a motor torque control signal according to the torque error value and the flux linkage error value.

In one embodiment, determining a torque measurement based on the second back emf and the rotor angular velocity comprises:

converting the second counter electromotive force into a counter electromotive force measured value of the motor under a three-phase static coordinate system;

and determining the torque measured value by using the counter electromotive force measured value and the rotor angular speed.

In one embodiment, determining the stator flux linkage magnitude from the stator phase voltage and the stator phase current comprises:

and determining the stator flux linkage amplitude according to the coordinate axis measured current and the coordinate axis measured voltage.

In one embodiment, generating a motor torque control signal based on the torque error value and the flux linkage error value comprises:

In one embodiment, the operating parameters include stator phase voltages, stator phase currents, stator spatial position, and rotor angular velocity; the target parameter comprises a target torque;

generating a motor torque control signal according to the second back emf, the operating parameter and a preset target parameter, including:

determining a torque measurement value according to the second counter potential and the rotor angular speed;

determining a torque error value based on the torque measurement and the target torque;

and generating a motor torque control signal according to the torque error value and the stator space position.

In one embodiment, generating a motor torque control signal based on the torque error value and the spatial position of the stator comprises:

determining a third control parameter using the torque error value as an input to the torque hysteresis regulator;

generating a corresponding switching signal according to the third control parameter and the stator space position; the switching signal is used for instructing the inverter to output a motor torque control signal.

In another aspect, there is provided a motor control apparatus including:

the acquisition module is used for acquiring the operating parameters of the motor;

the back electromotive force determining module is used for taking the operation parameters as the input of the back electromotive force sliding mode observer and determining a first back electromotive force of the motor through the back electromotive force sliding mode observer;

the filtering module is used for filtering the target sub-harmonic in the first counter electromotive force to obtain a second counter electromotive force;

and the control module is used for generating a motor torque control signal according to the second counter electromotive force, the operation parameter and a preset target parameter, and the motor torque control signal is used for controlling the motor to output stable torque.

In another aspect, a computer device is provided, which includes a memory and a processor, the memory stores a computer program, and the processor implements the steps of any one of the motor control methods provided in the above aspect when executing the computer program.

In another aspect, a computer-readable storage medium is provided, on which a computer program is stored, which when executed by a processor implements the steps of any one of the motor control methods provided in the above-mentioned aspect.

According to the high-speed switch motor drive control method, the operation parameters of the motor are obtained; taking the operation parameters as the input of a back-emf sliding-mode observer, and determining a first back-emf of the motor through the back-emf sliding-mode observer; filtering a target sub-harmonic in the first counter electromotive force to obtain a second counter electromotive force; and generating a motor torque control signal according to the second counter electromotive force, the operation parameter and a preset target parameter, wherein the motor torque control signal is used for controlling the motor to output stable torque. That is, the first back electromotive force determined by the back electromotive force sliding mode observer has harmonic waves, and the accuracy of the back electromotive force can be improved by filtering out the target subharmonic. And when the direct torque control is carried out on the motor through the second back electromotive force, the operation parameters and the preset target parameters, the torque pulsation caused by non-ideal back electromotive force harmonic waves and the torque pulsation caused by current phase commutation when the motor operates at a low speed can be effectively inhibited, so that the motor outputs stable torque.

Drawings

FIG. 1a is a schematic diagram of a motor control system according to an embodiment of the present application;

FIG. 1b is a schematic diagram of another exemplary motor control system of the present application;

FIG. 2 is a schematic flow chart of a motor control method according to an embodiment of the present application;

FIG. 3 is a schematic flow chart of a motor control method according to another embodiment of the present application;

FIG. 4 is a block diagram of a back-emf sliding-mode observer in an embodiment of the present application;

FIG. 5 is a schematic flow chart of a motor control method according to another embodiment of the present application;

FIG. 6 is a block diagram of the structure of an ANF filtering module in the back-emf sliding-mode observer according to an embodiment of the present application;

FIG. 7 is a block diagram of an adaptive notch filter according to an embodiment of the present application;

FIG. 8 is a schematic flow chart of a motor control method according to another embodiment of the present application;

FIG. 9 is a schematic flow chart of a motor control method according to another embodiment of the present application;

FIG. 10 is a block diagram of a motor control apparatus according to an embodiment of the present application;

FIG. 11 is a diagram illustrating an internal structure of a computer device in one embodiment.

Detailed Description

In order to make the objects, technical solutions and advantages of the present application more apparent, the present application is described in further detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative of the present application and are not intended to limit the present application.

Before explaining the control method of the motor provided by the present application, technical terms and application backgrounds related to the present application will be explained.

Brushless DC motor: a Brushless Direct Current Motor (BLDCM) is a new type of dc Motor developed along with the rapid development of electronic technology based on the conventional Motor. Compared with the traditional direct current motor, the brushless direct current motor uses electronic commutation to replace the mechanism commutation of the original direct current motor, and an electric brush and a phase converter are cancelled; the stator and the rotor in the original direct current motor are reversed, namely an armature winding is arranged on the stator and is conveniently connected with a static electronic commutation circuit, the armature winding is excited on the rotor and is a permanent magnet, the excitation winding is not needed, and a slip ring and an electric brush for electrifying the rotor are also not needed; in the original DC motor, the commutator is on the rotor, which can ensure the current change direction when the armature conductor is transferred from one stator magnetic pole to the other, the change of current direction in the armature winding of the brushless DC motor is controlled by the switch of the power tube, in order to ensure the switch signal and the position where the rotor magnetic pole is rotated, the sensor for detecting the rotor position angle is added in the motor control system.

The stator windings of the motor body are generally made in multiple phases (three, four, five, etc.). The rotor is composed of permanent magnetic steel according to a certain number of pole pairs. The position detection device is connected with the rotating shaft of the motor. When a certain phase of stator winding is electrified, the magnetic field produced by said current and the magnetic field produced by magnetic pole are interacted to drive rotor to rotate, then the position of rotor is converted into electric signal by position sensor to control inverter so as to make the stator windings be conducted according to a certain order, and the stator phase current can be changed in phase according to a certain order with the change of rotor position. Since the conduction sequence of the inverter is synchronized with the rotor angle, the same commutation as the mechanical commutator is performed.

Mathematical model of brushless dc motor: the process of establishing a differential equation model is described by taking a three-phase bridge type Y-connected brushless direct current motor as an example. The stator winding of the motor is a Y-connection concentrated integral pitch winding, the rotor adopts a non-salient pole inner rotor structure, and 3 Hall sensors are symmetrically arranged at intervals of 120 degrees. On the basis of this structure, the following assumptions are made:

(1) neglecting the saturation of the motor iron core and not counting the eddy current loss and the magnetic hysteresis loss;

(2) ignoring armature reaction, the air gap magnetic field distribution is approximately considered as a trapezoidal wave with a flat top width of 120 degrees of electrical angle;

(3) neglecting the cogging effect, the armature conductors are continuously and uniformly distributed on the surface of the armature;

(4) and the power tube and the freewheeling diode of the inverter circuit of the driving system have ideal switching characteristics.

The voltage balance equation for the three phase windings of the stator can be expressed as:

wherein u is_A、u_BAnd u_CFor stator three-phase winding voltage, R_sIs stator winding resistance, i_A、i_BAnd i_CFor stator three-phase winding currents, P is a differential operator, L_A、L_BAnd L_CFor self-inductance of the three-phase winding of the stator, L_AB、L_AC、L_BA、L_BC、L_CAAnd L_CBFor three-phase winding mutual inductance, e_A、e_BAnd e_CAn electromotive force is induced to the three-phase winding.

For the surface-mounted rotor structure, the self inductance and the mutual inductance are considered to be constant values, and are independent of the rotor position, namely, the following relations exist:

L_A＝L_B＝L_C＝L_s (2)

L_AB＝L_AC＝L_BA＝L_BC＝L_CA＝L_CB＝M (3)

wherein L is_sThe self inductance of each phase winding is obtained, and M is the mutual inductance of the phase windings.

Because of the fact that

i_A+i_B+i_C＝0 (4)

Therefore, it is not only easy to use

Mi_B+Mi_C＝-Mi_A (5)

From the above equations (4) and (5), the voltage balance equation of the three-phase winding of the brushless dc motor stator of the above equation (1) can be expressed as:

in the above formula:

wherein u is_A、u_BAnd u_CIs terminal voltage of the machine, u_nIs the neutral point voltage.

When the phase-change operation is not performed, if i, j two phases are turned on (i, j is A, B or C, and i ≠ j), it can be obtained from the above equations (4), (6) and (7):

when the phase change works, the following can be obtained:

the counter-electromotive force waveform of the brushless DC motor is trapezoidal wave, the counter-electromotive force is a quantity related to the space position angle, and the expression of the counter-electromotive force e can be written according to a piecewise function form, wherein the expression is expressed by e_AFor example, the following steps are carried out:

wherein e is_AIs the back-emf of the A-phase stator, k_eIs the back electromotive force coefficient, omega, of the motor_rIs the electrical angular velocity of the permanent magnet rotor, theta is the angle between the rotor and the coordinate axis a.

e_BAnd e_CRespectively lag behind e_A120 deg. and 240 deg. electrical angles.

Further, the torque equation of the motor is:

wherein e is_A、e_BAnd e_CBeing the back-emf of the motor, i_A、i_BAnd i_CFor stator phase current, ω is the mechanical angular velocity of the motor.

In order to generate a constant electromagnetic torque, i.e., a motor that can output a stable torque, the input is required to be a square-wave stator current, or when the stator current is a square wave, the back emf waveform is required to be a trapezoidal wave. And the duration of the square wave current is 120 electrical degrees in each half cycle, then the plateau portion of the trapezoidal wave back emf is also 120 electrical degrees and the two should be strictly synchronized. At any time, only two phases of the stator are conducting.

Direct torque control: direct Torque Control (DTC) is another high-dynamic-performance variable-voltage variable-frequency speed Control system of an alternating current motor developed after vector Control. The direct torque control technology uses a space vector analysis method to directly calculate and control the torque of an alternating current motor under a stator coordinate system, adopts stator magnetic field orientation, generates a pulse width modulation signal by means of discrete two-point regulation (Bang-Bang control), and directly performs optimal control on the switching state of an inverter so as to obtain high dynamic performance of the torque.

The direct torque control technology is to directly analyze a mathematical model of the alternating current motor and control the flux linkage and the torque of the motor under a stator coordinate system, adopts stator flux linkage magnetic field orientation, and can observe the stator flux linkage by using a back electromotive force integration method as long as the stator resistance is known.

That is, the direct torque control is a direct torque control, which directly controls the torque not by controlling the equivalent amount of current and flux linkage to indirectly control the torque, but by directly using the torque as a controlled amount. Therefore, the direct torque control emphasizes the torque control effect from the viewpoint of controlling the torque, and adopts the concept of a discrete voltage state and a hexagonal flux linkage locus or an approximately circular flux linkage locus. The control mode of the direct torque control technology for directly controlling the torque is to make a hysteresis comparison between a torque detection value and a torque set value through a torque two-point type regulator so as to limit the torque fluctuation within a certain tolerance range.

The most serious influence is caused by the torque ripple caused by non-ideal back electromotive force harmonic waves and the torque ripple caused by commutation of the motor, aiming at the cogging torque ripple of the brushless direct current motor, the torque ripple caused by non-ideal back electromotive force waveforms and the torque ripple caused by commutation.

And as can be seen from the above analysis, the commutation torque ripple of the motor is related to the electrical angular velocity of the permanent magnet rotor and the dc voltage across the inverter. If the direct-current voltage is not changed, the motor is in a low-speed working condition, the non-commutation phase current of the motor is increased, and the commutation torque is increased; when the motor is in a high-speed working condition, the non-commutation phase current of the motor is reduced, so that the commutation torque is reduced. During the phase change, when the phase current of the non-phase change is not changed, the torque of the motor is not fluctuated. Therefore, the motor can not generate torque pulsation only by adjusting the direct-current voltage in time through a direct torque control method.

Based on this, the back electromotive force of the motor is determined by the sliding mode observer and the self-adaptive filter, so that the torque pulsation is better inhibited while higher harmonics are filtered, the calculated amount is greatly reduced, and the accuracy and the stability of motor control are improved while the torque pulsation is inhibited.

After the technical terms and application scenarios related to the scheme of the present application are introduced, a description will be given to an applicable system of the motor control method of the present application with reference to fig. 1a and 1 b.

The motor control method provided by the application can be applied to a motor control system shown in fig. 1a or fig. 1 b. In the motor control system 100, the inverter 160 supplies power to the motor, thereby ensuring the normal operation of the motor. As shown in fig. 1a, the motor control system 100 includes a parameter acquisition device 110, a sliding-mode observer 120, a filter 130, a hysteresis regulator 140, a PID controller 150, and an inverter 160.

The Control of the motor comprises a rotational speed Control outer ring and a torque Control inner ring, the rotational speed Control outer ring is provided with a Proportional-Integral-Derivative (PID) controller 150, in particular, a PI controller. The PID controller 150 is configured to determine a target torque of the motor based on a difference between the rotor angular velocity of the motor and a preset target angular velocity.

The parameter collector 110 is used for collecting operation parameters of the motor, and may be disposed in the motor, or disposed at an output end or an input end of the motor, wherein the parameter collector is disposed at the input end of the motor and is used for collecting stator phase current and phase voltage of the motor, and the parameter collector is disposed at the output end of the motor and is used for collecting rotor angular velocity of the motor, or disposed in the motor and is used for obtaining a rotor magnetic field position. The parameter collector 110 may be an encoder, a resolver, or a hall sensor.

The sliding-mode observer 120 is used to estimate the back emf of the motor based on the collected stator phase voltages, phase currents and rotor angular velocity. The sliding mode surface of the sliding mode observer can be designed in advance according to the running state of a controlled motor, and the designed sliding mode observer can be directly used when the motor torque is controlled.

Because the back electromotive force is calculated according to the motor operating parameters, other noise signals are easily mixed in the data acquisition process, and harmonic components are contained in the back electromotive force. Thus, the filter 130 is provided to filter out the target sub-harmonic in the back emf to improve the accuracy of the back emf estimation.

In one possible implementation, as shown in fig. 1a, torque control may be controlled by two hysteresis regulators, including: a hysteresis regulator 140 for controlling torque and a hysteresis regulator 140 for controlling flux linkage, the two hysteresis regulators 140 outputting motor torque control signals according to the input torque error value and flux linkage error value. The applied voltage vector is selected based on the motor torque control signal and the space voltage vector to achieve direct control of the motor torque.

Since the brushless dc motor includes a position sensor, the torque control of the brushless dc motor can be performed by one hysteresis regulator 140 in addition to the control method shown in fig. 1 a.

In another possible implementation, as shown in fig. 1b, a hysteresis regulator 140 for controlling torque outputs a motor torque control signal based on the torque error value. The applied voltage vector is selected based on the motor torque control signal, the rotor field position and the space voltage vector to achieve direct control of the motor torque.

The inverter 160 is powered by a three-phase voltage source, and the output three-phase power is used for driving the brushless direct current motor to operate. That is, the inverter is directly controlled to output the corresponding space voltage vector by selecting a proper voltage vector, and then the output torque of the motor is directly controlled.

Based on the motor control system 100, in one possible implementation manner, the sliding mode observer 120, the filter 130, the PID controller 150, the hysteresis regulator 140, the voltage space vector and other corresponding functional modules and algorithms are integrated into one Digital Signal Processing (DSP) controller to be implemented, and a software program embedded in the DSP calls the corresponding functional units, and the specific algorithms and the like to work. The specific implementation process refers to the following embodiment corresponding to fig. 2. After the DSP outputs a control signal, the corresponding space voltage vector is selected according to the control signal to control the inverter, so that the motor outputs stable torque.

Based on the above motor control system, the motor control method of the present application will be explained with reference to the drawings.

In one embodiment, as shown in fig. 2, there is provided a motor control method, which may be applied to the motor control system shown in fig. 1a or fig. 1b, the method comprising the steps of:

step 210: and acquiring the operating parameters of the motor.

The operation parameters of the motor comprise phase current, phase voltage, rotor angular speed and rotor magnetic field position of a stator in the motor.

Referring to the motor control system shown in fig. 1a, the operation parameters of the motor may specifically be: three-phase current i of stator_A、i_B、i_CThree-phase voltage u of stator_A、u_B、u_CAnd rotor angular velocity ω.

Referring to the motor control system shown in fig. 1b, the operation parameters of the motor may specifically be: three-phase current i of stator_A、i_B、i_CThree-phase voltage u of stator_A、u_B、u_CThe spatial position of the rotor HA, HB, HC, and the rotor angular velocity ω.

Step 220: and determining a first counter electromotive force of the motor by using the operating parameters as input of a counter electromotive force sliding mode observer.

Since the back electromotive force of the motor cannot be directly measured, it is important to estimate the back electromotive force of the motor according to the operation parameters of the motor.

The sliding mode observation can be carried out in a dynamic process, and the system is forced to move according to a state track of a preset 'sliding surface' according to the current state (such as deviation, each order derivative thereof and the like) of the system in a purposeful and continuous change mode. The sliding surface can be designed and is irrelevant to the parameters and the disturbance of an object, so that the sliding mode control has the advantages of quick response, insensitive corresponding parameter change and disturbance, no need of system online identification, simple physical realization and the like.

Therefore, the present application employs a pre-designed back-emf sliding-mode observer to determine the back-emf of the motor. The back electromotive force sliding mode observer builds a construction model according to an actual model of the brushless direct current motor, is connected with the actual model in parallel, takes the state error of the actual model of the motor and the built construction model equal to zero as a control target, selects a switching function to continuously adjust parameters of the construction model, and extracts a back electromotive force parameter value of the motor after the back electromotive force parameter value is stable, namely an observed value.

Specifically, algorithms such as a state equation, a sliding mode surface equation, a switching function and a back electromotive force calculation equation of the brushless direct current motor are stored in the back electromotive force sliding mode observer, and the back electromotive force of the motor can be predicted through the back electromotive force sliding mode observer under the condition that the operation parameters of the motor are input. Here, the back electromotive force output by the back electromotive force sliding mode observer is determined as a first back electromotive force.

Step 230: and filtering the target subharmonic in the first counter electromotive force to obtain a second counter electromotive force.

Wherein the target subharmonic may be predetermined. The second counter potential is the filtered counter potential, which is subsequently used to determine a torque measurement of the electric machine.

As an example, fourier transform analysis (FFT) is continuously performed on the voltage of the motor, and it is known that 3 rd, 5 th, and 7 th harmonic components in the collected motor voltage are more, and the existence of the harmonic component affects the estimation result of the motor torque, so that the harmonic in the first back electromotive force is filtered after the first back electromotive force is determined, and the final torque calculation result is more accurate.

Step 240: and generating a motor torque control signal according to the second counter electromotive force, the operation parameter and a preset target parameter, wherein the motor torque control signal is used for controlling the motor to output stable torque.

The preset target parameters comprise a target torque and a target flux linkage amplitude.

Specifically, a torque measurement of the electric machine may be calculated based on the second back emf and an operating parameter of the electric machine; according to the operation parameters of the motor, the stator flux linkage amplitude of the motor can be calculated.

Referring to fig. 1a, in one possible implementation manner, the implementation process of step 240 is: a torque measurement of the electric machine is determined by the second counter-potential. A hysteresis regulator for controlling the torque, outputting a first control signal according to a difference between a target torque and a torque measurement value; and the hysteresis regulator for controlling the flux linkage outputs a second control signal according to the difference between the target flux linkage amplitude and the stator flux linkage amplitude. And selecting the applied voltage vector based on the first control signal, the second control signal and the space voltage vector to realize direct control of the motor torque, so that the motor can output stable torque.

As shown in fig. 1b, in another possible implementation manner, the implementation process of step 240 is: a torque measurement of the electric machine is determined by the second counter-potential. A hysteresis regulator for controlling the torque, outputting a first control signal according to a difference between a target torque and a torque measurement value; based on the first control signal, the rotor field position and the space voltage vector, the applied voltage vector is selected to achieve direct control of the motor torque so that the motor can output a stable torque.

In the motor control method, the operation parameters of the motor are obtained; taking the operation parameters as the input of a back-emf sliding-mode observer, and determining a first back-emf of the motor through the back-emf sliding-mode observer; filtering a target sub-harmonic in the first counter electromotive force to obtain a second counter electromotive force; and generating a motor torque control signal according to the second counter electromotive force, the operation parameter and a preset target parameter, wherein the motor torque control signal is used for controlling the motor to output stable torque. That is, the first back electromotive force determined by the back electromotive force sliding mode observer has harmonic waves, and the accuracy of the back electromotive force can be improved by filtering out the target subharmonic. And when the direct torque control is carried out on the motor through the second back electromotive force, the operation parameters and the preset target parameters, the torque pulsation caused by non-ideal back electromotive force harmonic waves and the torque pulsation caused by current phase commutation when the motor operates at a low speed can be effectively inhibited, so that the motor outputs stable torque.

Based on the corresponding embodiment shown in fig. 2, in an embodiment, as shown in fig. 3, the operation parameters include a stator phase voltage and a stator phase current, and a specific implementation manner of determining the first back electromotive force of the electric machine through the back electromotive force sliding mode observer by using the operation parameters as an input of the back electromotive force sliding mode observer (i.e., the step 220) includes the following steps:

step 310: and converting the stator phase current into coordinate axis measuring current under a two-phase static coordinate system and converting the stator phase voltage into coordinate axis measuring voltage under the two-phase static coordinate system through a preset first conversion function.

Wherein the first transformation function is a Clark transformation or 3/2 transformation for transforming a rotating vector from a three-phase stator coordinate system (a-B-C coordinate system) to a stator two-phase stationary coordinate system (α - β coordinate system). The above-mentioned coordinate axis current includes the α -axis measurement current i_αAnd beta axis measuring current i_β(ii) a The coordinate axis measurement voltage comprises an alpha axis measurement voltage u_αAnd beta axis measurement voltage u_β。

Stator phase current i_A、i_B、i_CConverting the measured current into coordinate axis measured current i under a two-phase static coordinate system_α、i_βThe process comprises the following steps:

the three-phase voltage u of the stator_A、u_B、u_CConverting into coordinate axis measurement voltage u under two-phase static coordinate system_α、u_βThe process comprises the following steps:

step 320: and taking the coordinate axis measured current and the coordinate axis measured voltage as the input of the counter electromotive force sliding mode observer to obtain the counter electromotive force of the motor under the two-phase static coordinate system, and determining the counter electromotive force under the two-phase static coordinate system as the first counter electromotive force.

In one possible implementation, the sliding-mode observer uses the following formula in determining the first back emf:

the voltage equation of the brushless direct current motor under an alpha-beta coordinate system is as follows:

wherein R is_sIs stator winding resistance, e_αCounter-potential of stator in alpha-axis, e_βCounter-potential of stator in beta axis, L_sThe self inductance of each phase winding is obtained, and M is the mutual inductance of the phase windings.

The state equation of the brushless direct current motor is as follows:

wherein the coefficient matrixCoefficient matrixStator current vector i ═ i_α i_β]^ΤDerivative of stator current vectorStator voltage vector u ═ u [ u ]_α u_β]^ΤMotor back electromotive force e ═ e_α e_β]^Τ。

The sliding mode surface equation is:

in the formula, "-" represents an error of the corresponding variable, and "^" represents an estimated value of the corresponding variable.

Taking phase A as an example to design an observer, and measuring a value u according to a coordinate axis_α、u_β、i_α、i_βStator resistor R of brushless DC motor_sAnd a stator inductance L_sM, establishing an equation of the brushless direct current motor back electromotive force sliding mode observer as follows:

the error dynamic equation of the back-emf sliding-mode observer is as follows:

where G is a continuous smooth arctangent function permutation sign function as proposed herein. The expression is g (x) arctan (x); k is a sliding mode gain value and is a constant.

According to the Lyapunov function stability theorem, the following can be deduced:

therefore, when K is a positive number which is large enough, the system enters the sliding mode surface to move, the error of the measured value of the stator current approaches to 0, and the error of the back electromotive force of the winding converges to 0.

By using the designed back electromotive force sliding mode observer, the first back electromotive force of the motor can be output under the condition that the coordinate axis current and the coordinate axis voltage under the two-phase static coordinate system are input and the stability of the back electromotive force sliding mode observer is met.

As an example, a structural block diagram of a back-emf sliding-mode observer is designed as shown in FIG. 4. Because a large amount of switching noise is often mixed in the observed first back emf signal, the first back emf signal needs to be filtered, and the low-pass filtering step is performed by a back emf sliding mode observer. Where τ is the filter time constant.

In the embodiment of the application, the designed back-emf sliding-mode observer has smaller ripple and has better inhibition effect on jitter generated by sliding-mode motion. The back electromotive force sliding mode observer can estimate the back electromotive force of the motor according to the stator phase current and the stator phase voltage of the motor, and filter noise signals in the estimated back electromotive force to obtain a first back electromotive force, so that the back electromotive force estimation accuracy is improved.

Based on the above embodiment, since a large amount of high-frequency switching noise is mixed in the first back-emf signal, the effect of filtering by using simple band-pass filtering in the back-emf sliding-mode observer is not good, and the high-order harmonics in the first back-emf signal still affect the back-emf calculation result.

Based on this, an Adaptive Notch Filter (ANF) is arranged at the output end of the back electromotive force sliding mode observer, and is used for filtering out the target higher harmonic in the first back electromotive force.

It should be noted that the trap is a filter that strongly attenuates signals at a specific frequency, that is, a band-stop filter with a very narrow band-stop bandwidth. In a conventional digital trap design, in order to achieve a sufficient attenuation for a signal at a certain frequency, it is common practice to select the order to be high enough to achieve a large attenuation; but at the same time the amount of calculations becomes larger. The working principle of the self-adaptive trap wave device provided by the application is as follows: knowing the frequency of the interfering signal (e.g., the most common 50Hz line frequency interference) in the original signal, it is only necessary to know the phase and amplitude of the interfering signal to completely "reproduce" the interfering signal, which is then subtracted from the original signal to obtain the desired signal component.

In addition, in specific implementation, the filter algorithm of the ANF may be stored in the back electromotive force sliding mode observer, the first back electromotive force is directly predicted by the back electromotive force sliding mode observer, and the second back electromotive force is obtained by filtering the higher harmonic in the first back electromotive force signal. The filtering process can be realized by an ANF (adaptive neural network) or a back electromotive force sliding mode observer without limitation.

Based on the corresponding embodiment shown in fig. 2, in an embodiment, as shown in fig. 5, a specific implementation manner of filtering out the target sub-harmonic in the first counter-electromotive force to obtain the second counter-electromotive force (i.e., the step 230) includes the following steps:

step 510: and converting the first counter electromotive force into a counter electromotive force under a rotating coordinate system through a preset second transformation function.

Wherein the second transformation function is a Park transformation or a rotation transformation for transforming a rotation vector from a two-phase stationary coordinate system (α - β coordinate system) to a two-phase rotating coordinate system (d-q coordinate system). The transformation relationship is as follows:

and theta is an included angle between the d axis of the d-q rotating coordinate system and the alpha axis of the alpha-beta coordinate system, namely the d axis is an included angle with the axis of the A-phase winding.

Mixing the above i_αAnd i_βConversion to i by Park transformation_dAnd i_qThe above u is introduced_αAnd u_βConversion to u by Park transformation_dAnd u_qAccording to i_d、i_q、u_dAnd u_qThe back electromotive force under the rotating coordinate system can be calculated.

In addition, it should be noted that, since the ANF can eliminate specific harmonics in the back electromotive force, and in practical applications, higher harmonics have a greater influence on the result, based on the analysis of the output waveform, it is found that after the first back electromotive force is low-pass filtered, the main higher harmonics still exist as second harmonics and sixth harmonics, and therefore, the second harmonics and sixth harmonics in the first back electromotive force need to be filtered out by using the ANF.

Step 510 is not a necessary step when using ANF filtering. That is, the filtering can be performed in a conventional α - β stationary coordinate system and in a reverse direction, and thus, 3 filtering links are required to perform filtering on the back electromotive force sliding mode observer by using the ANF.

Because the number of harmonic components determines the number of filter links in each branch, 3 filtering links are needed for filtering under the traditional alpha-beta static coordinate system, and only 2 filtering links are needed for filtering the back electromotive force sliding mode observer by using the ANF under the d-q coordinate system instead, so that the calculation amount is greatly reduced.

As an example, the block diagram of the structure of the ANF filtering module of the back-emf sliding-mode observer is shown in FIG. 6, e_A、e_BAnd e_CIn order to be the first counter-potential,andfor the filtered second back emf, abc/dq indicates the conversion from the three-phase stationary coordinate system to the two-phase rotating coordinate system, which can be achieved by the Clark transformation and the Park transformation. dq/abc denotes the conversion from a two-phase rotating coordinate system to a three-phase stationary coordinate system, which can be achieved by means of a Park inverse transform and a Clark inverse transform. ω is the motor rotor angular velocity.

The filtering includes two links for filtering out the first counter-potential e_dSecond harmonic and sixth harmonic in the filter to obtain a filterFiltering the first counter-potential e_qSecond harmonic and sixth harmonic in the filter to obtain a filter

Based on this, in order to reduce the amount of calculation, when performing the filtering process, the coordinate system conversion may be performed based on step 510, and then the filtering operation of step 520 described below may be performed.

Step 520: and filtering second harmonic and sixth harmonic in the counter electromotive force under the rotating coordinate system to obtain the counter electromotive force after filtering treatment under the rotating coordinate system.

As an example, a block diagram of the ANF structure is shown in fig. 7.

The transfer function of the ANF is:

where ω is the motor rotor angular velocity and h is the order of the major harmonic components; ξ is the damping coefficient. When the damping coefficient xi is larger, the dynamic response of the ANF filter is very quick, but the overshoot is larger; when the damping coefficient ξ is small, the overshoot amount of the ANF is effectively controlled, but the adjustment time becomes long. Through simulation analysis, the damping coefficient xi suitable for designing the ANF is 0.7.

Step 530: and converting the counter electromotive force after the filtering processing under the rotating coordinate system into the counter electromotive force after the filtering processing under the two-phase static coordinate system through the inverse transformation of the second transformation function, and determining the counter electromotive force after the filtering processing under the two-phase static coordinate system to be the second counter electromotive force.

Wherein the second transformation function is Park transformation or rotation transformation, and the inverse transformation is inverse Park transformation, and is used for transforming the filtered counter electromotive force under the rotation coordinate systemAndback electromotive force after filtering treatment under conversion into two-phase static coordinate systemAnd andi.e. the second counter potential.

In the embodiment of the application, the second harmonic and the sixth harmonic in the first counter electromotive force can be effectively filtered through the ANF, so that the counter electromotive force determination result is more accurate.

Based on the above-described corresponding embodiment of fig. 2 and the motor control system shown in fig. 1a, in one embodiment, as shown in fig. 8. The operating parameters include stator phase voltage, stator phase current, and rotor angular velocity; the target parameters comprise a target torque and a target flux linkage amplitude; a specific implementation of generating the motor torque control signal (i.e., the above step 240) according to the second back emf, the operating parameter, and the preset target parameter includes the steps of:

step 810: determining a torque measurement value according to the second counter potential and the rotor angular speed; and determining the stator flux linkage amplitude according to the stator phase voltage and the stator phase current.

In one possible implementation, the determination of the torque measurement is: and converting the second counter electromotive force into a counter electromotive force measured value of the motor under a three-phase static coordinate system, and determining a torque measured value by using the counter electromotive force measured value and the rotor angular speed.

The conversion of the second counter electromotive force into the counter electromotive force measurement value of the motor in the three-phase static coordinate system can be realized by Clark inverse conversion, and the second counter electromotive force is converted into the counter electromotive force measurement value of the motor in the three-phase static coordinate systemIs converted intoAnd

as one example, the torque measurement may be determined by the following equation:

wherein the content of the first and second substances,andfor a second counter-potential of the machine in a three-phase stationary frame, i_A、i_BAnd i_CFor stator phase currents, ω is the rotor angular velocity of the machine.

In one possible implementation, the stator flux linkage amplitude is determined by: converting the stator phase current into coordinate axis measuring current under a two-phase static coordinate system, and converting the stator phase voltage into coordinate axis measuring voltage under the two-phase static coordinate system; and determining the stator flux linkage amplitude according to the coordinate axis measured current and the coordinate axis measured voltage.

The conversion of stator phase voltage and phase current into coordinate axis measurement current in a two-phase static coordinate system can be realized by Clark conversion, and i is converted_A、i_B、i_CConversion to i_α、i_βWill u_A、u_B、u_CConversion to u_α、u_β。

As one example, the stator flux linkage amplitude may be determined by the following equation:

wherein psi_sIs the stator flux linkage amplitude phi psi in an alpha-beta static coordinate system_sα、ψ_sβFor the alpha and beta components of the stator flux linkage in the stationary frame, R_sIs the stator winding resistance.

Step 820: a torque error value is determined based on the torque measurement and the target torque, and a flux error value is determined based on the stator flux linkage amplitude and the target flux linkage amplitude.

In one possible implementation, the torque error value and the flux linkage error value may be determined by a comparator. The torque error value is a difference value between the torque measured value and the target torque, and the flux linkage error value is a difference value between the stator flux linkage amplitude and the target flux linkage amplitude.

Step 830: and generating a motor torque control signal according to the torque error value and the flux linkage error value.

Brushless dc motors generally adopt a two-by-two power-on mode, that is, 2 power tubes are switched on at every moment, and are commutated once at an electrical angle of 60 degrees, 1 power tube is commutated at every moment, and each power tube is switched on at an electrical angle of 120 degrees. Therefore, the voltage space vector can be represented by a six-bit binary number, and 6 non-zero voltage space vectors V1(100001), V2(001001), V3(011000), V4(010010), V5(000110), V6(100100), and 1 zero voltage space vector V0(000000) are obtained, where each bit from left to right represents the state of the upper and lower arm switching signals corresponding to the phases a, B, and C, respectively. The zero voltage space vector represents that the voltage at the stator terminal of the motor is zero when all the power tubes are turned off. The 6 non-zero voltage space vectors are mutually spaced by 60 degrees, and a three-phase coordinate system of the A-B-C stator is divided into 6 sectors, such as a space 6-sector division table shown in the following table 1:

TABLE 1

Electrical angle theta_eCan be determined by the following equation:

based on the analysis, the direct torque control is to perform hysteresis comparison control on the torque error value and the flux linkage error value, and then select a space voltage vector according to the output value of the hysteresis regulator and the current position of the stator flux linkage to realize the direct control on the motor torque.

In a possible implementation manner, the implementation process of the step 830 is as follows: determining a first control parameter by taking the torque error value as the input of a torque hysteresis regulator, and determining a second control parameter by taking the flux error value as the input of a flux hysteresis regulator; and generating a corresponding switching signal according to the first control parameter and the second control parameter, wherein the switching signal is used for indicating the inverter to output a motor torque control signal.

The flux linkage adopts three-value hysteresis comparison control, and the flux linkage hysteresis regulator output quantity psi_QWhen "1" indicates that the flux linkage is to be increased, the output amount ψ_QWhen "0", it means that the flux linkage is kept constant and the output quantity ψ is_QA value of "-1" indicates that the flux linkage is to be reduced; the torque is controlled by binary hysteresis comparison, and the output quantity T of the torque hysteresis regulator_QWhen "1", it means that the torque is to be increased, the output T is_QA "0" indicates that the torque is to be reduced. Phi (hand-to-hand)_Q、T_QThe space voltage vector determined together with the sector where the stator flux linkage is located is represented by a table, which is a switching table for direct torque control of the brushless dc motor, as shown in table 2 (a brushless dc motor torque control signal table):

TABLE 2

As can be seen from table 2, in any sector, if the actual stator flux linkage amplitude is equal to the target flux linkage amplitude, a non-zero vector and a zero vector are used to control the increase or decrease of the torque; when the actual stator flux linkage amplitude is smaller than the target flux linkage amplitude, increasing the flux linkage by using a non-zero vector; when the actual stator flux linkage amplitude is larger than the target flux linkage amplitude, the stator flux linkage is reduced by using another non-zero vector, and the selection of the space voltage vector is realized, so that the torque control of the brushless direct current motor can be realized.

In the embodiment of the application, the output psi of the flux linkage hysteresis loop regulator is determined_QOutput T of torque hysteresis regulator_QAnd the sector where the stator flux linkage is located, the conduction condition of a power tube in the inverter is determined in a table look-up 2 mode, the inverter is driven to be conducted to supply power to the motor, the torque pulsation can be restrained, and the motor can output stable torque.

Based on the above-mentioned corresponding embodiment of fig. 2 and the motor control system shown in fig. 1b, in another embodiment, as shown in fig. 9. The operation parameters comprise stator phase voltage, stator phase current, rotor magnetic field position and rotor angular speed; the target parameter comprises a target torque; a specific implementation of generating the motor torque control signal (i.e., the above step 240) according to the second back emf, the operating parameter, and the preset target parameter includes the steps of:

step 910: a torque measurement is determined based on the second back emf and the rotor angular velocity.

The specific implementation process can be referred to step 810 and formula (24), and is not described herein again.

Step 920: a torque error value is determined based on the torque measurement and the target torque.

In one possible implementation, a torque error value, i.e., the difference between the torque measurement and the target torque, may be determined by a comparator.

Step 930: and generating a motor torque control signal according to the torque error value and the rotor magnetic field position.

Because the brushless direct current motor is provided with a position sensor (generally, a Hall element is adopted as the position sensor), the motor selects a corresponding voltage space vector according to different logic combinations of 3 output signals HA, HB and HC of the Hall element, and the given voltage space vector can just generate a hexagonal flux linkage on a motor stator so as to realize continuous electric operation.

In a possible implementation manner, the implementation procedure of step 930 is: and determining a third control parameter by taking the torque error value as the input of the torque hysteresis regulator, and generating a corresponding switching signal according to the third control parameter and the position of the rotor magnetic field, wherein the switching signal is used for indicating the inverter to output the motor torque control signal.

When the brushless DC motor is applied with a non-zero basic voltage vector, the stator flux linkage moves along the direction of the applied voltage vector at the speed of the magnitude of the applied basic voltage vector; when the applied excitation is a zero voltage vector, the flux linkage is stationary. Therefore, by judging the area of the stator flux linkage and applying the corresponding basic voltage vector, the stator flux linkage can move along the hexagonal track formed by the connecting lines of the vertexes of the voltage vector, and the self-control of the stator flux linkage is realized.

Rotor magnetic field positions HA, HB and HC are collected through three Hall magnetic pole position sensors. Because the switching phase-locked Hall element is adopted, each output Hall signal is a square wave with the electric angle of 180 degrees and the width centered at the position where the Hall magnetic pole position sensor is placed. Thus, the position of the permanent magnet rotor flux linkage can be judged according to the Hall signal. And then the stator flux linkage is oriented to the rotor magnetic field, and the position of the stator flux linkage in the same phase is obtained. Therefore, the self-control of the stator flux linkage of the brushless direct current motor can be realized only by applying corresponding voltage according to the rotor magnetic field position detected by the Hall signal.

Specifically, the torque is controlled by adopting three-value hysteresis comparison, and when the output quantity T of the torque hysteresis regulator is_QWhen "1", it means that the torque is to be increased, the output T is_QWhen "0", it means that torque is to be maintained, the output T is_QA "-1" indicates that the torque is to be reduced. Phi (hand-to-hand)_QThe rotor magnetic field position and the space voltage vector determined by the sector corresponding to the rotor magnetic field position are represented by a table, namely a switch table for the direct torque control of the brushless DC motor, such as the torque control signal of the brushless DC motor shown in the following table 3Shown in the figure table:

TABLE 3

Referring to table 3, taking the stator flux linkage located in sector i as an example, when a voltage V2(001001) is applied, the stator flux linkage moves in the vertex direction of V6 to V1 and simultaneously drives the rotor to rotate counterclockwise, and the clockwise torque generated by the machine increases. Conversely, if voltage V5(000110) is applied, the motor generates a clockwise torque; when the zero voltage vector V0(000000) is applied, the motor output torque is zero. Similarly, other sectors have the same conclusions.

Therefore, as can be seen from table 3, when the torque of the motor is controlled, the output value T of the torque hysteresis regulator can be adjusted according to the sector in which the stator flux linkage is located_QThe applied voltage vector is selected so that dynamic direct control of the motor torque is achieved.

In the embodiment of the application, the output value T of the torque hysteresis regulator is used_QAnd the stator flux linkage is located in a sector, the conduction condition of a power tube in the inverter is determined by looking up a table 3, the inverter is driven to be conducted to supply power to the motor, the torque pulsation can be inhibited, and the motor can output stable torque.

It should be understood that, although the steps in the method flow chart corresponding to the above-described embodiment are sequentially displayed as indicated by the arrow, the steps are not necessarily sequentially executed as indicated by the arrow. The steps are not performed in the exact order shown and described, and may be performed in other orders, unless explicitly stated otherwise. Moreover, at least a part of the steps in the method flowcharts corresponding to the above embodiments may include multiple steps or multiple stages, which are not necessarily performed at the same time, but may be performed at different times, and the order of performing the steps or stages is not necessarily sequential, but may be performed alternately or alternately with other steps or at least a part of the steps or stages in other steps.

In one embodiment, as shown in fig. 10, there is provided a motor control apparatus 1000 including: an acquisition module 1010, a back emf determination module 1020, a filtering module 1030, and a control module 1040, wherein:

an obtaining module 1010, configured to obtain an operating parameter of a motor;

a back-emf determining module 1020, configured to determine a first back-emf of the motor through the back-emf sliding-mode observer using the operating parameter as an input of the back-emf sliding-mode observer;

the filtering module 1030 is configured to filter a target sub-harmonic in the first counter electromotive force to obtain a second counter electromotive force;

the control module 1040 is configured to generate a motor torque control signal according to the second back electromotive force, the operating parameter, and a preset target parameter, where the motor torque control signal is used to control the motor to output a stable torque.

In one embodiment, the operating parameters include a stator phase voltage and a stator phase current sum;

the back emf determination module 1020 is further configured to:

the back emf in the two-phase stationary frame is determined as the first back emf.

In one embodiment, the filtering module 1030 is further configured to:

converting the first counter electromotive force into a counter electromotive force under a rotating coordinate system through a preset second transformation function;

the control module 1040 includes:

a parameter determination submodule 1041 for determining a torque measurement value based on the second back emf and the rotor angular velocity; determining a stator flux linkage amplitude according to the stator phase voltage and the stator phase current;

the error determining submodule 1042 is used for determining a torque error value according to the torque measured value and the target torque, and determining a flux error value according to the stator flux linkage amplitude and the target flux linkage amplitude;

and the control submodule 1043 is configured to generate a motor torque control signal according to the torque error value and the flux linkage error value.

In one embodiment, the parameter determining submodule 1041 is further configured to:

converting the second counter electromotive force into a counter electromotive force measured value of the motor under a three-phase static coordinate system;

and determining the torque measured value by using the counter electromotive force measured value and the rotor angular speed.

In one embodiment, the parameter determining submodule 1041 is further configured to:

and determining the stator flux linkage amplitude according to the coordinate axis measured current and the coordinate axis measured voltage.

In one embodiment, the control sub-module 1043 is further configured to:

In one embodiment, the operating parameters include stator phase voltage, stator phase current, rotor angular velocity, and rotor magnetic field position; the target parameter comprises a target torque;

the control module 1040 includes:

a parameter determination submodule 1041 for determining a torque measurement value based on the second back emf and the rotor angular velocity;

an error determination submodule 1042 for determining a torque error value based on the torque measurement and the target torque;

and the control submodule 1043 is configured to generate a motor torque control signal according to the torque error value and the rotor magnetic field position.

In one embodiment, the control sub-module 1043 is further configured to:

determining a third control parameter using the torque error value as an input to the torque hysteresis regulator;

In the embodiment of the application, the motor control device acquires the operation parameters of the motor; taking the operation parameters as the input of a back-emf sliding-mode observer, and determining a first back-emf of the motor through the back-emf sliding-mode observer; filtering a target sub-harmonic in the first counter electromotive force to obtain a second counter electromotive force; and generating a motor torque control signal according to the second counter electromotive force, the operation parameter and a preset target parameter, wherein the motor torque control signal is used for controlling the motor to output stable torque. That is, the first back electromotive force determined by the back electromotive force sliding mode observer has harmonic waves, and the accuracy of the back electromotive force can be improved by filtering out the target subharmonic. And when the direct torque control is carried out on the motor through the second back electromotive force, the operation parameters and the preset target parameters, the torque pulsation caused by non-ideal back electromotive force harmonic waves and the torque pulsation caused by current phase commutation when the motor operates at a low speed can be effectively inhibited, so that the motor outputs stable torque.

In the motor control device provided in the above embodiment, when the motor is directly controlled by torque, only the division of the above functional modules is taken as an example, and in practical applications, the above function distribution may be completed by different functional modules according to needs, that is, the internal structure of the device is divided into different functional modules to complete all or part of the above described functions. In addition, each module in the above-described motor control device may be entirely or partially implemented by software, hardware, or a combination thereof. The modules can be embedded in a hardware form or independent from a processor in the computer device, and can also be stored in a memory in the computer device in a software form, so that the processor can call and execute operations corresponding to the modules.

It can be understood that the motor control device and the motor control method provided by the above embodiments belong to the same concept, and the specific implementation process thereof is detailed in the above stable motor control method embodiment, and is not described again here.

In one embodiment, a computer device is provided, which may be a terminal, and its internal structure diagram may be as shown in fig. 11. The computer device includes a processor, a memory, a communication interface, a display screen, and an input device connected by a system bus. The memory stores a computer program, and the processor implements all or part of the flow in the motor control method embodiment when executing the computer program.

In particular, the processor of the computer device is used to provide computing and control capabilities. The memory of the computer device comprises a nonvolatile storage medium and an internal memory. The non-volatile storage medium stores an operating system and a computer program. The internal memory provides an environment for the operation of an operating system and computer programs in the non-volatile storage medium. The communication interface of the computer device is used for carrying out wired or wireless communication with an external terminal, and the wireless communication can be realized through WIFI, an operator network, NFC (near field communication) or other technologies. The computer program is executed by a processor to implement a motor control method. The display screen of the computer equipment can be a liquid crystal display screen or an electronic ink display screen, and the input device of the computer equipment can be a touch layer covered on the display screen, a key, a track ball or a touch pad arranged on the shell of the computer equipment, an external keyboard, a touch pad or a mouse and the like.

Any reference to memory, storage, database, or other medium used in various embodiments of the motor control methods provided herein may include at least one of non-volatile and volatile memory. Non-volatile Memory may include Read-Only Memory (ROM), magnetic tape, floppy disk, flash Memory, optical storage, or the like. Volatile Memory can include Random Access Memory (RAM) or external cache Memory. By way of illustration and not limitation, RAM can take many forms, such as Static Random Access Memory (SRAM) or Dynamic Random Access Memory (DRAM), among others.

Those skilled in the art will appreciate that the architecture shown in fig. 11 is merely a block diagram of some of the structures associated with the disclosed aspects and is not intended to limit the computing devices to which the disclosed aspects apply, as particular computing devices may include more or less components than those shown, or may combine certain components, or have a different arrangement of components.

In an embodiment of the present application, a computer-readable storage medium is provided, on which a computer program is stored, which, when being executed by a processor, implements the flow of the above-mentioned respective motor control method embodiments.

Specifically, all or part of the flow of the motor control method embodiments may be implemented by a computer program that may be stored in a non-volatile computer-readable storage medium and that, when executed, may include the flow of the motor control method embodiments.

The technical features of the above embodiments can be arbitrarily combined, and for the sake of brevity, all possible combinations of the technical features in the above embodiments are not described, but should be considered as the scope of the present specification as long as there is no contradiction between the combinations of the technical features.

The above-mentioned embodiments only express several embodiments of the present application, and the description thereof is more specific and detailed, but not construed as limiting the scope of the invention. It should be noted that, for a person skilled in the art, several variations and modifications can be made without departing from the concept of the present application, which falls within the scope of protection of the present application. Therefore, the protection scope of the present patent shall be subject to the appended claims.

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High-speed switching motor drive control method

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