阅读说明：本技术 五桥臂逆变器双永磁电机无位置传感器控制装置和方法 (Position-sensorless control device and method for double permanent magnet motors of five-bridge-arm inverter ) 是由 耿强 张甍 夏长亮 陈炜 王慧敏 于 2021-01-27 设计创作，主要内容包括：本发明涉及五桥臂逆变器双永磁电机控制,为能够省去传统五桥臂逆变器双永磁电机系统硬件电路中的机械式光电编码器,从而减小系统体积、降低系统成本,并且提高系统适应性。本发明,五桥臂逆变器双永磁电机无位置传感器控制装置和方法,三相交流电经不可控整流桥转为直流电,再通过五桥臂逆变器驱动两台永磁同步电机,分别在两台电机的零电压矢量控制半周期进行高频方波信号注入,从而通过数学运算获取含有转子位置信息的高频电流响应信号；同时对两台永磁电机均采用矢量控制。本发明主要应用于五桥臂逆变器双永磁电机控制场合。(The invention relates to a five-bridge-arm inverter double-permanent magnet motor control, which can save a mechanical photoelectric encoder in a hardware circuit of a traditional five-bridge-arm inverter double-permanent magnet motor system, thereby reducing the system volume, reducing the system cost and improving the system adaptability. The invention relates to a position sensorless control device and a position sensorless control method for double permanent magnet motors of a five-bridge-arm inverter, wherein three-phase alternating current is converted into direct current through an uncontrollable rectifier bridge, then the five-bridge-arm inverter drives two permanent magnet synchronous motors, and high-frequency square wave signal injection is respectively carried out in a zero-voltage vector control half period of the two motors, so that a high-frequency current response signal containing rotor position information is obtained through mathematical operation; and simultaneously, vector control is adopted for the two permanent magnet motors. The invention is mainly applied to the control occasion of the double permanent magnet motor of the five-bridge arm inverter.)

1. A double-permanent magnet motor position sensorless control device of a five-bridge-arm inverter is characterized by comprising the following structures: three-phase alternating current is converted into direct current through an uncontrollable rectifier bridge, and then two permanent magnet synchronous motors are driven through a five-bridge arm inverter, wherein a bridge arm A, B, C in the five-bridge arm inverter is used for driving a motor PMSM1, a bridge arm A, D, E is used for driving the motor PMSM2, each bridge arm in the five-bridge arm inverter is composed of two IGBT switch tubes connected in series, and each IGBT switch tube is connected with a diode in an anti-parallel mode;

injecting high-frequency square wave signals in the zero-voltage vector control half period of the two motors respectively, so as to obtain high-frequency current response signals containing rotor position information through mathematical operation;

collecting direct-current side voltage and three-phase current of a motor by adopting a Hall current sensor and a Hall voltage sensor, and transmitting collected direct-current side voltage signals and collected current signals to a microprocessor; demodulating the high-frequency current response signal to calculate the position and the rotating speed information of the motor rotor; and the microprocessor respectively calculates the three-phase duty ratios of the two motors according to the rotating speed, the position, the current and the voltage signals, then calculates the five-phase duty ratio acting on the five-bridge-arm inverter through duty ratio correction, and independently controls the two permanent magnet motors.

2. A position sensorless control method for a double-permanent magnet motor of a five-bridge-arm inverter is characterized in that three-phase alternating current is converted into direct current through an uncontrollable rectifier bridge and then drives two permanent magnet synchronous motors through the five-bridge-arm inverter, wherein a bridge arm A, B, C in the five-bridge-arm inverter is used for driving a motor PMSM1, a bridge arm A, D, E is used for driving a motor PMSM2, each bridge arm in the five-bridge-arm inverter is composed of two IGBT switching tubes connected in series, and each IGBT switching tube is connected with a diode in an anti-parallel mode; injecting high-frequency square wave signals in the zero-voltage vector control half period of the two motors respectively, so as to obtain high-frequency current response signals containing rotor position information through mathematical operation; and simultaneously, vector control is adopted for the two permanent magnet motors, namely a sector where a vector is located is calculated through a reference voltage vector, the action time of two zero vectors and non-zero vectors is further calculated, the switching point of each vector is solved, then a triangular carrier signal is compared with the vector switching point of each sector to generate duty ratios required by the control of the two motors, and finally PWM (pulse Width modulation) pulse signals required by the five-leg inverter are generated through duty ratio correction.

3. The position sensorless control method of the double permanent magnet motors of the five-bridge arm inverter is characterized in that a Hall current sensor and a Hall voltage sensor are adopted to collect direct-current side voltage and three-phase current of the motors, and collected direct-current side voltage signals and collected current signals are transmitted to a microprocessor; demodulating the high-frequency current component to calculate the position and rotation speed information of the motor rotor; and the microprocessor respectively calculates the three-phase duty ratios of the two motors according to the rotating speed, the position, the current and the voltage signals, then calculates the five-phase duty ratio acting on the five-bridge-arm inverter through duty ratio correction, and independently controls the two permanent magnet motors.

4. The five-leg inverter double-permanent-magnet motor position sensorless control method as claimed in claim 2, wherein, in particular, subscript i represents permanent-magnet motor 1 and permanent-magnet motor 2, respectively, where i is 1, 2; all variable subscripts d and q respectively represent d-axis and q-axis components under a two-phase rotating d-q coordinate system; all variable subscripts alpha and beta respectively represent alpha axis and beta axis components under a static alpha-beta coordinate system; omega_1refAnd ω_2refReference mechanical rotating speeds of the two motors are respectively set; omega₁And ω₂Are respectively twoThe actual mechanical speed of the motor;andrespectively obtaining mechanical rotating speeds observed by the two motors; i.e. i_{d1_ref}And i_{d2_ref}Respectively setting the current values of the d-axis stators of the two motors; i.e. i_d1And i_d2Actual values of d-axis stator currents of the two motors are respectively obtained; i.e. i_{q1_ref}And i_{q2_ref}Respectively setting the q-axis stator current values of the two motors; i.e. i_q1And i_q2Actual values of q-axis stator currents of the two motors are respectively; l is_diAnd L_qiInductance values for d-axis and q-axis, respectively; v_FOCRepresents an effective voltage vector; v_injRepresents a high frequency injection signal; t is_sRepresents one control cycle; FL-VSI represents a five-leg voltage source inverter;

the voltage equation under the two-phase static coordinate system of the double-permanent magnet synchronous motor is as follows:

in the formula u_αi、u_βiStator voltage components of an alpha axis and a beta axis; r_siIs a stator resistor; i.e. i_αi、i_βiStator current components of an alpha axis and a beta axis; p is a differential factor; psi_αi、ψ_βiStator flux linkage components of the α and β axes;

ψ_αi、ψ_βiexpressed as:

in the formula, L_avgiIs mean inductance, denoted as L_avgi＝(L_di+L_qi)/2；L_difiIs a differential inductance, denoted L_difi＝(L_di-L_qi)/2；ψ_fiIs a rotor permanent magnet flux linkage; theta_eiIs the actual rotor electrical angular position;

because the frequency of the high-frequency injection signal is far higher than the fundamental wave operating frequency, the influence of the resistance voltage drop of the stator, the rotating voltage and the back electromotive force is neglected, and the built-in permanent magnet synchronous motor can be regarded as a pure inductive load;

in the formula u_αhi、u_βhiHigh-frequency voltage components of an alpha axis and a beta axis, respectively; i.e. i_αhi、i_βhiHigh-frequency current response components of an alpha axis and a beta axis, respectively;

defining static alpha-beta shafting, synchronous rotating d-q shafting and estimating rotationRelationship of axis system, orderObserving the rotor electrical angle positions for the two motors;the difference value between the actual value and the observed value of the electrical angle position of the two motor rotors isConversion matrix R (theta) from static alpha-beta shafting to synchronous rotating d-q shafting_ei) Comprises the following steps:

and estimate the rotationD-q shafting conversion matrix from shafting to synchronous rotationComprises the following steps:

the method specifically comprises the following steps of:

1. the collection and calculation of the electric quantity of the two permanent magnet motor systems comprise:

1.1, the speed loop adopts a PI controller, the actual rotating speeds of the two motors are calculated by adopting a position-sensorless algorithm, the rotating speed error required by the control motor is obtained by subtracting the calculated rotating speed signal from the reference rotating speed of each motor, and the difference value between the reference rotating speed and the actual rotating speed is used for generating a q-axis given current i through the PI controller_{qi_ref}D-axis given reference current i_{di_ref}Is 0;

1.2, in a control period, collecting three-phase stator currents of two motors by using a Hall current sensor, and performing Clark conversion on the three-phase stator currents in a microprocessor to convert the three-phase stator currents into stator current components on a two-phase static alpha-beta coordinate system:

in the formula i_a1、i_b1、i_c1、i_a2、i_b2、i_c2Three-phase stator currents of two motors respectively; i.e. i_α1、i_β1、i_α2、i_β2The stator current components of the two motors on a two-phase static alpha-beta coordinate system are respectively;

1.3, in a control period, acquiring rotor position information of two motors by adopting a position sensorless algorithm, and carrying out Park transformation on stator current components of the two motors on a two-phase stationary alpha-beta coordinate system to convert the stator current components into stator current components on a two-phase rotating d-q coordinate system:

and 1.4, collecting the voltage of the direct current side by adopting a Hall voltage sensor, and transmitting the collected voltage signal of the direct current side to a microprocessor.

The method comprises the steps of acquiring rotor position information of two motors by adopting a position-sensorless algorithm, specifically, selecting zero-voltage vector half cycles of the two motors, injecting high-frequency voltage components into stationary beta shafting of the two motors respectively, separating the injection time of effective voltage vectors from high-frequency signals, injecting the high-frequency voltage signals to pass through a motor model to generate response current, and decomposing the response current into fundamental frequency current and high-frequency current containing the rotor position information; obtaining an orthogonal signal containing rotor position information by demodulating the high-frequency current component; and the rotating speed and the position information of the rotor can be obtained through the position observer.

5. The method for controlling the position sensorless of the dual permanent magnet motor of the five-leg inverter according to claim 4, wherein the high-frequency signal separation and demodulation step is,

2.1.1, after injecting high-frequency voltage into the static beta shafting of the motor, the response current of the static alpha-beta shafting of the motor contains a fundamental frequency current component i_αfiAnd i_βfiWith a high-frequency current component i_αhiAnd i_βhi. The high frequency current component has the same frequency as the injection frequency and contains rotor position information, so the sampling current is written as:

in the formula i_αf1、i_βf1Fundamental frequency current components of the first motor on an alpha axis and a beta axis respectively; i.e. i_αf2、i_βf2The fundamental frequency current components of the second motor on an alpha axis and a beta axis respectively; i.e. i_αh1、i_βh1Height of the first motor in alpha and beta axes respectivelyA frequency-responsive current component; i.e. i_αh2、i_βh2High-frequency response current components of the second motor on an alpha axis and a beta axis respectively;

2.1.2, in order to extract a high-frequency current component containing rotor position information, injecting a high-frequency voltage signal into a zero vector half period in a traditional modulation strategy of a five-bridge arm inverter, separating the time of an effective voltage vector from the high-frequency injection signal, enabling the injection frequency to be one half of a PWM frequency, sampling current twice in each PWM period, and enabling the sampling time to be the time before and after the high-frequency signal injection, wherein the sampling frequency is high, a fundamental frequency current is regarded as a constant value in two continuous sampling, and in order to eliminate a voltage error caused by inverter nonlinearity and realize accurate estimation of the position of a rotor of a motor, a positive voltage vector is injected into a first control period, and a negative voltage vector is injected into a second control period, so that the two motors are respectively sampled for four times in the two control periods, and the sampling current is shown as a formula (9):

in the formula (I), the compound is shown in the specification,current sampling values of the motor 1 on an alpha axis and a beta axis at sampling moments 1,2, 3 and 4 respectively;current sampling values of the motor 2 on an alpha axis and a beta axis at sampling moments 1,2, 3 and 4 respectively;the fundamental frequency current components of the motor 1 on the alpha axis and the beta axis at the sampling instants 1,2, 3 and 4 respectively, the fundamental frequency current components of the motor 2 on the alpha and beta axes at the sampling instants 1,2, 3 and 4 respectively, the high-frequency current components on the alpha axis and the beta axis at the sampling time 1,2, 3 and 4 of the motor 1 respectively,sampling high-frequency current components of an alpha axis and a beta axis at the sampling moments 1,2, 3 and 4 of the motor 2;

because the motor runs at low speed and the frequency of the injected high-frequency voltage signal is half of the PWM frequency and is far higher than the fundamental frequency, the fundamental frequency current can be regarded as a fixed value in two continuous samplings; therefore, the high-frequency response current component in the first control period and the high-frequency response current component in the second control period are expressed as:

in the formula (I), the compound is shown in the specification,high-frequency response current components on an alpha axis and a beta axis of a first motor in a first control period respectively;high-frequency response current components of the second motor on an alpha axis and a beta axis in a first control period respectively;for the first motor in the second control cycle, respectivelyA high frequency response current component on the shaft;high-frequency response current components of the second motor on an alpha axis and a beta axis in a second control period respectively;

2.2 high frequency response current envelope extraction

2.2.1, in the signal processing process, obtaining the position information of the rotor by utilizing the high-frequency current component of the static alpha-beta shaft system, and injecting high-frequency square wave voltage into the static alpha-beta shaft system as shown in the formula (11).

2.2.2, obtaining a high-frequency mathematical model of the built-in permanent magnet synchronous motor under the static alpha-beta shafting by the formula (3), and substituting the formula (11) into the formula (3) to obtain:

when the injection signal is in the first control period, equation (12) can be summarized as:

in the formula, delta t is half of a control period; delta I_αhi、ΔI_βhiHigh-frequency current variation amounts of an alpha axis and a beta axis, respectively;

when the injection signal is in the second control period, equation (12) can be summarized as:

performing signal processing on the formula (13) and the formula (14), and multiplying the high-frequency response current generated in the injection period by a sign function, wherein the sign function is expressed as the formula (15):

the high-frequency current response envelope of the motor is obtained at the moment:

in the formula i_cosi、i_siniThe high-frequency current envelope signal containing position information;

according to formula (17), I_NAnd I_MThe parameters in the expression are all known quantities, so that the direct current offset I can be obtained by calculation_NAnd coefficient I_MTherefore, simplified formula (16) can be simplified to obtain:

in the formula I_cosi、I_siniThe orthogonal signals containing the position information of the motor rotor are obtained;

2.3 principle and design of position observer

2.3.1 estimating rotor position, high frequency response current i in stationary alpha-beta shafting by using orthogonal phase locked loop_αhi、i_βhiObtaining orthogonal signal I containing rotor position information after processing_cosi、I_siniThe rotor position information can be extracted from the signal through an orthogonal phase-locked loop control link;

the quadrature phase-locked loop is composed of three parts, namely a Phase Detector (PD), a loop oscillator (LPF) and a Voltage Controlled Oscillator (VCO). The effect of the phase detector is to compare the estimated phase with the actual phaseBit comparison, i.e. two input signals I_cosi、I_siniAnd two output signalsThe processing and comparison is performed to obtain a position error signal:

wherein ε is the position error signal;

formula (18) may be substituted for formula (19):

when the estimated rotor position signal limit of the machine approaches the actual rotor position, i.e. whenWhen there is, formula (21) holds

The closed loop transfer function of the quadrature phase locked loop is as follows:

wherein s denotes a complex frequency in the complex frequency domain; k_pThe coefficient of a proportional regulator of a proportional integral regulator inside the quadrature phase-locked loop; k_iThe integral regulator coefficient of a proportional-integral regulator inside the quadrature phase-locked loop;

the steady state error transfer function of the quadrature phase locked loop, which can be derived from equation (22), is:

when the motor is in steady state operation, the input theta_eiAs a ramp function, the steady state error of the quadrature phase locked loop at this time is:

the loop filter adopts the adjustment effect of a PI (proportional-integral) regulator to filter out high-frequency components and noise in epsilon, a voltage-controlled oscillator is controlled by control voltage, the frequency of the voltage-controlled oscillator is close to the frequency of an input signal until the frequency difference is eliminated, then the frequency is locked, and the influence generated by higher harmonics can be eliminated by a phase-locked loop.

Technical Field

The invention relates to a high-frequency square wave injection control method for a double-permanent magnet motor of a five-bridge-arm inverter, and belongs to the field of multi-motor control. In particular to a low-speed position-sensorless control method applied to a five-bridge-arm inverter double-permanent magnet synchronous motor system.

Background

With the continuous development of industrial automation, two motors are often required to be cooperatively controlled in industries such as electric vehicles, wind power generation, heavy load lifting and the like, so as to solve the problems of low driving reliability, poor control performance, complex mechanical transmission mechanism of a system, high power requirement of a single motor and the like of the traditional single motor. In conventional industrial applications, a permanent magnet synchronous motor is generally driven by a voltage-type inverter composed of semiconductor switching devices IGBT. Researches show that approximately half of inverter faults are caused by damage of a switching tube, when one phase of bridge arm of a traditional six-bridge arm inverter fails, a five-bridge arm voltage source inverter serves as a good fault-tolerant control scheme and can independently control two permanent magnet synchronous motors, and in recent years, attention of numerous scholars is paid, and the method can save two power switching devices, so that the system size can be reduced, and the system cost can be reduced. However, the motor needs real-time feedback of position information and rotation speed information in the operation process to form closed-loop control, and is mostly realized by adopting a mechanical encoder at present, but the use of the mechanical encoder can increase the system volume, improve the system cost and reduce the system reliability. In addition, in some occasions with complex working conditions, such as aerospace, mining industry, wind power generation and the like, higher requirements are provided for the volume and the reliability of the system, so that the dual-motor position-sensor-free control technology has important research significance

The control target of the position sensorless control system of the double permanent magnet motors of the five-bridge arm inverter is as follows: under the condition of keeping the control independence of the two three-phase permanent magnet synchronous motors, the structure is simplified, the system reliability is improved, the dynamic performance is improved, the control precision is improved, and the like. The traditional mechanical position sensor increases the installation and maintenance difficulty, and is easily influenced by conditions such as temperature and humidity, so that the volume of the permanent magnet motor system is increased, and the reliability is reduced.

Disclosure of Invention

In order to fill the gap of the prior art, the invention aims to provide a method which can save a mechanical photoelectric encoder in a hardware circuit of a traditional five-bridge-arm inverter dual-permanent magnet motor system, thereby reducing the system volume, reducing the system cost and improving the system adaptability. The invention is realized by the following technical scheme.

A three-phase alternating current is converted into direct current through an uncontrollable rectifier bridge, and then two permanent magnet synchronous motors are driven through a five-bridge-arm inverter, a bridge arm A, B, C in the five-bridge-arm inverter is used for driving a motor PMSM1, a bridge arm A, D, E is used for driving the motor PMSM2, each bridge arm in the five-bridge arm is composed of two IGBT switching tubes connected in series, and each IGBT switching tube is connected with a diode in an anti-parallel mode; injecting high-frequency square wave signals in the zero-voltage vector control half period of the two motors respectively, so as to obtain high-frequency current response signals containing rotor position information through mathematical operation; and simultaneously, vector control is adopted for the two permanent magnet motors, namely a sector where a vector is located is calculated through a reference voltage vector, the action time of two zero vectors and non-zero vectors is further calculated, the switching point of each vector is solved, then a triangular carrier signal is compared with the vector switching point of each sector to generate duty ratios required by the control of the two motors, and finally PWM (pulse Width modulation) pulse signals required by the five-leg inverter are generated through duty ratio correction.

Collecting direct-current side voltage and three-phase current of a motor by adopting a Hall current sensor and a Hall voltage sensor, and transmitting collected direct-current side voltage signals and collected current signals to a microprocessor; demodulating the high-frequency current component to calculate the position and rotation speed information of the motor rotor; and the microprocessor respectively calculates the three-phase duty ratios of the two motors according to the rotating speed, the position, the current and the voltage signals, then calculates the five-phase duty ratio acting on the five-bridge-arm inverter through duty ratio correction, and independently controls the two permanent magnet motors.

Specifically, subscript i represents permanent magnet machine 1 and permanent magnet machine 2, i ═ 1, 2; all variable subscripts d and q respectively represent d-axis and q-axis components under a two-phase rotating d-q coordinate system; all ofThe variable subscripts alpha and beta respectively represent alpha axis and beta axis components under a static alpha-beta coordinate system; omega_1refAnd ω_2refReference mechanical rotating speeds of the two motors are respectively set; omega₁And ω₂The actual mechanical rotating speeds of the two motors are respectively;andrespectively obtaining mechanical rotating speeds observed by the two motors; i.e. i_{d1_ref}And i_{d2_ref}Respectively setting the current values of the d-axis stators of the two motors; i.e. i_d1And i_d2Actual values of d-axis stator currents of the two motors are respectively obtained; i.e. i_{q1_ref}And i_{q2_ref}Respectively setting the q-axis stator current values of the two motors; i.e. i_q1And i_q2Actual values of q-axis stator currents of the two motors are respectively; l is_diAnd L_qiInductance values for d-axis and q-axis, respectively; v_FOCRepresents an effective voltage vector; v_injRepresents a high frequency injection signal; t is_sRepresents one control cycle; FL-VSI represents a five-leg voltage source inverter;

the voltage equation under the two-phase static coordinate system of the double-permanent magnet synchronous motor is

In the formula u_αi、u_βiStator voltage components of an alpha axis and a beta axis; r_siIs a stator resistor; i.e. i_αi、i_βiStator current components of an alpha axis and a beta axis; p is a differential factor; psi_αi、ψ_βiStator flux linkage components of the α and β axes;

ψ_αi、ψ_βiexpressed as:

in the formula, L_avgiIs mean value electricityFeeling, expressed as L_avgi＝(L_di+L_qi)/2；L_difiIs a differential inductance, denoted L_difi＝(L_di-L_qi)/2；ψ_fiIs a rotor permanent magnet flux linkage; theta_eiIs the actual rotor electrical angular position;

because the frequency of the high-frequency injection signal is far higher than the fundamental wave operating frequency, the influence of the resistance voltage drop of the stator, the rotating voltage and the back electromotive force is neglected, and the built-in permanent magnet synchronous motor can be regarded as a pure inductive load;

in the formula u_αhi、u_βhiHigh-frequency voltage components of an alpha axis and a beta axis, respectively; i.e. i_αhi、i_βhiHigh-frequency current response components of an alpha axis and a beta axis, respectively;

defining static alpha-beta shafting, synchronous rotating d-q shafting and estimating rotationRelationship of axis system, orderObserving the rotor electrical angle positions for the two motors;the difference value between the actual value and the observed value of the electrical angle position of the two motor rotors isConversion matrix R (theta) from static alpha-beta shafting to synchronous rotating d-q shafting_ei) Comprises the following steps:

and estimate the rotationD-q shafting conversion matrix from shafting to synchronous rotationComprises the following steps:

the method specifically comprises the following steps of:

1. the collection and calculation of the electric quantity of the two permanent magnet motor systems comprise:

1.1, the speed loop adopts a PI controller, the actual rotating speeds of the two motors are calculated by adopting a position-sensorless algorithm, the rotating speed error required by the control motor is obtained by subtracting the calculated rotating speed signal from the reference rotating speed of each motor, and the difference value between the reference rotating speed and the actual rotating speed is used for generating a q-axis given current i through the PI controller_{qi_ref}D-axis given reference current i_{di_ref}Is 0;

1.2, in a control period, collecting three-phase stator currents of two motors by using a Hall current sensor, and performing Clark conversion on the three-phase stator currents in a microprocessor to convert the three-phase stator currents into stator current components on a two-phase static alpha-beta coordinate system:

in the formula i_a1、i_b1、i_c1、i_a2、i_b2、i_c2Three-phase stator currents of two motors respectively; i.e. i_α1、i_β1、i_α2、i_β2The stator current components of the two motors on a two-phase static alpha-beta coordinate system are respectively;

1.3, in a control period, acquiring rotor position information of two motors by adopting a position sensorless algorithm, and carrying out Park transformation on stator current components of the two motors on a two-phase stationary alpha-beta coordinate system to convert the stator current components into stator current components on a two-phase rotating d-q coordinate system:

and 1.4, collecting the voltage of the direct current side by adopting a Hall voltage sensor, and transmitting the collected voltage signal of the direct current side to a microprocessor.

The method comprises the steps of acquiring rotor position information of two motors by adopting a position-sensorless algorithm, specifically, selecting zero-voltage vector half cycles of the two motors, injecting high-frequency voltage components into stationary beta shafting of the two motors respectively, separating the injection time of effective voltage vectors from high-frequency signals, injecting the high-frequency voltage signals to pass through a motor model to generate response current, and decomposing the response current into fundamental frequency current and high-frequency current containing the rotor position information; obtaining an orthogonal signal containing rotor position information by demodulating the high-frequency current component; and the rotating speed and the position information of the rotor can be obtained through the position observer.

The high-frequency signal separating and demodulating step is,

2.1.1, after injecting high-frequency voltage into the static beta shafting of the motor, the response current of the static alpha-beta shafting of the motor contains a fundamental frequency current component i_αfiAnd i_βfiWith a high-frequency current component i_αhiAnd i_βhi. The high frequency current component has the same frequency as the injection frequency and contains rotor position information, so the sampling current is written as:

in the formula i_αf1、i_βf1Fundamental frequency current components of the first motor on an alpha axis and a beta axis respectively; i.e. i_αf2、i_βf2The fundamental frequency current components of the second motor on an alpha axis and a beta axis respectively; i.e. i_αh1、i_βh1High-frequency response current components of the first motor on an alpha axis and a beta axis respectively; i.e. i_αh2、i_βh2High-frequency response current components of the second motor on an alpha axis and a beta axis respectively;

2.1.2, in order to extract a high-frequency current component containing rotor position information, injecting a high-frequency voltage signal in a zero vector half period in a traditional modulation strategy of a five-bridge arm inverter, separating the time of an effective voltage vector from the high-frequency injection signal, enabling the injection frequency to be one half of a PWM frequency, sampling current twice in each PWM period, and enabling the sampling time to be the time before and after the high-frequency signal injection, wherein the sampling frequency is high, a fundamental frequency current is regarded as a constant value in two continuous sampling, and in order to eliminate a voltage error caused by inverter nonlinearity and realize accurate estimation of the position of a rotor of a motor, a positive voltage vector is injected in a first control period, a negative voltage vector is injected in a second control period, so that in the two motors are respectively sampled for four times in the two control periods, and the sampling current is shown as a formula (9).

In the formula (I), the compound is shown in the specification,current sampling values of the motor 1 on an alpha axis and a beta axis at sampling moments 1,2, 3 and 4 respectively;current sampling values of the motor 2 on an alpha axis and a beta axis at sampling moments 1,2, 3 and 4 respectively;the fundamental frequency current components of the motor 1 on the alpha axis and the beta axis at the sampling instants 1,2, 3 and 4 respectively,the fundamental frequency current components of the motor 2 on the alpha and beta axes at the sampling instants 1,2, 3 and 4 respectively, the high-frequency current components on the alpha axis and the beta axis at the sampling time 1,2, 3 and 4 of the motor 1 respectively, sampling high-frequency current components of an alpha axis and a beta axis at the sampling moments 1,2, 3 and 4 of the motor 2;

because the motor runs at low speed and the frequency of the injected high-frequency voltage signal is half of the PWM frequency and is far higher than the fundamental frequency, the fundamental frequency current can be regarded as a fixed value in two continuous samplings; therefore, the high-frequency response current component in the first control period and the high-frequency response current component in the second control period are expressed as:

in the formula (I), the compound is shown in the specification,high-frequency response current components on an alpha axis and a beta axis of a first motor in a first control period respectively;high-frequency response current components of the second motor on an alpha axis and a beta axis in a first control period respectively;high-frequency response current components of the first motor on an alpha axis and a beta axis in a second control period respectively;high-frequency response current components of the second motor on an alpha axis and a beta axis in a second control period respectively;

2.2 high frequency response current envelope extraction

2.2.1, in the signal processing process, obtaining the position information of the rotor by utilizing the high-frequency current component of the static alpha-beta shaft system, and injecting high-frequency square wave voltage into the static alpha-beta shaft system as shown in the formula (11).

2.2.2, obtaining a high-frequency mathematical model of the built-in permanent magnet synchronous motor under the static alpha-beta shafting by the formula (3), and substituting the formula (11) into the formula (3) to obtain:

when the injection signal is in the first control period, equation (12) can be summarized as:

in the formula, delta t is half of a control period; delta I_αhi、ΔI_βhiHigh-frequency current variation amounts of an alpha axis and a beta axis, respectively;

when the injection signal is in the second control period, equation (12) can be summarized as:

performing signal processing on the formula (13) and the formula (14), and multiplying the high-frequency response current generated in the injection period by a sign function, wherein the sign function is expressed as the formula (15):

the high-frequency current response envelope of the motor is obtained at the moment:

in the formula i_cosi、i_siniThe high-frequency current envelope signal containing position information;

according to formula (17), I_NAnd I_MThe parameters in the expression are all known quantities, so that the direct current offset I can be obtained by calculation_NAnd coefficient I_MTherefore, simplified formula (16) can be simplified to obtain:

in the formula I_cosi、I_siniThe orthogonal signals containing the position information of the motor rotor are obtained;

2.3 principle and design of position observer

2.3.1 estimating rotor position, high frequency response current i in stationary alpha-beta shafting by using orthogonal phase locked loop_αhi、i_βhiObtaining orthogonal signal I containing rotor position information after processing_cosi、I_siniThe rotor position information can be extracted from the signal through an orthogonal phase-locked loop control link;

the quadrature phase-locked loop is composed of three parts, namely a Phase Detector (PD), a loop oscillator (LPF) and a Voltage Controlled Oscillator (VCO). The effect of the phase detector is to compare the estimated phase with the actual phase, i.e. to compare the two input signals I_cosi、I_siniAnd two output signalsThe processing is performed and compared to obtain a position error signal.

Wherein ε is the position error signal;

formula (18) may be substituted for formula (19):

when the estimated rotor position signal limit of the machine approaches the actual rotor position, i.e. whenWhen there is, formula (21) holds

The closed loop transfer function of the quadrature phase locked loop is as follows:

wherein s denotes a complex frequency in the complex frequency domain; k_pThe coefficient of a proportional regulator of a proportional integral regulator inside the quadrature phase-locked loop; k_iThe integral regulator coefficient of a proportional-integral regulator inside the quadrature phase-locked loop;

the steady state error transfer function of the quadrature phase locked loop, which can be derived from equation (22), is:

when the motor is in steady state operation, the input theta_eiAs a ramp function, the steady state error of the quadrature phase locked loop at this time is:

the loop filter adopts the adjustment effect of a PI (proportional-integral) regulator to filter out high-frequency components and noise in epsilon, a voltage-controlled oscillator is controlled by control voltage, the frequency of the voltage-controlled oscillator is close to the frequency of an input signal until the frequency difference is eliminated, then the frequency is locked, and the influence generated by higher harmonics can be eliminated by a phase-locked loop.

A position-sensorless control device of a five-bridge-arm inverter double-permanent magnet motor system is structurally characterized in that:

three-phase alternating current is converted into direct current through an uncontrollable rectifier bridge, and then two permanent magnet synchronous motors are driven through a five-bridge arm inverter, wherein a bridge arm A, B, C in the five-bridge arm inverter is used for driving a motor PMSM1, a bridge arm A, D, E is used for driving the motor PMSM2, each bridge arm in the five-bridge arm inverter is composed of two IGBT switch tubes connected in series, and each IGBT switch tube is connected with a diode in an anti-parallel mode;

injecting high-frequency square wave signals in the zero-voltage vector control half period of the two motors respectively, so as to obtain high-frequency current response signals containing rotor position information through mathematical operation;

collecting direct-current side voltage and three-phase current of a motor by adopting a Hall current sensor and a Hall voltage sensor, and transmitting collected direct-current side voltage signals and collected current signals to a microprocessor; demodulating the high-frequency current response signal to calculate the position and the rotating speed information of the motor rotor; and the microprocessor respectively calculates the three-phase duty ratios of the two motors according to the rotating speed, the position, the current and the voltage signals, then calculates the five-phase duty ratio acting on the five-bridge-arm inverter through duty ratio correction, and independently controls the two permanent magnet motors.

The invention has the characteristics and beneficial effects that:

the invention discloses a control method applied to a five-bridge-arm inverter double-permanent-magnet motor system, provides a low-speed position-sensorless control method, saves the use of a mechanical photoelectric encoder and a filter in the position observation process, saves the system space, improves the dynamic performance of the system, improves the reliability of the system and reduces the cost of the system, and is a good fault-tolerant control scheme.

Description of the drawings:

fig. 1 is a circuit topology structure diagram of a five-bridge-arm inverter dual-permanent magnet motor system.

Fig. 2 shows a space voltage vector diagram of two permanent magnet motors.

Fig. 3 is a control structure diagram of a position sensorless system of a double-permanent magnet motor of a five-bridge-arm inverter.

FIG. 4 is a schematic diagram of coordinate system axis definition.

Fig. 5 is a high frequency square wave voltage signal injection timing diagram.

FIG. 6 is a diagram of a high frequency square wave voltage signal injection.

FIG. 7 is a schematic block diagram of a position observer.

FIG. 8 is a Bode diagram of a position observer.

FIG. 9 hardware experiment system platform.

FIG. 10 is a waveform of a speed step experiment at idle time.

FIG. 11 is a no load speed step current response plot.

Fig. 12 is a waveform diagram of a load step experiment.

Fig. 13 is a current response diagram at a load step.

FIG. 14 is a waveform diagram of the positive and negative rotation test with load.

FIG. 15 is a current response diagram for positive and negative load rotation.

Detailed Description

The invention constructs a high-frequency square wave voltage injection position-free sensor control method based on a static shafting, which is characterized in that on the basis of a five-bridge arm traditional modulation strategy, high-frequency square wave signal injection is respectively carried out in a zero voltage vector control half period of two motors, the zero vector half period is utilized, the effective voltage vector is separated from the high-frequency signal injection moment, the use of a filter in the high-frequency signal extraction process is omitted, and due to the fact that a common bridge arm is arranged, the duty ratio generated by the high-frequency signal injection half period of one motor and the duty ratio generated by the vector control half period of the other motor are corrected, so that the duty ratio required by the system is generated. The control method obtains the rotating speed and position information of the rotors of the two motors according to the high-frequency response currents of the two motors.

A five-bridge-arm inverter double-permanent magnet motor position-sensorless control method is used for controlling two permanent magnet synchronous motors based on a traditional modulation method of the five-bridge-arm inverter, and the two inverters share an A-phase bridge arm. Injecting high-frequency square wave signals in the zero-voltage vector control half period of the two motors respectively, so as to obtain high-frequency current response signals containing rotor position information through mathematical operation; meanwhile, the two permanent magnet motors are controlled by the traditional vector, namely, the sector of the vector is calculated by a reference voltage vector, the action time of two zero vectors and non-zero vectors is further calculated, the switching point of each vector is solved, then a triangular carrier signal with a certain frequency is compared with the switching point of each sector vector to generate the duty ratio required by the control of the two motors, and finally a PWM (pulse Width modulation) pulse signal required by the five-bridge-arm inverter is generated by duty ratio correction.

In the invention, subscripts i of all variables respectively represent a permanent magnet motor 1 and a permanent magnet motor 2, wherein i is 1 and 2; all variable subscripts d and q respectively represent d-axis and q-axis components under a two-phase rotating d-q coordinate system; all variable subscripts alpha and beta respectively represent alpha axis and beta axis components under a static alpha-beta coordinate system; omega_1refAnd ω_2refReference mechanical rotating speeds of the two motors are respectively set; omega₁And ω₂The actual mechanical rotating speeds of the two motors are respectively;andrespectively obtaining mechanical rotating speeds observed by the two motors; i.e. i_{d1_ref}And i_{d2_ref}Respectively setting the current values of the d-axis stators of the two motors; i.e. i_d1And i_d2Actual values of d-axis stator currents of the two motors are respectively obtained; i.e. i_{q1_ref}And i_{q2_ref}Respectively setting the q-axis stator current values of the two motors; i.e. i_q1And i_q2Actual values of q-axis stator currents of the two motors are respectively; l is_diAnd L_qiInductance values for d-axis and q-axis, respectively; v_FOCRepresents an effective voltage vector; v_injRepresents a high frequency injection signal; t is_sRepresents one control cycle; FL-VSI representationA five bridge arm voltage source inverter; PMSM1 and PMSM2 are two permanent magnet synchronous machines.

A position-sensorless control device of a five-bridge-arm inverter double-permanent magnet motor system is structurally characterized in that:

three-phase alternating current is converted into direct current through an uncontrollable rectifier bridge, and then two permanent magnet synchronous motors are driven through a five-bridge arm inverter, wherein a bridge arm A, B, C in the five-bridge arm inverter is used for driving a motor PMSM1, a bridge arm A, D, E is used for driving the motor PMSM2, each bridge arm in the five-bridge arm inverter is composed of two IGBT switch tubes connected in series, and each IGBT switch tube is connected with a diode in an anti-parallel mode;

and obtaining the position and the rotating speed information of the rotors of the two motors by a high-frequency square wave voltage injection method.

Collecting direct-current side voltage and three-phase current of a motor by adopting a Hall current sensor and a Hall voltage sensor, and transmitting collected direct-current side voltage signals and collected current signals to a microprocessor; demodulating the high-frequency current component to calculate the position and rotation speed information of the motor rotor; the microprocessor respectively calculates the three-phase duty ratios of the two motors according to the rotating speed, the position, the current and the voltage signals, then calculates the five-phase duty ratio acting on the five-bridge-arm inverter through duty ratio correction, and independently controls the two permanent magnet motors; specifically, when the a-phase duty ratio generated by the effective voltage vector control in the first half period of the PMSM1 is not equal to the a-phase duty ratio generated by the high-frequency injection signal in the first half period of the PMSM2, the action requirements of the common bridge arm are not consistent; similarly, this problem is also caused when the a-phase duty ratio generated by the high-frequency injection signal in the latter half period of PMSM1 is not equal to the a-phase duty ratio generated by the effective voltage vector control in the latter half period of PMSM 2. In order to realize the independent control of the two motors, the duty ratio generated by the high-frequency injection half period of one motor and the duty ratio generated by the vector control half period of the other motor need to be corrected, so that the independent control of the two motors is realized under the condition that the duty ratios of the common bridge arms are consistent.

The present invention will be described in further detail with reference to the accompanying drawings and specific examples.

The circuit topology structure of the five-bridge-arm inverter double-permanent-magnet motor system comprises a three-phase power grid, an uncontrollable rectifier bridge, a five-bridge-arm inverter, two permanent-magnet synchronous motors (PMSM) PMSM1 and PMSM2, wherein a bridge arm A, B, C is used for driving a motor PMSM1, and a bridge arm A, D, E is used for driving a motor PMSM 2. In the present invention, the subscripts i (i ═ 1,2) of all variables represent PMSM1 and PMSM2, respectively. It can be seen that compared to the conventional six-leg inverter, the five-leg structure reduces the number of power switching devices, and two inverters share the a-phase leg.

In this embodiment, let S_x(x ═ A, B, C, D, E) is a switching function, and when the switch tube on the corresponding bridge arm is switched on and the lower switch tube is switched off, S is made_x1 is ═ 1; on the contrary, when the upper switch tube is turned off and the lower switch tube is turned on, the S is enabled_x0. For each individual motor, the voltage vector controlling each motor comprises 6 effective vectors u_in(n-1, …,6) and 2 zero vectors u_im(m is 0, 7). The space voltage vector of each motor is divided into six sectors of I, II, III, IV, V and VI; u. of_i1[1,0,0]，u_i2[1,1,0]，u_i3[0,1,0]，u_i4[0,1,1]，u_i5[0,0,1]，u_i6[1,0,1]6 effective voltage vectors; u. of_i0[0,0,0]，u_i7[1,1,1]Is 2 zero voltage vectors.

In the aspect of a control structure of the invention, a Proportional Integral (PI) controller is adopted for a speed loop, subscripts i of all variables respectively represent a permanent magnet synchronous motor 1 and a permanent magnet synchronous motor 2, and i is 1 and 2; all variable subscripts d and q respectively represent d-axis and q-axis components under a two-phase rotating d-q coordinate system; all variable subscripts alpha and beta respectively represent alpha axis and beta axis components under a static alpha-beta coordinate system; omega_1refAnd ω_2refReference rotating speeds of the two motors are respectively set; omega₁And ω₂The actual mechanical rotating speeds of the two motors are respectively;andare respectively two motorsObserving the obtained mechanical rotating speed; i.e. i_{d1_ref}And i_{d2_ref}Respectively setting the current values of the d-axis stators of the two motors; i.e. i_d1And i_d2Actual values of d-axis stator currents of the two motors are respectively obtained; i.e. i_{q1_ref}And i_{q2_ref}Respectively setting the q-axis stator current values of the two motors; i.e. i_q1And i_q2Actual values of q-axis stator currents of the two motors are respectively; l is_diAnd L_qiInductance values for d-axis and q-axis, respectively; v_FOCRepresents an effective voltage vector; v_injRepresents a high frequency injection signal; t is_sRepresents one control cycle; FL-VSI represents a five-leg voltage source inverter; PMSM1 and PMSM2 are two permanent magnet synchronous machines.

The voltage equation under the two-phase static coordinate system of the double-permanent magnet synchronous motor is

In the formula u_αi、u_βiStator voltage components of an alpha axis and a beta axis; r_siIs a stator resistor; i.e. i_αi、i_βiStator current components of an alpha axis and a beta axis; p is a differential factor; psi_αi、ψ_βiThe stator flux linkage components are the alpha and beta axes.

ψ_αi、ψ_βiIs shown as

In the formula, L_avgiIs mean inductance, denoted as L_avgi＝(L_di+L_qi)/2；L_difiIs a differential inductance, denoted L_difi＝(L_di-L_qi)/2；ψ_fiIs a rotor permanent magnet flux linkage; theta_eiIs the actual rotor electrical angular position.

Because the frequency of the high-frequency injection signal is far higher than the fundamental wave operating frequency, the influence of the resistance voltage drop of the stator, the rotating voltage and the back electromotive force is ignored, and the built-in permanent magnet synchronous motor can be regarded as a pure inductive load

In the formula u_αhi、u_βhiHigh-frequency voltage components of an alpha axis and a beta axis, respectively; i.e. i_αhi、i_βhiHigh frequency current response components of the alpha and beta axes, respectively.

Defining static alpha-beta shafting, synchronous rotating d-q shafting and estimating rotationRelationship of axis system, orderObserving the rotor electrical angle positions for the two motors;the difference value between the actual value and the observed value of the electrical angle position of the two motor rotors isConversion matrix R (theta) from static alpha-beta shafting to synchronous rotating d-q shafting_ei) Comprises the following steps:

estimating rotationD-q shafting conversion matrix from shafting to synchronous rotationComprises the following steps:

the control without the position sensor further comprises the following specific steps:

1. the collection and calculation of the electric quantity of the two permanent magnet motor systems comprise:

1.1, the speed loop adopts a PI controller, the actual rotating speeds of the two motors are calculated by adopting a position-sensorless algorithm, the rotating speed error required by the control motor is obtained by subtracting the calculated rotating speed signal from the reference rotating speed of each motor, and the difference value between the reference rotating speed and the actual rotating speed is used for generating a q-axis given current i through the PI controller_{qi_ref}D-axis given reference current i_{di_ref}Is 0.

1.2, in a control period, collecting three-phase stator currents of two motors by using a Hall current sensor, and performing Clark conversion on the three-phase stator currents in a microprocessor to convert the three-phase stator currents into stator current components on a two-phase static alpha-beta coordinate system:

in the formula i_a1、i_b1、i_c1Three-phase stator currents, i, of the first motor, respectively_a2、i_b2、i_c2The three-phase stator currents of the second motor are respectively; i.e. i_α1、i_β1The stator current components, i, of the first motor in a stationary two-phase alpha-beta coordinate system_α2、i_β2Respectively, the stator current components of the second motor in the two-phase stationary alpha-beta coordinate system.

1.3, in a control period, rotor position information of the two motors is obtained by adopting a position sensorless algorithm, and stator current components of the two motors on a two-phase static alpha-beta coordinate system are subjected to Park conversion to be converted into stator current components on a two-phase rotating d-q coordinate system.

And 1.4, collecting the voltage of the direct current side by adopting a Hall voltage sensor, and transmitting the collected voltage signal of the direct current side to a microprocessor.

2. Position sensorless control

In a traditional modulation strategy of a five-bridge arm inverter, a control cycle is divided into two half cycles with equal length, in the former half cycle, a first motor carries out effective voltage vector modulation, and a second motor carries out zero voltage vector modulation; in the latter half period, the second motor performs effective voltage vector modulation, and the first motor performs zero voltage vector modulation. The method is based on a traditional modulation strategy of a five-bridge-arm inverter, zero-voltage vector half cycles of two motors are selected, high-frequency voltage components are respectively injected into static beta shafting of the two motors, the injection time of effective voltage vectors and high-frequency signals is separated, the high-frequency voltage signals are injected to pass through a motor model to generate response current, and the response current is decomposed into fundamental frequency current and high-frequency current containing rotor position information; obtaining an orthogonal signal containing rotor position information by demodulating the high-frequency current component; and the rotating speed and the position information of the rotor can be obtained through the position observer.

2.1 high-frequency Signal separation and demodulation

2.1.1, after injecting high-frequency voltage into the static beta shafting of the motor, the response current of the static alpha-beta shafting of the motor contains a fundamental frequency current component i_αfiAnd i_βfiWith a high-frequency current component i_αhiAnd i_βhi. The high-frequency current component has the same frequency as the injection frequency and contains rotor position information, so the sampling current can be written as:

in the formula i_αf1、i_βf1Fundamental frequency current components of the first motor on an alpha axis and a beta axis respectively; i.e. i_αf2、i_βf2The fundamental frequency current components of the second motor on an alpha axis and a beta axis respectively; i.e. i_αh1、i_βh1High-frequency response current components of the first motor on an alpha axis and a beta axis respectively; i.e. i_αh2、i_βh2The high-frequency response current components of the second motor on the alpha axis and the beta axis respectively.

2.1.2, in order to extract a high-frequency current component containing rotor position information, in the invention, high-frequency voltage signal injection is carried out in a zero vector half period in a traditional modulation strategy of the five-bridge arm inverter, and the time of an effective voltage vector is separated from the time of a high-frequency injection signal. Let the injection frequency be one-half of the PWM frequency. The current sampling is carried out twice in each PWM period, the sampling time is before and after the high-frequency signal is injected, and the fundamental frequency current can be regarded as a fixed value in two continuous sampling due to the fact that the sampling frequency is high. In order to eliminate voltage error caused by inverter nonlinearity and realize accurate estimation of the position of the motor rotor, a positive voltage vector is injected in a first control period, and a negative voltage vector is injected in a second control period. Therefore, in two control periods, the two motors need to perform four times of sampling respectively, and the sampling current is shown as the formula (9).

In the formula (I), the compound is shown in the specification,current sampling values of the motor 1 on an alpha axis and a beta axis at sampling moments 1,2, 3 and 4 respectively;current sampling values of the motor 2 on an alpha axis and a beta axis at sampling moments 1,2, 3 and 4 respectively;the fundamental frequency current components of the motor 1 on the alpha axis and the beta axis at the sampling instants 1,2, 3 and 4 respectively,the fundamental frequency current components of the motor 2 on the alpha and beta axes at the sampling instants 1,2, 3 and 4 respectively, the high-frequency current components on the alpha axis and the beta axis at the sampling time 1,2, 3 and 4 of the motor 1 respectively, the high frequency current components of the alpha and beta axes are sampled at times 1,2, 3, 4 for the motor 2.

Because the motor runs at low speed and the frequency of the injected high-frequency voltage signal is half of the PWM frequency and is far higher than the fundamental frequency, the fundamental frequency current can be regarded as a fixed value in two continuous samplings; therefore, the high-frequency response current component in the first control period and the high-frequency response current component in the second control period are expressed as:

in the formula (I), the compound is shown in the specification,high-frequency response current components on an alpha axis and a beta axis of a first motor in a first control period respectively;high-frequency response current components of the second motor on an alpha axis and a beta axis in a first control period respectively;high-frequency response current components of the first motor on an alpha axis and a beta axis in a second control period respectively;the high frequency response current components on the alpha axis and the beta axis of the second motor in the second control period are respectively.

2.2 high frequency response current envelope extraction

2.2.1, in the signal processing process, obtaining the position information of the rotor by utilizing the high-frequency current component of the static alpha-beta shaft system, and injecting high-frequency square wave voltage into the static alpha-beta shaft system as shown in the formula (11).

2.2.2, the high-frequency mathematical model of the built-in permanent magnet synchronous motor under the static alpha-beta shafting can be obtained by the formula (3), and the formula (11) is substituted for the formula (3) to obtain:

when the injection signal is in the first control period, equation (12) can be summarized as:

in the formula, delta t is half of a control period; delta I_αhi、ΔI_βhiThe high-frequency current variation amounts of the α axis and the β axis, respectively.

When the injection signal is in the second control period, equation (12) can be summarized as:

the equations (13) and (14) are signal-processed, and the high-frequency response current generated in the injection period is multiplied by a sign function, which is expressed as equation (15).

The high-frequency current response envelope of the motor can be obtained as follows:

in the formula i_cosi、i_siniIs a high frequency current envelope signal containing position information.

According to formula (17), I_NAnd I_MThe parameters in the expression are all known quantities, so that the direct current offset I can be obtained by calculation_NAnd coefficient I_MTherefore, simplified formula (16) can be simplified to obtain:

in the formula I_cosi、I_siniIs a quadrature signal containing information on the position of the rotor of the motor.

2.3 principle and design of position observer

2.3.1, the position observer is a key component for realizing the estimation of the rotor position in the high-frequency square wave injection method, and the invention adopts an orthogonal phase-locked loop to estimate the rotor position. High-frequency response current i in static alpha-beta shafting_αhi、i_βhiObtaining orthogonal signal I containing rotor position information after processing_cosi、I_siniAnd the rotor position information can be extracted from the signal through an orthogonal phase-locked loop control link.

The quadrature phase-locked loop is composed of three parts, namely a Phase Detector (PD), a loop oscillator (LPF) and a Voltage Controlled Oscillator (VCO). The effect of the phase detector is to compare the estimated phase with the actual phase, i.e. to compare the two input signals I_cosi、I_siniAnd two output signalsProcessing and comparing to obtain position errorA difference signal.

Where ε is the position error signal.

Formula (18) may be substituted for formula (19):

when the estimated rotor position signal limit of the machine approaches the actual rotor position, i.e. whenWhen there is, formula (21) holds

The closed loop transfer function of the quadrature phase locked loop is as follows:

wherein s denotes a complex frequency in the complex frequency domain; k_pThe coefficient of a proportional regulator of a proportional integral regulator inside the quadrature phase-locked loop; k_iThe integral regulator coefficient of a proportional integral regulator inside the quadrature phase-locked loop.

The steady state error transfer function of the quadrature phase locked loop, which can be derived from equation (22), is:

when the motor is in steady state operation, the input theta_eiAs a ramp function, the steady state error of the quadrature phase locked loop at this time is:

the loop filter adopts the adjustment effect of the PI regulator to filter out high-frequency components and noise in epsilon, so as to ensure the required performance in the loop and increase the stability of the system. The voltage controlled oscillator is controlled by a control voltage such that the frequency of the voltage controlled oscillator is close to the frequency of the input signal until after the frequency difference is removed and then the frequency is locked. The phase-locked loop can eliminate the influence caused by higher harmonics, so that the position of the rotor output by the phase-locked loop is smoother and more stable.

2.4 five bridge arm voltage source inverter duty cycle correction

The method provided by the invention effectively utilizes the zero-voltage vector half period in the traditional modulation method of the five-bridge arm inverter, and when a novel high-frequency injection method is adopted, when the A-phase duty ratio generated by a high-frequency injection signal in the zero-voltage vector half period of the PMSM1 is not equal to the A-phase duty ratio generated by the PMSM2 in the effective voltage vector control half period, the action requirements of a public bridge arm are not consistent; similarly, the A-phase duty ratio generated by the high-frequency injection signal in the zero-voltage vector half period of the PMSM2 and the A-phase duty ratio generated by the PMSM1 in the effective voltage vector control half period are not equal. In order to realize the independent control of the two motors, the duty ratio generated by the high-frequency injection of one motor into a half period and the duty ratio generated by the vector control half period of the other motor need to be corrected, and the correction method can be realized by adopting the existing vector correction method of the five-bridge-arm inverter.

The method is based on the idea of traditional modulation of the five-bridge-arm inverter, high-frequency square wave voltage signals are injected into a static alpha-beta shafting of the two motors, the injection time of effective voltage vectors is separated from the injection time of the high-frequency signals, high-frequency response currents generated by the two motors are used for extracting the rotor position and rotating speed information of the two motors, and the duty ratio of the five-bridge-arm inverter needs to be corrected, so that the two motors are independently controlled. Compared with the traditional control method, the invention omits the use of a mechanical photoelectric encoder and the use of a filter during high-frequency signal extraction and position information demodulation, reduces the system volume, improves the system dynamic performance and improves the system applicability.

The present invention provides a low-speed position sensorless control method for a dual permanent magnet motor of a five-leg inverter, which is described in detail below with reference to the embodiments and the accompanying drawings.

In this example, a DSP (TMS320F28377D) microprocessor of TI corporation is selected for formula calculation, algorithm processing, signal acquisition, and switching tube switching signal generation. The circuit topology structure of the five-bridge-arm inverter double-permanent magnet motor system is shown in figure 1, the left side is provided with a three-phase power grid and an uncontrollable rectifier bridge, wherein V_sa、V_sb、V_scIs a three-phase grid phase voltage; v_dcIs the DC side capacitor voltage; the right side is provided with a five-bridge arm inverter and two Permanent Magnet Synchronous Motors (PMSM) PMSM1 and PMSM2, wherein a bridge arm A, B, C is used for driving a motor PMSM1, and a bridge arm A, D, E is used for driving a motor PMSM 2. In the present invention, the subscripts i (i ═ 1,2) of all variables represent PMSM1 and PMSM2, respectively. It can be seen that compared with the conventional six-leg inverter, the five-leg inverter control system structure reduces the number of power switching devices, and in the method, two motors share the a-phase leg.

In this embodiment, let S_x(x ═ A, B, C, D, E) is a switching function, and when the switch tube on the corresponding bridge arm is switched on and the lower switch tube is switched off, S is made_x1 is ═ 1; on the contrary, when the upper switch tube is turned off and the lower switch tube is turned on, the S is enabled_x0. For each individual motor, the voltage vector controlling each motor comprises 6 effective vectors u_in(n-1, …,6) and 2 zero vectors u_im(m ═ 0,7), as shown in fig. 2. The space voltage vector of each motor is divided into six sectors of I, II, III, IV, V and VI; u. of_i1[1,0,0]，u_i2[1,1,0]，u_i3[0,1,0]，u_i4[0,1,1]，u_i5[0,0,1]，u_i6[1,0,1]6 effective voltage vectors; u. of_i0[0,0,0]，u_i7[1,1,1]The stator voltage reference vectors of PMSM1 and PMSM2 are u as 2 zero voltage vectors_1refAnd u_2refThe phase angle of the reference voltage is theta_e1And theta_e2And at an angular velocity ω, respectively₁And ω₂And (4) rotating.

The control structure diagram of the present invention is shown in fig. 3, a Proportional Integral (PI) controller is adopted for a speed loop, subscripts i of all variables represent a permanent magnet synchronous motor 1 and a permanent magnet synchronous motor 2, i is 1, 2; all variable subscripts d and q respectively represent d-axis and q-axis components under a two-phase rotating d-q coordinate system; all variable subscripts alpha and beta respectively represent alpha axis and beta axis components under a static alpha-beta coordinate system; omega_1refAnd ω_2refReference rotating speeds of the two motors are respectively set; omega₁And ω₂The actual rotating speeds of the two motors are respectively;andrespectively obtaining mechanical rotating speeds observed by the two motors; i.e. i_{d1_ref}And i_{d2_ref}Respectively setting the current values of the d-axis stators of the two motors; i.e. i_d1And i_d2Actual values of d-axis stator currents of the two motors are respectively obtained; i.e. i_{q1_ref}And i_{q2_ref}Respectively setting the q-axis stator current values of the two motors; i.e. i_q1And i_q2Actual values of q-axis stator currents of the two motors are respectively; l is_diAnd L_qiInductance values for d-axis and q-axis, respectively; v_FOCRepresents an effective voltage vector; v_injRepresents a high frequency injection signal; t is_sRepresents one control cycle; FL-VSI represents a five-leg voltage source inverter; PMSM1 and PMSM2 are two permanent magnet synchronous machines.

The voltage equation under the two-phase static coordinate system of the double-permanent magnet synchronous motor is

In the formula u_αi、u_βiStator voltage components of an alpha axis and a beta axis; r_siIs a stator resistor; i.e. i_αi、i_βiStator current components of an alpha axis and a beta axis; p is a differential factor; psi_αi、ψ_βiThe stator flux linkage components are the alpha and beta axes.

ψ_αi、ψ_βiIs shown as

In the formula, L_avgiIs mean inductance, denoted as L_avgi＝(L_di+L_qi)/2；L_difiIs a differential inductance, denoted L_difi＝(L_di-L_qi)/2；ψ_fiIs a rotor permanent magnet flux linkage; theta_eiIs the actual rotor electrical angular position.

Because the frequency of the high-frequency injection signal is far higher than the fundamental wave operating frequency, the influence of the resistance voltage drop of the stator, the rotating voltage and the back electromotive force is ignored, and the built-in permanent magnet synchronous motor can be regarded as a pure inductive load

In the formula u_αhi、u_βhiHigh-frequency voltage components of an alpha axis and a beta axis, respectively; i.e. i_αhi、i_βhiHigh frequency current response components of the alpha and beta axes, respectively.

As shown in FIG. 4, a stationary alpha-beta axis system, a synchronous rotating d-q axis system and an estimated rotation are definedRelationship of axis system, orderObserving the rotor electrical angle positions for the two motors;the difference value between the actual value and the observed value of the electrical angle position of the two motor rotors isConversion matrix R (theta) from static alpha-beta shafting to synchronous rotating d-q shafting_ei) Comprises the following steps:

estimating rotationD-q shafting conversion matrix R (theta) from shafting to synchronous rotation_e) Comprises the following steps:

the control without the position sensor further comprises the following specific steps:

1. the collection and calculation of the electric quantity of the two permanent magnet motor systems comprise:

1.1, the speed loop adopts a PI controller, the actual rotating speeds of the two motors are calculated by adopting a position-sensorless algorithm, the rotating speed error required by the control motor is obtained by subtracting the calculated rotating speed signal from the reference rotating speed of each motor, and the difference value between the reference rotating speed and the actual rotating speed is used for generating a q-axis given current i through the PI controller_{qi_ref}D-axis given current i_{di_ref}Is 0.

1.2, in a control period, collecting three-phase stator currents of two motors by using a Hall current sensor, and performing Clark conversion on the three-phase stator currents in a microprocessor to convert the three-phase stator currents into stator current components on a two-phase static alpha-beta coordinate system:

in the formula i_a1、i_b1、i_c1Are respectively the firstThree-phase stator current of stage motor i_a2、i_b2、i_c2The three-phase stator currents of the second motor are respectively; i.e. i_α1、i_β1The stator current components, i, of the first motor in a stationary two-phase alpha-beta coordinate system_α2、i_β2Respectively, the stator current components of the second motor in the two-phase stationary alpha-beta coordinate system.

1.3, in a control period, rotor position information of the two motors is obtained by adopting a position sensorless algorithm, and stator current components of the two motors on a two-phase static alpha-beta coordinate system are subjected to Park conversion to be converted into stator current components on a two-phase rotating d-q coordinate system.

1.4, collecting direct-current side voltage by using a Hall voltage sensor, and transmitting a collected direct-current side voltage signal to a microprocessor;

2. position sensorless control

As shown in fig. 5, in the conventional modulation strategy of the five-leg inverter, one control cycle is divided into two half cycles with equal length, and in the former half cycle, the first motor performs effective voltage vector modulation, and the second motor performs zero voltage vector modulation, as shown by the shaded part in the figure; in the latter half period, the second motor performs active voltage vector modulation and the first motor performs zero voltage vector modulation, as shown by the shaded portion. The basic idea of the two-motor position-sensorless control method of the five-bridge-arm inverter provided by the invention is as follows: based on a traditional modulation strategy of a five-bridge-arm inverter, selecting zero-voltage vector half cycles of two motors, respectively injecting high-frequency voltage signals into a static beta shaft system of the two motors, separating the injection moments of effective voltage vectors and the high-frequency signals while utilizing the zero-vector half cycles, injecting the high-frequency voltage signals to pass through a motor model to generate response current, and decomposing the response current into fundamental frequency current components and high-frequency current components containing rotor position information; obtaining an orthogonal signal containing rotor position information by demodulating the high-frequency current component; and the rotating speed and the position information of the rotor can be obtained through the position observer.

The conventional high-frequency voltage signal injection method superimposes an injected high-frequency voltage on a magnetic field orientation control voltage, and thus a filter is required when a high-frequency response signal is extracted. When the five-bridge arm voltage source inverter adopts a traditional modulation strategy, a control period is divided into two half periods with equal length, wherein one half period is under the action of an effective vector, and the other half period is under the action of a zero vector. Aiming at the traditional modulation method of the five-bridge arm inverter, a high-frequency voltage signal is injected in the zero vector half period of the five-bridge arm inverter.

In the five-bridge-arm dual-motor high-frequency voltage injection position-free sensor control method, the high-frequency injection time is shown in figure 5, the A phases of two inverters are used as a common phase, the shaded parts in the figure are zero vector half periods of PMSM1 and PMSM2 in a control period respectively, and a high-frequency voltage signal is injected into PMSM2 in the former half period; a high frequency voltage signal is injected into the PMSM1 during the last half cycle.

In order to eliminate voltage errors caused by inverter nonlinearity and realize accurate estimation of the position of a motor rotor, a positive voltage vector is injected in a first control period, and a negative voltage vector is injected in a second control period. In order to reduce the current harmonic of the common bridge arm, a beta axis in a static alpha-beta axis system of the motor is selected for high-frequency square wave injection, because when the amplitude of the injected voltage signals is equal and the phases are opposite, the reference voltage vector is limited in sectors II and V, current oscillation only exists in phases B and C, phase A current theoretically does not have oscillation, and a high-frequency voltage signal injection diagram is shown in FIG. 6.

2.1 high-frequency Signal separation and demodulation

2.1.1, as shown in FIG. 6, after injecting high frequency voltage into the stationary beta axis of the motor, the response current of the stationary alpha-beta axis of the motor contains fundamental frequency current component i_αfiAnd i_βfiWith a high-frequency current component i_αhiAnd i_βhi. The high-frequency current component has the same frequency as the injection frequency and contains rotor position information, so the sampling current can be written as:

in the formula i_αf1、i_βf1Fundamental frequency current components of the first motor on an alpha axis and a beta axis respectively; i.e. i_αf2、i_βf2The fundamental frequency current components of the second motor on an alpha axis and a beta axis respectively; i.e. i_αh1、i_βh1High-frequency response current components of the first motor on an alpha axis and a beta axis respectively; i.e. i_αh2、i_βh2The high-frequency response current components of the second motor on the alpha axis and the beta axis respectively.

2.1.2, in order to extract a high-frequency current component containing rotor position information, in the invention, high-frequency voltage signal injection is carried out in a zero vector half period in a traditional modulation strategy of the five-bridge arm inverter, and the time of an effective voltage vector is separated from the time of a high-frequency injection signal. Let the injection frequency be one-half of the PWM frequency. The current sampling is carried out twice in each PWM period, the sampling time is before and after the high-frequency signal is injected, and the fundamental frequency current can be regarded as a fixed value in two continuous sampling due to the fact that the sampling frequency is high. And in order to eliminate voltage error caused by inverter nonlinearity and realize accurate estimation of the position of the motor rotor, a positive voltage vector is injected in the first control period, and a negative voltage vector is injected in the second control period, as shown in fig. 6. Therefore, in two control periods, the two motors need to perform four times of sampling respectively, and the sampling current is shown as the formula (9).

In the formula (I), the compound is shown in the specification,current sampling values of the motor 1 on an alpha axis and a beta axis at sampling moments 1,2, 3 and 4 respectively;current sampling values of the motor 2 on an alpha axis and a beta axis at sampling moments 1,2, 3 and 4 respectively;the fundamental frequency current components of the motor 1 on the alpha axis and the beta axis at the sampling instants 1,2, 3 and 4 respectively,the fundamental frequency current components of the motor 2 on the alpha and beta axes at the sampling instants 1,2, 3 and 4 respectively, the high-frequency current components on the alpha axis and the beta axis at the sampling time 1,2, 3 and 4 of the motor 1 respectively, the high frequency current components of the alpha and beta axes are sampled at times 1,2, 3, 4 for the motor 2.

Because the motor runs at low speed and the frequency of the injected high-frequency voltage signal is half of the PWM frequency and is far higher than the fundamental frequency, the fundamental frequency current can be regarded as a fixed value in two continuous samplings; therefore, the high-frequency response current component in the first control period and the high-frequency response current component in the second control period can be expressed as:

in the formula (I), the compound is shown in the specification,high-frequency response current components on an alpha axis and a beta axis of a first motor in a first control period respectively;are respectively the second stationHigh-frequency response current components on an alpha axis and a beta axis in a first control period;high-frequency response current components of the first motor on an alpha axis and a beta axis in a second control period respectively;the high frequency response current components on the alpha axis and the beta axis of the second motor in the second control period are respectively.

2.2 high frequency response current envelope extraction

2.2.1, in the signal processing process, obtaining the position information of the rotor by utilizing the high-frequency current component of the static alpha-beta shaft system, and injecting high-frequency square wave voltage into the static alpha-beta shaft system as shown in the formula (11).

2.2.2, the high-frequency mathematical model of the built-in permanent magnet synchronous motor under the static alpha-beta shafting can be obtained by the formula (3), and the formula (11) is substituted for the formula (3) to obtain:

when the injection signal is in the first control period, equation (12) can be summarized as:

in the formula, delta t is half of a control period; delta I_αhi、ΔI_βhiThe high-frequency current variation amounts of the α axis and the β axis, respectively.

When the injection signal is in the second control period, equation (12) can be summarized as:

the equations (13) and (14) are signal-processed, and the high-frequency response current generated in the injection period is multiplied by a sign function, which is expressed as equation (15).

The high-frequency current response envelope of the motor can be obtained as follows:

in the formula i_cosi、i_siniIs a high frequency current envelope signal containing position information.

According to formula (17), I_NAnd I_MThe parameters in the expression are all known quantities, so that the direct current offset I can be obtained by calculation_NAnd coefficient I_MTherefore, simplified formula (16) can be simplified to obtain:

in the formula I_cosi、I_siniFor quadrature signals containing information on the position of the rotor of the machine

2.3 principle and design of position observer

2.3.1, the position observer is a key component for realizing the estimation of the rotor position in the high-frequency square wave injection method, and the invention adopts an orthogonal phase-locked loop to estimate the rotor position. High-frequency response current i in static alpha-beta shafting_αhi、i_βhiObtaining orthogonal signal I containing rotor position information after processing_cosi、I_siniAnd the rotor position information can be extracted from the signal through an orthogonal phase-locked loop control link. A quadrature phase locked loop control block diagram is shown in fig. 7.

The quadrature phase-locked loop is composed of three parts, namely a Phase Detector (PD), a loop oscillator (LPF) and a Voltage Controlled Oscillator (VCO). The effect of the phase detector is to compare the estimated phase with the actual phase, i.e. to compare the two input signals I_cosi、I_siniAnd two output signalsThe processing and comparison is performed to obtain a position error signal:

where ε is the position error signal.

Formula (18) may be substituted for formula (19):

when the estimated rotor position signal limit of the machine approaches the actual rotor position, i.e. whenWhen the formula (21) is satisfied.

The closed loop transfer function of the quadrature phase locked loop is as follows

Wherein s denotes a complex frequency in the complex frequency domain; k_pThe coefficient of a proportional regulator of a proportional integral regulator inside the quadrature phase-locked loop; k_iThe integral regulator coefficient of a proportional integral regulator inside the quadrature phase-locked loop.

The steady state error transfer function of the quadrature phase locked loop obtained from equation (22) is:

when the motor is in steady state operation, the input theta_eiAs a ramp function, the steady state error of the quadrature phase locked loop at this time is:

the system is a second-order system, the bandwidth of the system is directly related to the response capability and the anti-interference capability of the system, and the bandwidth of the system is reduced to improve the anti-interference capability of the system under the condition that proper parameters are selected to ensure that the system can normally run. The parameter selected by the invention is K_p＝141.4，K_i10000, a bode diagram of the system is made through a transfer function as shown in fig. 8, it can be known that the system bandwidth is 49.1Hz at this time, and it can be seen that the system has better dynamic response capability and interference resistance capability.

2.4 five bridge arm voltage source inverter duty cycle correction

The method provided by the invention effectively utilizes the zero-voltage vector half period in the traditional modulation method of the five-bridge arm inverter, and when a novel high-frequency injection method is adopted, when the A-phase duty ratio generated by a high-frequency injection signal in the zero-voltage vector half period of the PMSM1 is not equal to the A-phase duty ratio generated by the PMSM2 in the effective voltage vector control half period, the action requirements of a public bridge arm are not consistent; similarly, the A-phase duty ratio generated by the high-frequency injection signal in the zero-voltage vector half period of the PMSM2 and the A-phase duty ratio generated by the PMSM1 in the effective voltage vector control half period are not equal. In order to realize the independent control of the two motors, the duty ratio generated by the high-frequency injection of one motor into a half period and the duty ratio generated by the vector control half period of the other motor need to be corrected, and the correction method can be realized by adopting the existing vector correction method of the five-bridge-arm inverter.

The method is based on the idea of traditional modulation of the five-bridge-arm inverter, high-frequency square wave voltage components are injected into a static alpha-beta shafting of the two motors, the effective voltage vectors are separated from the injection time of high-frequency signals, the high-frequency response currents generated by the two motors are used for extracting the rotor position and rotating speed information of the two motors, and the duty ratio of the five-bridge-arm inverter needs to be corrected, so that the two motors are independently controlled. Compared with the traditional control method, the invention omits the use of a mechanical photoelectric encoder and the use of a filter during high-frequency signal extraction and position information demodulation, reduces the system volume, improves the system dynamic performance and improves the system applicability.

In order to verify the effectiveness of the control method of the five-bridge-arm inverter double-permanent-magnet motor position-sensorless control method, experiments are carried out on two 2.8kW IPMSM, and an experiment platform adopted in the experiments is shown in FIG. 9. In the experiment, the main control is completed by a 32-bit floating point type dual-core digital processor TMS320F28377 produced by TI company and a Cyclic V series FPGA produced by Intel company, wherein the DSP is mainly responsible for an algorithm execution part, and the FPGA is mainly responsible for external ADC sampling, DAC conversion, pulse distribution and the like. The experiment adopts a 6000-line incremental encoder to measure the actual rotor position and the rotating speed information of the motor, and is used for comparing the actual rotor position with the observed rotor position, and the experiment results are obtained in a non-position control mode.

TABLE 1 Motor parameters

The switching frequency used in the experiment was 5KHz, the injection voltage amplitude was 60V, and the frequency was half of the switching frequency, and the two motors had the same parameters, as shown in table 1.

The experiment is analyzed from the following working conditions, wherein the first working condition is that the two motors are unloaded and speed step change is realized, the specific working condition is that the PMSM1 is started at the rotating speed of 100r/min, the rotating speed is stepped to 300r/min at 3s, and the rotating speed is stepped to 100r/min at 6 s; the PMSM2 operates at a constant speed of 300 r/min. As shown in fig. 10, the actual rotor speed, the estimated rotor speed, the rotor speed error, and the actual rotor position, the estimated rotor position, and the rotor position error of PMSM1 and PMSM2 are given, respectively. It can be seen from the figure that under the working condition, the rotating speed tracking of the two motors is better, and the rotating speed tracking error is small; the position error of the PMSM1 is 9 degrees at most, the average error is 6.5 degrees, the position error of the PMSM2 is 7 degrees at most, the average error is 5.6 degrees, and the position estimation precision is high. The experimental result shows that the two motors can independently run and have good dynamic speed control performance.

The method provided by the invention avoids the use of a filter in the process of extracting the high-frequency signal, as shown in fig. 11, the method is a quadrature signal of the rotor positions of the two motors under the first working condition, so that the quadrature of the sine and cosine signals of the rotor positions of the two motors can be seen, and the effectiveness of the method can be verified by combining the operation result of fig. 10.

The second working condition is an operation working condition when the load of the motor is increased or reduced, the specific working condition is that the PMSM1 applies 3Nm load and starts at the rotating speed of 300r/min, the load is stepped to 6Nm when 3s, and the load is stepped to 3Nm when 6 s; the PMSM2 applied a 6Nm load and was operated at a constant speed of 300 r/min. As shown in fig. 12, the actual rotor speed, the estimated rotor speed, the rotor speed error, and the actual rotor position, the estimated rotor position, and the rotor position error of PMSM1 and PMSM2 are given, respectively. It can be seen from the figure that under the working condition, the rotating speed tracking of the two motors is better, and the rotating speed tracking error is small; the position error of the PMSM1 is 7 degrees at most, the average error is 6 degrees, the position error of the PMSM2 is 5.5 degrees at most, the average error is 5 degrees, and the position tracking precision is high. The experimental result shows that the two motors can independently run and have good dynamic torque control performance.

As shown in fig. 13, the waveforms of the three-phase current and the d-q axis current of the two motors under the second working condition show that the motor current can be smoothly transited when the motors are subjected to sudden loading and sudden unloading, the dynamic performance is good, the common phase of the two motors, namely, the harmonic content of the phase a current is obviously less than that of the other two phases, and the effectiveness of the method provided by the invention can be verified by combining the operation result of fig. 12.

The third working condition is the running condition of the motor with constant load and when the motor rotates positively and negatively, the specific working condition is that the PMSM1 applies 6Nm load and starts up at the rotating speed of 200r/min, the speed is stepped to-200 r/min at 3s, and the rotating speed is stepped to 200r/min at 7 s; the PMSM2 applied a 3Nm load and started at 300r/min, stepped to-300 r/min at 4s and 300r/min at 8 s. As shown in fig. 14, the actual rotor speed, the estimated rotor speed, the rotor speed error, and the actual rotor position, the estimated rotor position, and the rotor position error of PMSM1 and PMSM2 are given, respectively. It can be seen from the figure that under the working condition, the rotating speed tracking of the two motors is better, and the rotating speed tracking error is small; the position error of the PMSM1 is 15 degrees at most, the average error is 5.6 degrees, the position error of the PMSM2 is 12 degrees at most, the average error is 4.7 degrees, the position tracking precision is high, and the experimental result shows that the two motors can independently run and have good dynamic control performance.

As shown in fig. 15, the waveforms of d-q axis currents of the two motors under the third working condition show that the currents of the rotating shafts of the two motors are jittered when the two motors are loaded and rotate in the forward and reverse directions, but the overall control effect is better and the dynamic performance is good.

In summary, the invention provides a position sensorless control method for a five-bridge-arm inverter-driven double-permanent magnet synchronous motor at a low speed. Firstly, based on the idea of traditional modulation of a five-bridge arm inverter, high-frequency signal injection is carried out in a zero vector half period, so that separation of an effective voltage vector and high-frequency signal injection time is realized, and the use of a filter during high-frequency signal extraction and position information demodulation is omitted; secondly, high-frequency square wave signals are injected into a beta axis of a motor static shafting, so that current resonance of a five-bridge-arm public bridge arm is reduced; then, the duty ratio of the five-bridge-arm inverter needs to be corrected, so that the two motors can be independently controlled; finally, experiments prove that the method provided by the invention can accurately estimate the position of the motor rotor when the two motors independently operate at low speed, and the method reduces the system volume, improves the system dynamic performance, increases the system reliability and improves the system applicability.

While the present invention has been described with reference to the accompanying drawings, the present invention is not limited to the above-described embodiments, which are intended to be illustrative rather than restrictive, and many modifications may be made by those skilled in the art without departing from the spirit of the present invention within the scope of the appended claims.