阅读说明：本技术 确定无刷永磁电机的转子的位置的方法 (Method for determining the position of the rotor of a brushless permanent magnet motor ) 是由 N.施密特 于 2020-03-19 设计创作，主要内容包括：一种确定无刷永磁电机(14)的转子(18)位置的方法(100)包括在电机(14)的相绕组(22)激励期间测量(102)流过相绕组(22)的相电流,以及在相绕组激励期间测量(103)施加到电机的相绕组的电压。该方法(100)包括使用测量相电流和测量电压计算(104)在相绕组(22)中感应的反EMF的相位。该方法(100)包括使用在相绕组(22)中感应的反EMF的计算相位来确定(106)在相绕组(22)中感应的反EMF的过零点。该方法(100)包括当在相绕组(22)中感应的反EMF处于过零点时,确定(108)无刷永磁电机的转子(18)的对准位置。(A method (100) of determining a position of a rotor (18) of a brushless permanent magnet motor (14) includes measuring (102) phase current flowing through a phase winding (22) of the motor (14) during excitation of the phase winding (22), and measuring (103) voltage applied to the phase winding of the motor during excitation of the phase winding. The method (100) includes calculating (104) a phase of a back EMF induced in the phase winding (22) using the measured phase currents and the measured voltages. The method (100) includes determining (106) zero-crossings of back EMF induced in the phase winding (22) using a calculated phase of the back EMF induced in the phase winding (22). The method (100) includes determining (108) an alignment position of a rotor (18) of the brushless permanent magnet motor when a back EMF induced in the phase winding (22) is at a zero crossing.)

1. A method of determining the position of a rotor of a brushless permanent magnet electric machine, the method comprising measuring instantaneous phase current flowing through a phase winding of the electric machine during excitation of the phase winding; measuring a voltage applied to a phase winding of the electric machine during excitation of the phase winding; calculating a phase of a back EMF induced in the phase winding using the measured instantaneous phase current and the measured voltage; determining zero-crossings of the back EMF induced in the phase winding using the calculated phase of the back EMF induced in the phase winding; and determining an alignment position of a rotor of the brushless permanent magnet motor when the back EMF induced in the phase windings is at a zero crossing.

2. The method of claim 1, wherein an equation is usedCalculating the phase of the back EMF induced in the phase winding, wherein E is the back EMF induced in the phase winding, L is the inductance of the phase winding, I is the phase current flowing through the phase winding, R is the resistance of the phase winding, and V_PHIs the voltage across the phase winding.

3. A method as claimed in claim 1 or 2, wherein the method comprises measuring phase current applied to a plurality of excitation pulses of the phase winding.

4. A method as claimed in claim 3, wherein the method comprises measuring the phase current applied to each energising pulse of the phase winding.

5. The method of any preceding claim, wherein measuring the current flowing through the phase windings of the electric machine comprises measuring an average phase current flowing through the phase windings of the electric machine and/or measuring an instantaneous phase current flowing through the phase windings of the electric machine.

6. The method of any preceding claim, wherein measuring the voltage applied to a phase winding of the electric machine during excitation of the phase winding comprises measuring an average DC voltage and/or an instantaneous DC voltage applied to a phase winding of the electric machine during excitation of the phase winding.

7. A method as claimed in any preceding claim, wherein the method comprises commutating the phase winding relative to a determined zero crossing of the back EMF induced in the phase winding.

8. The method of any preceding claim, wherein determining a zero crossing of the back EMF induced in the phase winding comprises using any one or any combination of a calculated phase of the back EMF induced in the phase winding, an amplitude representative of an amplitude of the back EMF induced in the phase winding, and a frequency representative of a frequency of the back EMF induced in the phase winding.

9. The method of any preceding claim, wherein calculating the phase of the back EMF induced in the phase winding comprises commutingIntegration is performed to obtain a relationship representing the integrated back EMF.

10. The method of claim 9, wherein calculating the phase of the back EMF induced in the phase winding comprises equating the integrated back EMF as an integral of a sinusoidal waveform representing the back EMF induced in the phase winding.

11. The method of claim 10, wherein calculating the phase of the back EMF induced in the phase winding comprises equating a relationship representing an integrated back EMF as an integral of a sinusoidal waveform representing the back EMF induced in the phase winding.

12. The method of any of claims 1 to 8, wherein calculating the phase of the back EMF induced in the phase winding comprises calculating at least one instantaneous back EMF value using the measured instantaneous phase currents and the measured voltage.

13. The method of claim 12, wherein an equation is usedTo calculate the instantaneous back EMF value.

14. The method of claim 12 or 13, wherein calculating the phase of the back EMF induced in the phase winding comprises integrating the calculated instantaneous back EMF value.

15. A brushless permanent magnet electric machine comprising a controller configured to measure instantaneous phase current flowing through a phase winding of the electric machine during excitation of the phase winding; measuring a voltage applied to a phase winding of the electric machine during excitation of the phase winding; calculating a phase of a back EMF induced in the phase winding using the measured instantaneous phase current and the measured voltage; determining zero-crossings of the back EMF induced in the phase winding using the calculated phase of the back EMF induced in the phase winding; and determining an alignment position of a rotor of the brushless permanent magnet motor when the back EMF induced in the phase windings is at a zero crossing.

16. A data carrier comprising machine readable instructions for operation of a controller of a brushless permanent magnet motor to measure instantaneous phase current flowing through a phase winding of the motor during excitation of the phase winding; measuring a voltage applied to a phase winding of the electric machine during excitation of the phase winding; calculating a phase of a back EMF induced in the phase winding using the measured instantaneous phase current and the measured voltage; determining zero-crossings of the back EMF induced in the phase winding using the calculated phase of the back EMF induced in the phase winding; the alignment position of the rotor of the brushless permanent magnet motor is determined when the back EMF induced in the phase windings is at a zero crossing.

Technical Field

The present invention relates to a method of determining the position of a rotor of a brushless permanent magnet electric machine.

Background

Knowledge of the rotor position is essential in order to commutate the phase windings of a brushless permanent magnet motor at the correct time. Brushless permanent magnet motors typically include hall effect sensors that output signals indicative of the position of the rotor. While the component costs of the sensor are relatively inexpensive, integrating the sensor within the motor typically complicates the design and manufacture of the motor. In addition, the signal output by the sensor is typically susceptible to electromagnetic noise generated within the motor.

Sensorless solutions for indirectly determining the rotor position are known. For permanent magnet motors, the transition in polarity of the back EMF induced in the phase windings can be used to determine rotor position. For a multi-phase motor, rotor position may be determined by sensing the back EMF induced in the non-energized phase windings. For single phase motors, this type of control is not feasible due to the lack of additional phase windings. However, the position of the rotor may be determined by suspending the excitation at a point in the electrical cycle where a polarity transition of the back EMF is expected. Unfortunately, suspending energization has the disadvantage of reducing the electrical energy that can be driven into the motor.

A sensorless solution to alleviate the above drawbacks has previously been proposed in PCT patent application WO 2013/132249. While this approach may alleviate the above-mentioned disadvantages to some extent, the approach disclosed therein utilizes a complex hardware arrangement, which may increase the overall cost of the motor control system.

Disclosure of Invention

According to a first aspect of the present invention, there is provided a method of determining the position of a rotor of a brushless permanent magnet electric machine, the method comprising measuring phase current flowing through a phase winding of the electric machine during excitation of the phase winding; measuring a voltage applied to a phase winding of the electric machine during excitation of the phase winding; calculating a phase of a back EMF induced in the phase winding using the measured phase current and the measured voltage; determining zero-crossings of the back EMF induced in the phase winding using the calculated phase of the back EMF induced in the phase winding; and determining an alignment position of a rotor of the brushless permanent magnet motor when the back EMF induced in the phase windings is at a zero crossing.

The method according to the first aspect of the present invention may in principle be advantageous in that the method comprises calculating a phase of the back EMF induced in the phase winding using the measured phase currents and the measured voltages, determining a zero crossing of the back EMF induced in the phase winding using the calculated phase of the back EMF induced in the phase winding, and determining the alignment position of the rotor of the brushless permanent magnet motor when the back EMF induced in the phase winding is at the zero crossing.

In particular, it is known that the back EMF of a brushless permanent magnet motor can have a substantially sinusoidal form, and the amplitude and frequency of the back EMF induced in the phase winding can be obtained from past measurements or simulations or from real-time calculations. By using the measured instantaneous phase currents and the measured voltages to calculate the phase of the back EMF induced in the phase winding, this information can then be used in conjunction with known amplitudes and frequencies to provide a relatively accurate representation of the waveform of the back EMF induced in the phase winding. The waveform representation of the back EMF induced in the phase windings can then be used to determine the zero-crossing of the back EMF induced in the phase windings and hence the alignment position of the rotor when the back EMF induced in the phase windings crosses zero.

The method according to the first aspect of the invention is advantageous in that it allows to determine zero-crossing points, wherein the zero-crossing points are located outside the excitation period of the phase winding. In particular, by utilizing a representation of the back EMF induced in the phase winding in the manner described above, zero crossings can be determined regardless of whether they are located within the excitation period of the phase winding. This enables efficient operation over a wider power range than, for example, arrangements in which the rotor position is determined by hardware. In particular, a motor in which the zero-crossing points can only be determined during the excitation period may have a lower operating efficiency for a given power than a motor controlled according to the method of the first aspect of the invention, since zero-crossing points lying outside the excitation period cannot be determined with any accuracy, resulting in a commutation inefficiency.

For example, the method may be performed using software, rather than using hardware. Thus, the number of components and/or the overall cost of the control system implementing the method according to the first aspect of the invention may be reduced, for example with respect to schemes using hardware to predict or calculate zero-crossings of the back EMF.

The zero crossing of the back EMF induced in the phase winding refers to the point at which the back EMF value reaches zero during the transition between a positive polarity back EMF value and a negative polarity back EMF value, and vice versa.

Equations may be usedCalculating the phase of the back EMF induced in the phase winding, wherein E is the back EMF induced in the phase winding, L is the inductance of the phase winding, I is the phase current flowing through the phase winding, R is the resistance of the phase winding, and V_PHIs the voltage across the phase winding.

The method may include measuring phase current applied to a plurality of excitation pulses of the phase winding, such as to each excitation pulse of the phase winding or to every other excitation pulse of the phase winding. Measuring the phase current applied to each excitation pulse of the phase winding may be beneficial because it may improve the accuracy of the determined zero crossings. The method may include measuring phase current at the beginning of each excitation pulse applied to the phase winding and at the end of each excitation pulse applied to the phase winding. The method may include measuring phase current for substantially the entire duration of each excitation pulse applied to the phase winding.

Measuring the current flowing through the phase windings of the motor may include measuring an average phase current flowing through the phase windings of the motor and/or measuring an instantaneous phase current flowing through the phase windings of the motor.

Measuring the voltage applied to the phase windings of the motor during excitation of the phase windings may include measuring an average DC voltage and/or measuring an instantaneous DC voltage applied to the phase windings of the motor during excitation of the phase windings.

The method may include commutating the phase winding relative to a determined zero-crossing of a back EMF induced in the phase winding. For example, the method may comprise advancing the commutation of the phase winding relative to determining a zero crossing, and/or the method may comprise commutating the phase winding synchronously with determining a zero crossing, and/or the method may comprise delaying the commutation of the phase winding relative to determining a zero crossing.

Determining an alignment position of a rotor of the brushless permanent magnet motor when the back EMF induced in the phase windings is at a zero-crossing may include determining a future alignment position of the rotor when the back EMF is at a future zero-crossing. For example, using the calculated phase of the back EMF induced in the phase winding to determine zero crossings of the back EMF induced in the phase winding may include determining future zero crossings of the back EMF induced in the phase winding.

The back EMF induced in the phase winding may comprise a sinusoidal waveform, e.g., having an amplitude and frequency. Determining the zero-crossing of the back EMF induced in the phase winding may comprise using any one or any combination of the calculated phase of the back EMF induced in the phase winding, an amplitude representative of the amplitude of the back EMF induced in the phase winding, and a frequency representative of the frequency of the back EMF induced in the phase winding.

The amplitude indicative of the amplitude of the back EMF induced in the phase winding may comprise a predetermined amplitude and/or the frequency indicative of the frequency of the back EMF induced in the phase winding may comprise a predetermined frequency. For example, the predetermined amplitude and/or the predetermined frequency may be obtained by pre-measurement or simulation and may be stored in a memory of a controller of the brushless permanent magnet motor. The amplitude representing the amplitude of the back EMF induced in the phase winding may comprise, for example, a calculated amplitude calculated in real time, and/or the frequency representing the frequency of the back EMF induced in the phase winding may comprise, for example, a calculated frequency calculated in real time. The amplitude representing the amplitude of the back EMF induced in the phase winding and/or the frequency representing the frequency of the back EMF induced in the phase winding may be velocity dependent. For example, higher rotational speeds of the rotor of a brushless permanent magnet motor may result in greater amplitude and/or frequency.

Calculating the phase of the back EMF induced in the phase winding may include the equation The integration is performed to obtain a relationship representing the integrated back EMF, e.g., a relationship between the integrated back EMF, the integrated and/or instantaneous phase current flowing through the phase winding, and the integrated voltage applied to the phase winding. Equation of pairsPerforming integration may include integrating an equation between boundaries set by phase angles at the beginning and end of an excitation pulse applied to a phase winding.

Calculating the phase of the back EMF induced in the phase winding may include equaling the integrated back EMF to an integral of a sinusoidal waveform representing the back EMF induced in the phase winding. Calculating the phase of the back EMF induced in the phase winding may include equating a relationship representing an integrated back EMF with an integration of a sinusoidal waveform representing the back EMF induced in the phase winding.

Calculating the phase of the back EMF induced in the phase winding may include calculating at least one instantaneous back EMF value using the measured instantaneous phase current and the measured voltage. Calculating the phase of the back EMF induced in the phase winding may include calculating at least one instantaneous back EMF value using any one or any combination of the voltage across the phase winding, the DC supply voltage of the motor, the inductance of the phase winding, and the resistance of the phase winding. Calculating the phase of the back EMF induced in the phase windings may include calculating an instantaneous back EMF value for substantially each measured instantaneous phase current value.

Equations may be usedCalculating an instantaneous back EMF value, where E is the back EMF induced in the phase winding, L is the inductance of the phase winding, I is the instantaneous phase current flowing through the phase winding, R is the resistance of the phase winding, and V_PHIs the voltage across the phase winding. The inductance of the phase winding may include a self-inductance of the phase winding and a mutual inductance of the phase winding, e.g., a difference between the self-inductance of the phase winding and the mutual inductance of the phase winding.

Calculating at least one instantaneous back EMF value using the measured instantaneous phase currents may include calculating an instantaneous back EMF value applied to a plurality of excitation pulses of the phase winding, such as an instantaneous back EMF value applied to each excitation pulse of the phase winding. Calculating at least one instantaneous back EMF value using the measured instantaneous phase currents may include calculating an instantaneous back EMF value at the beginning of each excitation pulse applied to the phase winding and at the end of each excitation pulse applied to the phase winding.

Calculating the phase of the back EMF induced in the phase winding may include integrating the calculated instantaneous back EMF value, for example, between boundaries set by phase angles at the beginning and end of an excitation pulse applied to the phase winding. This may be beneficial because it may be able to calculate the phase of the back EMF induced in the winding while also filtering any noise that may be introduced during calculation of the instantaneous back EMF value using the measured instantaneous phase currents, for example, noise introduced by differentiation.

The method may include normalizing the integrated calculated instantaneous back EMF value.

According to a second aspect of the present invention there is provided a brushless permanent magnet electric machine comprising a controller configured to measure instantaneous phase current flowing through a phase winding of the electric machine during excitation of the phase winding, measure voltage applied to the phase winding of the electric machine during excitation of the phase winding, calculate a phase of a back EMF induced in the phase winding using the measured instantaneous phase current and the measured voltage, determine zero crossings of the back EMF induced in the phase winding using the calculated phase of the back EMF induced in the phase winding, and determine an alignment position of a rotor of the brushless permanent magnet electric machine when the back EMF induced in the phase winding is at the zero crossings.

According to a third aspect of the present invention there is provided a data carrier comprising machine readable instructions for operation of a controller of a brushless permanent magnet motor to measure instantaneous phase current flowing through a phase winding of the motor during excitation of the phase winding, to measure voltage applied to the phase winding of the motor during excitation of the phase winding, to calculate a phase of a back EMF induced in the phase winding using the measured instantaneous phase current and the measured voltage, to determine zero-crossing points of the back EMF induced in the phase winding using the calculated phase of the back EMF induced in the phase winding, and to determine an alignment position of a rotor of the brushless permanent magnet motor when the back EMF induced in the phase winding is at the zero-crossing points.

Preferred features of each aspect of the invention may be equally applicable to other aspects of the invention where appropriate.

Drawings

For a better understanding of the present invention, and to show more clearly how it may be carried into effect, reference will now be made, by way of example, to the following drawings in which:

FIG. 1 is a block diagram of an electric motor system according to the present invention;

FIG. 2 is a schematic view of the motor system of FIG. 1;

FIG. 3 details an enable state of an inverter of the motor system of FIG. 1 in response to a control signal issued by a controller of the motor system;

FIG. 4 is a schematic flow chart illustrating a method according to the present invention; and

figure 5 is a graph of back EMF versus sampling time illustrating a portion of the method of figure 4.

Detailed Description

The motor system 10 of fig. 1 and 2 is powered by an AC mains power supply 12 and includes a brushless motor 14 and a control system 16. Although the embodiments described herein relate to a brushless permanent magnet motor powered by an AC mains power supply 12, it will be appreciated that the teachings herein are also applicable to a brushless permanent magnet motor powered by a DC mains power supply, with appropriate modifications apparent to those skilled in the art.

The motor 14 includes a four-pole permanent magnet rotor 18 that rotates relative to a four-pole stator 20. The wire is wound around the stator 20 and coupled together (e.g., in series or parallel) to form a single phase winding 22. Although the embodiments described herein relate to a single phase brushless permanent magnet motor 14, it should be understood that the teachings herein are also applicable to a multi-phase, e.g., three-phase brushless permanent magnet motor, with appropriate modifications apparent to those skilled in the art.

The control system 16 includes a rectifier 24, a DC link filter 26, an inverter 28, a gate driver module 30, a main supply voltage sensor 32, a current sensor 34, a back EMF sensor 36, and a controller 38. Those skilled in the art will appreciate that the back EMF sensor 36 herein is part of the controller and indeed may be considered to comprise an algorithm running on the controller 36.

Rectifier 24 is a full wave bridge D1-D4 that rectifies the output of AC main power supply 12 to provide a DC link voltage.

The DC-link filter 26 includes a capacitor C1 that smoothes relatively high frequency ripples generated by the switching of the inverter 28.

The inverter 28 includes a full bridge of four power switches Q1-Q4 that couple the DC link voltage to the phase windings 22. Each switch Q1-Q4 includes a freewheeling diode.

The gate driver module 30 drives the opening and closing of the switches Q1-Q4 in response to control signals received from the controller 38.

The main supply voltage sensor 32 outputs a signal V to the controller 38 and to the back EMF sensor 36_DCWhich determines the operating conditions of the motor system 10 in the steady-state mode, as will be discussed in more detail below.

The current sensor 34 includes a sense resistor R1 located on the low side of the inverter 28. The voltage across the current sensor 34 is output as a current SENSE signal I SENSE to the back EMF sensor 36 and the controller 38.

The back EMF sensor 36 generates a digital signal BEMF that is used by the controller 38 to determine the signal provided to the gate driver module 30.

The controller 38 includes a microcontroller having a processor, a memory device, and a plurality of peripherals (e.g., ADCs, comparators, timers, etc.). The storage device stores instructions executed by the processor, as well as control parameters used by the processor (e.g., current limits, rise time thresholds, speed thresholds, freewheel cycles, advance cycles, delay cycles, power-on cycles, etc.).

The controller 38 is responsible for controlling the operation of the motor 14 and generates four control signals S1-S4 for controlling each of the four power switches Q1-Q4. The control signals are output to the gate driver module 30, and in response, the gate driver module 30 drives the opening and closing of the switches Q1-Q4.

FIG. 3 summarizes the enable states of the switches Q1-Q4 in response to the control signals S1-S4 output by the controller 38. Hereinafter, the terms "set" and "clear" will be used to indicate that the signals are logically pulled high and low, respectively, and vice versa. As can be seen in fig. 3, the controller 38 sets S1 and S4 and clears S2 and S3 to energize the phase winding 22 from left to right. Conversely, the controller 38 sets S2 and S3 and clears S1 and S4 to energize the phase winding 22 from right to left. The controller 38 clears S1 and S3 and sets S2 and S4 to freewheel the phase winding 22. Freewheeling enables current in the phase winding 22 to recirculate around the low-side loop of the inverter 28. In this embodiment, the power switches Q1-Q4 are capable of bidirectional conduction. Thus, the controller 38 closes the two low-side switches Q2, Q4 during freewheeling such that current flows through the switches Q2, Q4 instead of the less efficient diodes. It is envisioned that inverter 28 may include power switches that conduct in only a single direction. In this case, the controller 38 will clear S1, S2, and S3 and set S4 to freewheel the phase winding 22 from left to right. The controller 38 will then clear S1, S3, and S4 and set S2 to freewheel the phase winding 22 from right to left. The current in the low side loop of the inverter 28 then flows down through the closed low side switch (e.g., Q4) and up through the diode of the open low side switch (e.g., Q2).

Controller 38 operates in one of three modes depending on the speed of rotor 18. When rotor 18 is stationary, controller 38 operates in a start mode that is only used to start rotor 18 moving in the forward direction. Once the rotor 18 moves forward, the controller 38 switches to the acceleration mode. Controller 38 operates in the acceleration mode until the speed of rotor 18 exceeds a speed threshold, after which controller 38 switches to the steady-state mode. Within each mode of operation, the controller 38 employs a different scheme to control the motor 14 without the need for a dedicated rotor sensor.

Start-up mode

When operating in the start mode, controller 38 does not attempt to determine the position of rotor 18. In contrast, controller 38 energizes phase windings 22 in a predetermined sequence, which ensures that rotor 18 is driven in the forward direction regardless of the position at which rotor 18 has stopped.

The details of the start-up mode are not relevant to the present invention and will not be described here for the sake of brevity. A suitable start-up mode may be, for example, the start-up mode described in PCT patent application WO 2013/132249.

Acceleration mode

When operating in the acceleration mode, the controller 38 needs to know the position of the rotor 18, and therefore the controller 38 operates the first sensorless scheme to determine the position of the rotor 18. The details of the first sensorless scheme are not relevant to the present invention and will not be described here for the sake of brevity. A suitable acceleration mode and first sensorless control scheme may be, for example, the acceleration mode and first sensorless control scheme described in PCT patent application WO 2013/132249.

Steady state mode

When operating in the steady-state mode, the controller 38 employs a second sensorless approach to determining the position of the rotor 18, and the second sensorless approach corresponds to a method according to the present invention.

The method according to the invention is schematically illustrated in fig. 4, generally designated 100.

The method 100 includes measuring 102 an instantaneous phase current flowing through the phase winding 22 during excitation of the phase winding 22, and measuring 103 an average DC voltage applied to the phase winding 22 during excitation of the phase winding 22. The phase of the back EMF induced in the phase winding 22 is calculated 104 using the measured instantaneous phase current and the measured average DC voltage. The calculated phase of the back EMF induced in the phase winding 22 is used to determine 106 zero-crossings of the back EMF induced in the phase winding 22. The alignment position of rotor 18 is determined 108 when the back EMF induced in phase winding 22 is at a zero crossing.

The method 100 utilizes a back EMF sensor 36, and as will now be explained, the back EMF sensor 36 outputs a digital signal BEMF indicative of zero-crossing points of the back EMF.

Without any significant saturation or significance, the voltage equation for the phase winding 22 can be expressed as:

where E is the back EMF induced in the phase winding 22, L is the inductance of the phase winding 22, I is the instantaneous phase current flowing through the phase winding 22, R is the resistance of the phase winding 22, and V_PHIs the voltage across the phase winding 22. The inductance L may be further divided into self-inductances (denoted as L) of the phase windings 22_s) And mutual inductance (denoted L) of the phase winding 22_m) The difference between them.

As can be seen in fig. 2, and as previously described, the back EMF sensor 36, i.e., the controller 38, receives the signal V from the main supply voltage sensor 32_DCAnd a signal V_DCMay be considered to represent the phase voltage V across the phase winding 22_PH. The back EMF sensor 36 also receives a signal I SENSE from the current sensor 34 that is representative of the instantaneous phase current flowing through the phase winding 22. Considering the nature of the current sensor 34, i.e., the SENSE resistance R1, the signal I SENSE is only output to the back EMF sensor 36 and the controller 38 during excitation of the phase winding 22, i.e., when an excitation pulse is delivered to the phase winding 22.

The inductance L and resistance R of the phase winding 22, which are known quantities, are used in conjunction with the signal V when an excitation pulse is delivered to the phase winding 22_DCAnd I SENSE, the back EMF sensor 36 can calculate the value of the instantaneous back EMF using the voltage equation for the phase winding 22 described above.

It may be that zero-crossings of back EMF occur when the phase winding is energized, in which case it is envisioned that a calculated value of instantaneous back EMF may be used to indicate zero-crossings of back EMF. However, in practice, the calculated value of the instantaneous back EMF may be inaccurate, especially because of noise of the signal (which may be generated by the differential component of the voltage equation). Furthermore, this does not allow to determine zero crossings of the back EMF with any accuracy, which zero crossings occur when the phase winding is not energized. It is therefore desirable to provide an alternative method of determining the zero-crossing of the back EMF induced in the phase winding 22, which the inventors of the present application have devised, as will now be described.

The first method comprises using the signal V_DCAnd I _ SENSE when the pulse is excitedWhen a pulse is delivered to the phase winding 22, the value of the instantaneous back EMF is calculated using the voltage equation for the phase winding 22, as described above. The instantaneous back EMF value is then integrated to remove noise and is equivalent to a sinusoidal waveform representing the back EMF induced in the phase winding 22 so that the phase of the back EMF induced in the phase winding 22 can be calculated. This phase can then be used to determine the zero-crossing of the back EMF induced in the phase winding 22.

These steps are substantially similar to the steps outlined in relation to the second method below, but it is critical that the calculated value of the instantaneous back EMF is integrated here. Integrating each value of the calculated instantaneous back EMF can be computationally intensive, which can lead to processing power and thus size and cost that are difficult to use in practice with this approach.

The inventors of the present application have therefore devised a second method, which will now be described,

the integrated back EMF equation gives the following relationship:

where-a and a are boundary values at the beginning and end of the application of the excitation pulse to the phase winding 22, respectively. This equation can be used to obtain an estimated back EMF integral over the measurement interval, but the estimation requires normalization.

The back EMF induced in the phase winding 22 can also be approximated fairly accurately by a sinusoidal waveform having the following equation:

wherein E (t) is the back EMF, A is the amplitude of the back EMF, ω is the angular frequency of the back EMF in radians/second,is the phase of the back EMF in radians. noise (t) represents any noise present in the back EMF signal.

The integral of the noise component of the back EMF equation is close to zero and therefore effectively negligible.

If we set F_sIs the sampling frequency in the measurement interval from-h to h, we set s to be the time in the sample, t to be the time in seconds, such that s-F_st, then according to the measured phase current sum V_DCThe value bemf int of the value calculation can also be written as the interval [ -h, h in the sample]Estimated integration of the upper sinusoidal back EMF waveform:

if we substitute s ═ F_sω) x, then we get:

it can be seen that the normalization constant of the integration given above is a. (F)_sω) where F_sIs the sampling frequency. It can also be observed that the integral limitIs half of the measurement interval of the angle of the back EMF in radians.

The amplitude a depends linearly on the motor speed, usually by a motor specific constant M_100KDenotes that M_100KIs the amplitude in volts at a speed of 100000 RPM. This constant depends on the motor structure, varies slightly with temperature, and can be determined by the characteristics during the resynchronization phase of the motor 14. The amplitude is thus given by:

wherein f is_RPMIs the motor speed in RPM.

Thus, the normalization constant of the back EMF integral becomes:

expression (60F)_s)/(2·10⁵) Equal to the number of samples per electrical cycle of a four-pole motor at 100000RPM, i.e., at a speed of a specified M constant. This can be considered as a frequency normalization factor, whereas M.10³Can be considered as an amplitude normalization factor.

Thus, it can be seen that for known values of amplitude and frequency of the back EMF, we can calculate the phase of the back EMF induced in the phase winding 22 using the following relationship:

the period is 2 pi for a unit amplitude.

From the integration of the back EMF equation above, we know that:

then, using the measured instantaneous phase current value, the measured average voltage value applied to the phase winding, and converting the denominator argument to radians, we can determineThe value of (c). The phase is then obtained by applying an arcsine function

Once the phase has been calculated 104, the known amplitude and/or frequency values for the given rotor speed, or indeed the calculated amplitude and/or frequency values for the given rotor speed, stored in memory may be combined with the phase to determine 106 zero-crossings of the back EMF induced in the phase winding 22, for example using a representation of the back EMF waveform. An example of such a representation is shown relative to the hypothetical calculated value of instantaneous back EMF in figure 5. The noise-assumed computed value of the instantaneous back EMF is indicated by 200 in fig. 5, and the fitted curve segment is indicated by 204.

The zero-crossings of the back EMF induced in the phase windings correspond to the aligned position of rotor 18. The controller 38 then uses the information about the zero-crossings of the back EMF to commutate the phase winding 22 of the electric machine 14 in a desired manner, i.e., advance commutation with respect to the zero-crossings, synchronize commutation with the zero-crossings, or retard commutation with respect to the zero-crossings.

By using the measured instantaneous phase currents and the measured voltages to calculate the phase of the back EMF induced in the phase winding 22, this information can then be used in conjunction with known amplitudes and frequencies to provide a relatively accurate representation of the waveform of the back EMF induced in the phase winding. The representation of the waveform of the back EMF induced in the phase windings can then be used to determine the zero-crossing of the back EMF induced in the phase windings and hence the alignment position of the rotor when the back EMF induced in the phase windings is at the zero-crossing.

The method 100 is capable of determining zero crossings, where they are outside the excitation period of the phase winding. In particular, by utilizing a representation of the back EMF induced in the phase winding in the manner described above, zero crossings can be determined regardless of whether they are located within the excitation period of the phase winding. This enables efficient operation over a wider power range than, for example, arrangements in which the rotor position is determined by hardware. In particular, a motor controlled in relation to the method according to the first aspect of the invention, in which the zero-crossing points can only be determined during the excitation period, may have a lower operating efficiency for a given power, since the zero-crossing points lying outside the excitation period cannot be determined with any accuracy, resulting in an inefficient commutation.

The method 100 is performed using software rather than using hardware. Thus, the number of components and/or the overall cost of the control system implementing the method 100 may be reduced, for example, relative to schemes that use hardware to predict or calculate zero-crossings of back EMF.